Information
-
Patent Grant
-
6414866
-
Patent Number
6,414,866
-
Date Filed
Monday, November 15, 199924 years ago
-
Date Issued
Tuesday, July 2, 200222 years ago
-
Inventors
-
Original Assignees
-
Examiners
Agents
- Zak, Jr. Esq.; William J.
-
CPC
-
US Classifications
Field of Search
US
- 363 124
- 363 89
- 363 80
- 363 21
- 363 27
- 363 74
- 323 222
- 323 237
- 323 282
- 323 283
- 323 290
- 237 556
- 237 552
-
International Classifications
-
Abstract
An active filter is coupled to a dc line of a converter such as a dc-dc converter. The active filter traps a harmonic frequency component of chopped current on the dc line. A passive EMI filter may also be coupled to the dc line to remove higher harmonic frequency components of the chopped current on the dc line.
Description
BACKGROUND OF THE INVENTION
The present invention relates to electrical power supplies. More specifically, the invention relates to a converter having a dc line.
A down chopper of a dc-dc converter receives dc current from a dc power supply, modulates or “chops” the current to reduce current amplitude, and provides the chopped current on a dc line. The reduction in current is proportional to duty cycle of the chopping.
On-off action of the down chopper creates a pulse train-type pattern in the chopped current. Consequently, the chopped current on the dc line contains harmonic frequency content.
The dc-dc converter may also include an EMI filter for reducing the harmonic content to acceptable limits. A conventional EMI filter includes passive inductance, capacitance and resistance elements, the values of which are determined by power rating, chopping frequency and filter attenuation requirements. The EMI filter typically constitutes a significant amount of the overall weight of the dc-dc converter.
Reducing the weight of EMI filters for dc-dc converters used in aircraft would be very desirable. Reducing the weight would lower aircraft fuel consumption. Lowering the fuel consumption, in turn, would lower the cost of flying the aircraft.
SUMMARY OF THE INVENTION
According to one aspect of the present invention, an apparatus includes a converter having a dc line, and an active filter coupled to the dc line. The converter is operable to provide a pulsed current on the dc line at a chopping frequency; and the active filter is operable to trap a fundamental of the pulsed current. The active filter allows a substantially smaller passive filter to remove remaining harmonic frequency components.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1
is an illustration of a dc-dc converter including an active filter;
FIG. 1
a
is an illustration of a switch of the active filter;
FIG. 2
is an illustration of a fundamental frequency component of resonant current and a symmetrical deadband;
FIG. 3
is an illustration of phase and amplitude controls for the dc-dc converter of
FIG. 1
;
FIGS. 4
a
to
4
d
are illustrations of chopper duty cycle, chopper fundamental current, a sine wave reference signal, and a cosine wave reference signal, respectively; and
FIG. 5
is an illustration of alternative phase and amplitude controls for the dc-dc converter of FIG.
1
.
DETAILED DESCRIPTION OF THE INVENTION
Reference is made to
FIG. 1
, which illustrates a dc-dc converter
100
including a conventional down chopper
102
. A dc power source
104
is on the “source” side of the chopper
102
, and a dc line
106
is also on the source side of the chopper. The chopper
102
modulates or “chops” the current from the dc source
104
and thereby reduces the average amplitude of the chopped current on the dc line
106
. Current flowing through the chopper
102
is regulated by a chopper control
108
. The chopper control
108
generates a DC signal that modulates a solid-state power switch
107
in the chopper
102
. The chopper switch
107
is modulated such that (a) current flows from the dc source
104
onto the dc line
106
while the chopper switch
107
is closed; and (b) the current circulates through a freewheeling diode
110
while the chopper switch
107
is open. The reduction in current LINE on the dc line
106
is proportional to duty cycle of the chopping. An inductor
112
smoothes the current-ripple in the output of the chopper
102
.
The on-off action of the chopper
102
creates a pulse-train type flow pattern in the chopper current I
CHOPPER
(see
FIG. 4
a
). The interval T between pulses is proportional to the pulse width modulated (“PWM”) or chopping frequency f
chop
. That is, T=1/f
chop
.
The chopper current I
CHOPPER
has a dc content and a harmonic frequency content. The harmonic frequency content includes a fundamental frequency component and its associated higher harmonic frequency components. The frequency of the fundamental is equal to the chopping frequency f
chop
of the chopper
102
. To assure a small ripple current in the current on the dc line
106
, the chopping frequency is relatively high (e.g., 20 kHz).
The dc-dc converter
100
further includes an active filter
114
for trapping (by canceling or at least by reducing) the fundamental frequency component that would otherwise appear in the line current I
LINE
on the dc line
106
without the active filter
114
. The active filter
114
includes an adjustable power dissipating element
116
, a tunable trap circuit
118
and a bidirectional first switch
124
coupled between the dc line
106
and the trap circuit
118
. The trap circuit
118
includes a capacitor
126
and an inductor
128
. The power-dissipating element
116
includes a second switch
130
and a resistor
132
.
The bi-directional switch
124
may have an H-bridge configuration operated during positive and negative resonant half-cycles, as shown in
FIG. 1
a.
The bi-directional switch
124
includes transistor-type switches
124
a
and
124
b,
blocking diodes
124
c
and
124
d,
and reverse-current diodes
124
e
and
124
f.
The first switch
124
is closed to couple the trap circuit
118
to the dc line
106
. The second switch
130
is closed to discharge the capacitor
126
in the trap circuit
118
.
The fundamental frequency component in the line current I
LINE
is minimized by two independent decoupled controls: a phase control
120
and an amplitude control
122
. The phase control
122
modulates the first switch
124
to produce a resonant current I
RES
. The amplitude control
120
modulates the second switch
130
to modulate the amplitude of the resonant current I
RES
in the trap circuit
118
. This is accomplished by increasing or decreasing the charge on the capacitor
126
relative to the dc line
106
.
Appropriate operation of the switches
124
and
130
produces a resonant current I
RES
that is in phase and equal in magnitude with the fundamental frequency component of the chopper current I
CHOPPER
. To achieve this, phase and amplitude of the resonant current I
RES
are controlled. First, the phase of the resonant current I
RES
is controlled with respect to the fundamental frequency current component produced by the chopper
102
. Second, the amplitude of the resonant current I
RES
is adjusted to match the amplitude of the fundamental frequency current component produced by the chopper
102
as the duty cycle of the chopper
102
is varied. As long as the resonant current I
RES
is in phase and has about the same amplitude as the fundamental frequency component of the chopper current I
CHOPPER
, the fundamental frequency component will not appear in the current I
LINE
on dc line
106
.
Additional reference is made to FIG.
2
. The phase of the resonant current I
RES
may be adjusted by creating a first dead-band DB
1
before the first switch
124
is closed and a second dead band DB
2
after the first switch is closed. During a deadband DB
1
or DB
2
, little to no current flows through the trap circuit
118
. The first deadband DBl may be created by holding-off the gating of the first switch
124
(which is in series with the resonant LC components
126
and
128
of the trap circuit
118
) every half cycle. The time during which the gating is held off may be adjusted so that the first and second deadbands DB
1
and DB
2
are symmetrical or asymmetrical.
The amplitude of the resonant current I
RES
may be adjusted by controlling the voltage of the capacitor
126
just prior to closing the first switch
124
. This is accomplished by closing and then opening the second switch
130
. Consider two cases: (1) the voltage Vcap across the capacitor
126
is less than the dc line voltage Vdc across a capacitor
142
(that is, Vcap <Vdc), and (2) the capacitor voltage Vcap is greater than the dc line voltage Vdc (that is, Vcap>Vdc). When Vcap<Vdc just prior to closing the first switch
124
, the momentary closing of second switch
130
results in the capacitor
126
being (partially) discharged through the resistor
132
to a grounded negative potential and the peak of the subsequent resonant current I
RES
half-cycle will increase. Similarly, when Vcap>Vdc just prior to closing the switch
124
, the momentary closing of second switch
130
(partially) discharges the capacitor
124
but, this time, the peak of the resonant current I
RES
during the subsequent resonant half-cycle will decrease.
The frequency of the resonant current I
RES
is selected to be higher (e.g., approximately
10
% higher) than that of the chopper fundamental current I
FUND
produced by the chopper
12
to accommodate a range of amplitude and phase values that are associated with changes in the duty cycle of the chopper
12
. That is, the higher frequency allows space for phase-shifting using the aforementioned deadbands DB
1
and DB
2
.
The dc-dc converter
100
further includes a conventional EMI filter
134
coupled to the dc line
106
. The EMI filter
134
, which may have a standard construction (e.g., passive elements
136
,
138
,
140
and
142
), is designed to filter the remaining harmonic components from the current I
LINE
on dc line
106
. Since the active filter
114
has already trapped the fundamental frequency component from the line current
LINE
, the EMI filter
134
does not have to deal with the fundamental frequency component. Therefore, the size of inductors
138
and
140
in the EMI filter
134
can be reduced. Reducing the inductances of the EMI filter
134
can result in a significant decrease in the overall weight, since the reduction in the EMI filter inductor weight is greater than the corresponding weight increase due to the addition of the active filter
114
. Thus, the combined weight of the active filter
114
and the EMI filter
134
is lower than the weight of a corresponding conventional EMI filter.
FIG. 1
shows that the three currents I
LINE
, I
RES
and I
CHOPPER
are measured by three sensors
144
,
146
and
148
. However, the two sensors that measure the residual and chopper currents I
RES
and I
CHOPPER
are not used by the controls
120
and
122
of the active filter
114
and are shown only for descriptive purposes.
FIG. 3
shows the amplitude and phase controls
120
and
122
in greater detail. The fundamental in-phase and quadrature components of the line current I
LINE
are used to control the phase and amplitude of the resonant current I
RES
The fundamental in-phase and quadrature components of the line current I
LINE
are extracted by a synchronous demodulator
200
, which heterodynes the measured line current I
LINE
with respect to two reference signals: (1) a sine wave S
IP
for phase control and (2) a cosine wave C
Q
for amplitude control. Both reference signals S
IP
and C
Q
are synchronized with respect to the PWM frequency of the chopper
102
.
FIG. 4
a
shows the duty cycle of the chopper
12
,
FIG. 4
b
shows the chopper fundamental current,
FIG. 4
c
shows the sine wave reference signal S
IP
, and
FIG. 4
d
shows the cosine wave reference signal C
Q
.
Output signals S
1
and C
1
of the synchronous demodulator
200
are passed through low pass filters
202
and
204
to produce two filtered signals S
2
and C
2
. Due to the heterodyning action of the synchronous demodulator
200
, the filtered signals S
2
and C
2
are decoupled dc signals that represent the in-phase and quadrature components of the fundamental frequency content of the line current I
LINE
.
The filtered signal S
2
representing the in-phase component may be used as amplitude feedback, and the filtered signal C
2
representing the quadrature component may be used as phase feedback. A first summing element
206
generates an in-phase error signal P
ERR
representing the difference between the phase feedback C
2
and a reference phase signal P
REF
. The error signal P
ERR
is integrated over a complete “sampling period” by a phase shift regulator
208
, where the sampling period equals a chopping frequency cycle. The output of the phase shift regulator
208
is used to determine when to close the first switch
124
and thereby adjust the symmetry of the dead-band until the phase error signal P
ERR
vanishes. When the reference phase signal P
REF
=0 and the phase error signal P
ERR
vanishes, the deadband should be symmetrical.
A second summing element
210
generates an amplitude error signal A
ERR
representing the difference between the amplitude feedback S
2
and a reference amplitude signal A
REF
. The amplitude error signal A
ERR
is integrated over a complete sampling period by an amplitude control regulator
212
. An output of the amplitude control regulator
212
provides the signal S
3
. The magnitude and the polarity (sign) of the signal S
3
is provided to selector logic
214
which is used to modulate the second switch
130
such that the amplitude error signal A
ERR
vanishes. The polarity of the signal S
3
determines the resonant half-cycle in which the second switch
130
closed, and the magnitude of the signal S
3
determines how long the second switch
130
shall be closed. For example, the second switch
130
remains closed while the magnitude is negative, but opens once the amplitude transitions from negative to positive.
Thus, the reference signals P
REF
and A
REF
indicate the desired amount of phase offset and amplitude. The reference signals P
REF
=0 and A
REF
=0 cause the complete trapping of the fundamental frequency component of the chopper current I
CHOPPER
by the active filter
14
.
The invention is not limited to a synchronous demodulator for creating two independent, decoupled control signals that are orthogonal to each other with respect to a common synchronizing signal. Such orthogonal signals may be created by a Park-vector based decoupler.
Referring now to
FIG. 5
, a Park-vector based decoupler
300
is shown in a reference frame synchronized with respect to the chopper current I
CHOPPER
. In this reference frame the d-components and the q-components of the Park-vectors are inherently decoupled. A Park vector inherently contains information on both the instantaneous magnitudes and phase relationship of a rotating field with respect to a reference coordinate system. A Park vector, in general, is a mathematical representation that describes the locus of an electrical quantity in the complex space domain (where time is a parameter). The current Park vector is defined with the vector's amplitude and the vector's direction in spatial relation to the reference coordinate system. A general discussion of Park vectors can be found in P. K. Kovacs, “Transient Phenomena in Electrical Machines,” Elsevier Science Publishing Co. (1984).
A synchronization signal SYNC may be generated by the gating of the chopper switch
107
or by equivalent converter switching. A second block
304
converts the synchronization signal SYNC using a high frequency (e.g., greater than 100 MHz) clock (e.g. counter) to an equivalent angular value with respect to the repetition rate of the synchronization signal SYNC, where one repetition equals 360 degrees. A first block
302
represents a hardware or software implementation of a Park vector algorithm. A third block
306
represents a hardware or software implementation of a transformation algorithm which converts the Park vector signal (in stationary coordinates) to a complex vector (in a synchronous coordinates) with respect to the synchronization signal SYNC.
A fourth block
308
represents a hardware or software implementation for performing a d-q transformation. The d-q transformation includes a decomposition of the current Park vector with respect to the current voltage into in-phase and out-of-phase orthogonal components.
A first low pass filter
310
extracts the quadrature component C
2
from the imaginary portion of the decomposition, and a second low pass filter
312
extracts the in-phase component S
2
from real portion of the decomposition. Summing junctions
314
and
316
sum these components C
2
and S
2
with phase and amplitude reference signals P
REF
and A
REF
to produce error signals A
ERR
and P
ERR
.
The error signals P
ERR
and A
ERR
are supplied to phase and amplitude regulators
318
and
320
. An output of the phase regulator
318
is used to control the first switch
124
. Sign and magnitude of an output of the amplitude regulator
320
are used to control the second switch
130
.
The invention has been described above in connection with a dc-dc converter, and particularly in connection with a dc-dc converter including a down-chopper (buck-chopper). However, the invention is not so limited. The invention may also be applied to a dc-dc converter including an up-chopper (boost-chopper).
The invention may be used for dc-ac conversion, where the chopper down-chopper (buck-chopper). However, the invention is not so limited. The invention may also be applied to a dc-dc converter including an up-chopper (boost-chopper).
The invention may be used for dc-ac conversion, where the chopper is replaced by a three-phase inverter, and the duty cycle control is replaced by a PWM frequency commutation logic. The commutation logic generates commutation commands that cause the inverter to modulate the dc line current at the commutation frequency. The three-phase ac current may be supplied to a load such as permanent magnet motor, induction motor or switched reluctance motor.
The invention may also be used in combination with a switched reluctance machine that is used as a switched reluctance generator. The switched reluctance generator supplies ac current to an ac-dc converter. The ac-dc converter, in turn, supplies pulsed current on a dc line. An active filter removes the fundamental frequency component on the dc line. The active filter can reduce capacitor size at the output terminals of the dc line.
The common element in all of these embodiments is that the dc line current has a high frequency content at the commutation frequency and at the associated higher harmonics. The phase and amplitude controls just described are capable of filtering the fundamental frequency harmonic component since the fundamental frequency is an integer multiple of the PWM frequency.
The invention is not limited to an active filter having a single trap circuit. Additional trap circuits may be used to sink additional harmonic frequency components
Accordingly, the invention is not limited to the specific embodiments described above. Instead, the invention is construed according to the claims that follow.
Claims
- 1. Apparatus comprising:a converter having a dc line, the converter being operable to provide a pulsed current on the dc line at a chopping frequency; and an active filter including a trap circuit, coupled to the dc line, for trapping a fundamental of the pulsed current, and a control for controlling phase and amplitude of trap circuit current to cancel ripple current produced by the converter.
- 2. The apparatus of claim 1, wherein the active filter includes a tuned trap circuit, the trap circuit being tuned to about the fundamental frequency; a first switch, coupled between the dc line and the trap circuit, for creating at least one deadband to adjust phase of current flowing through the trap circuit; and a power dissipation element and second switch coupled to the tuned trap circuit for diverting at least some stored current in the tuned trap circuit to control amplitude of the current flowing through the tuned trap circuit.
- 3. The apparatus of claim 2, wherein the first switch has an H-bridge configuration.
- 4. The apparatus of claim 2, wherein the trap circuit is tuned to a frequency that is about 10% higher than an integer multiple of the chopping frequency.
- 5. The apparatus of claim 2, further comprising a controller for creating two independent, decoupled control signals that are orthogonal to each other with respect to a common synchronizing signal, a first of the two signals being used to control the first switch, a second of the two signals being used to control the second switch, whereby the first signal is used to control the phase and the second signal is used to control the amplitude.
- 6. The apparatus of claim 5, wherein the controller includes a synchronous demodulator for creating the two independent, decoupled control signals.
- 7. The apparatus of claim 5, wherein the controller includes a Park vector-based decoupler for creating the two independent, decoupled control signals.
- 8. The apparatus of claim 5, further comprising an EMI filter, coupled to the dc line, for attenuating at least some higher harmonic frequency components of the pulsed current on the dc line.
- 9. An active filter for a converter having a dc line, the converter being operable to provide a pulsed current on the dc line at a chopping frequency, the active filter comprising:a trap circuit tuned to a fundamental frequency component of the pulsed current; a first switch, coupled between the dc line and the trap circuit; a controller for causing the first switch to create at least one deadband to adjust phase of current flowing through the trap circuit; and a power dissipation element and second switch coupled to the trap circuit, the controller controlling the second switch to divert at least some stored current in the trap circuit, the current being diverted to control amplitude of the current flowing through the tuned trap circuit; whereby the active filter is operable to trap the fundamental component of the pulsed current.
- 10. The filter of claim 9, wherein the first switch has an H-bridge configuration.
- 11. The filter of claim 9, wherein the trap circuit is tuned to a frequency that is about 10% higher than an integer multiple of the chopping frequency.
- 12. The filter of claim 9, further wherein the controller creates two independent, decoupled control signals that are orthogonal to each other with respect to a common synchronizing signal, a first of the two signals being used to control the first switch, a second of the two signals being used to control the second switch, whereby the first signal is used to control the phase and the second signal is used to control the amplitude.
- 13. The filter of claim 12, wherein the controller includes a synchronous demodulator for creating the two independent, decoupled control signals.
- 14. The filter of claim 12, wherein the controller includes a Park vector-based decoupler for creating the two independent, decoupled control signals.
- 15. A method of filtering pulsed current on a dc line, the method comprising:trapping a fundamental component of the switched current; using a trap circuit to create a resonant frequency and at least one deadband in the trapped current, the deadband being controlled to control phase of the trapped current; and diverting at least some of the trapped current to control amplitude of the switched current; whereby phase and amplitude of the trapped current can be made to cancel the fundamental of the switched current.
- 16. The method of claim 15, wherein the trap circuit includes a charge storage device, and wherein the method includes the steps of connecting and disconnecting the trap circuit to the dc line to control the phase; and connecting and disconnecting a power dissipation element across the charge storage device to control the amplitude; whereby the capacitor is discharged through the power dissipation element when the power dissipation element is connected across the charge storage device.
- 17. The filter of claim 16, further comprising the step of creating two independent, decoupled control signals that are orthogonal to each other with respect to a common synchronizing signal, a first of the two signals being used to determine when the trap circuit is connected to the dc line, a second of the two signals being used to determine when the power dissipation element is connected across the charge storage device.
- 18. The filter of claim 17, wherein synchronous demodulation is used to create the two independent, decoupled control signals.
- 19. The method of claim 17, wherein Park vector-based decoupling is used to create the two independent, decoupled control signals.
- 20. The method of claim 15, further comprising the step of using an EMI filter to attenuate at least some higher harmonic frequency components of the pulsed current on the dc line.
US Referenced Citations (8)