AM modulated wave eliminating circuit

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to an AM modulated wave eliminating circuit which extracts a digital modulated wave by eliminating an AM modulated wave from an AM data multiplex modulated wave consisting of an AM stereo modulated wave multiplexed with a digital modulated wave.

2. Related Background Art

Japanese Laid-Open Patent Application No. 9-326836 discloses an AM data multiplex modulation system which multiplexes an AM modulated wave with a digital signal so as not to influence an AM synchronous detection wave output at AM synchronous detection.

The AM data multiplex modulation system mentioned above will be described with reference to

FIG. 6

taking an example wherein modulation system uses the QPSK (four-phase PSK) modulation mode for a digital wave.

An analog signal wave such as a voice signal (hereinafter also referred to simply as a signal) is supplied to an AM modulator

31

and a carrier wave having a frequency fc is subjected to AM modulation with the signal wave. An AM modulated wave VAMt) output from the AM modulator

31

is expressed by the following Equation (1):

νAM(t)={1+κνm(t)}cos ωct (1)

wherein an amplitude of the carrier wave is taken as 1, the reference symbol ωc(rad/s) represents an angular frequency of the carrier wave, the reference symbol κ designates a modulation degree and a reference symbol νm(t) denotes the signal wave.

Arrays of digital signals of I and Q which are generated by a QPSK baseband digital signal generator

32

will be represented as In and Qn. Let us assume that In=±1 and Qn=±1.

The output signal from the QPSK baseband digital signal generator

32

is split into two, one being input into a quadrature modulator

33

to which a carrier wave having a frequency (fc+fα is supplied for DC-AC modulation of a carrier wave having an angular frequency (ωc+ωα)(rad/s) with a complex signal array. An output signal νDH(t) from the quadrature modulator

33

is as expressed by the following Equation (2):

νDH(t)=In cos(ωc+ωαt+Qn sin(ωc+ωα)t (2)

On the other hand, the QPSK baseband digital signals which are output from the QPSK baseband digital signal generator

32

are supplied to a sign inverter

34

for conversion into (−In) and (−Qn). The QPSK baseband digital signals which are subjected to the signal inversion by the sign inverter

34

are supplied to a complex conjugater

35

and made conjugate in complex, whereby the sign of the Qn signal array of the QPSK baseband digital signals is inverted, whereby the signal arrays are converted into (−In) and (Qn). That is, the sign inverter

34

and the complex conjugater

35

invert a sign of components which have a vector deviation corresponding to dibits formed by the QPSK baseband digital signals and the same phase as that of a standard carrier wave.

The complex signal arrays which are made conjugate in complex by the complex conjugater

35

are input into a quadrature modulator

36

to which the carrier wave having the frequency (fc−fα) is supplied and the carrier wave having the angular frequency (fc−fα) (rad/s) is subjected to quadrature conversion with the complex signal array. An output signal νDL(t) from the quadrature modulator

36

is as expressed by the following Equation (3):

νDL(t)=−In cos(ωc−ωα)t+Qn sin(ωc−ωα)t (3)

The output signals νDH(t) and νDL(t) expressed by the Equations (2) and (3) are added to each other with an adder

37

, which provides an addition output of νD(t) expressed by the following Equation (4):

νD(t)=νDH(t)+νDL(t)

=In cos(ωc+ωα)t+Qn sin(ωc+ωα)t

−In cos(ωc−ωα)t+Qn sin(ωc−ωα)t (4)

The AM modulated wave νAM(t) and the digital modulated wave νD(t) are input for addition into an adder

38

, which transforms the modulated waves expressed by the Equations (1) and (4) into an AM data multiplex modulated wave ν(t) expressed by the following Equation (5):

ν(t)=νAM(t)+νD(t)

={1+κνm(t)}cos ωct

+In cos(ωc+ωα)t+Qn sin(ωc+ωα)t

−In cos(ωc−ωα)t+Qn sin(ω−ωα)t (5)

A process to prepare the AM data multiplex modulation wave in the AM data multiplex wave modulation system is shown in

FIG. 7

, wherein the AM modulated wave output from the AM modulator

31

is indicated by a, the output signal from the quadrature modulator

36

, that is, the digital modulated wave, is indicated by b and the output signal from the quadrature modulator

33

, that is, the digital modulated wave is indicated by c. The digital modulated wave output from the adder

37

is a sum of the signals indicated by b and c in

FIG. 7

, and the AM multiplex modulated wave output from the adder

38

is indicated by d in FIG.

7

.

The AM data multiplex wave modulation system does not influence an AM synchronous detection wave output at an AM synchronous detection of the AM data multiplex modulated wave since the digital modulated wave signals are multiplexed at a location of the frequency (fc+fα) and a location of the frequency (fc−fα) which are axially symmetrical with regard to the carrier wave fc on a frequency axis.

However, the AM data multiplex modulation system may adopt the AM stereo modulation mode in place of the digital modulation mode for the AM data multiplex modulation. When the digital modulation mode is replaced with the AM stereo modulator, the AM data multiplex modulation system is incapable of extracting a digital modulated wave since a phase modulated wave and a digital modulated wave have similar characteristics in an AM stereo modulated wave.

Furthermore, the modulation system described above does not permit taking out desired digital data at an optionally selected time at which data are multiplexed or at an optionally selected frequency band at which data are multiplexed since it multiplexes, at the same frequency band and at the same time, an AM modulated component and a data modulated component of an AM data multiplex modulated wave which is multiplexed with a digital modulated wave and modulated in the AM data multiplex modulation mode though the system modulates an amplitude of a carrier wave having a frequency fc by an analog signal wave with an AM modulator, and multiplexes the digital modulated signals at the location of the frequency (fc+fα) and the location of the frequency (fc−fα) which are axially symmetrical with regard to the carrier wave having the frequency fc on the frequency axis.

A primary object of the present invention is to provide an AM modulated wave eliminating circuit which is capable of extracting a digital modulated wave by cancelling an AM modulated wave from an AM data multiplex modulated wave.

The AM modulated wave eliminating circuit according to the present invention extracts a digital modulated wave by cancelling an AM modulated wave from an AM data multiplex modulated wave which is multiplexed with an AM stereo modulated wave and a digital modulated wave within the same frequency band of an AM stereo modulated wave, the AM modulated wave eliminating circuit comprising extracting means to extract a composite wave composed of an AM carrier wave in-phase signal, an AM carrier wave reverse-phase signal and a digital modulated wave from an AM data multiplex modulated wave, presuming means to presume a value on the basis of a phase modulated component of an AM stereo modulated wave from the composite wave, and operating means to multiply a presumed value on the basis of the phase modulated component by the AM carrier wave in-phase component and add the multiplication output to the composite wave, characterized in that an output of the operating means provides a digital modulated wave.

The AM modulated wave eliminating circuit according to the present invention extracts the composite wave composed of the AM carrier wave in-phase signal, the AM carrier wave reverse-phase signal and the digital modulated wave from the AM data multiplex modulated wave by the extracting means, presumes the value from the extracted composite wave on the basis of the phase modulated component of the AM stereo modulated wave by the presuming means, multiplies a presumed value on the basis of the phase modulated component of the AM stereo modulated wave by AM carrier wave in-phase component and adds the multiplication output to the composite wave, thereby providing a digital modulated wave.

Furthermore, the AM modulated wave eliminating circuit according to the present invention performs AM modulation of a carrier wave having a frequency fc with an analog signal wave, and eliminates an AM modulated wave from an AM data multiplex modulated wave composed of digital modulated wave signals multiplexed at a location of a frequency (fc+fα and a location of a frequency (fc−fα) which are axially symmetrical with regard to a frequency axis of the carrier wave having a frequency fc within a frequency band of the AM modulated wave, characterized in that it comprises A/D converter means which samples the AM data multiplex modulated wave with sampling pulses having a frequency four times as high as that of the carrier wave and performs A/D conversion of sampled signals, a sampling point exchanging circuit which exchanges a discrete value output subjected to the A/D conversion at a sampling point (4m:m=0, 1, 2, 3 . . . ) with a discrete value output subjected to the A/D conversion at a sampling point (4m+3) and exchanges a discrete value output subjected to the A/D conversion at a sampling point (4m+1) with a discrete value output subjected to the A/D conversion at a sampling point (4m+2), and an adder which adds the discrete value output exchanged by the sampling point exchanging circuit to a discrete value output subjected to the A/D conversion by the A/D converter means.

In the AM modulated wave eliminating circuit according to the present invention, the AM data multiplex modulated wave is sampled at the frequency four times as high as the frequency of the carrier wave and the sampled signals are subjected to the A/D conversion by the A/D converter means. When m=0, 1, 2, 3, . . . , the discrete value output subjected to the A/D conversion at the sampling point (4m) is exchanged with the discrete value output subjected to the A/D conversion at the sampling point (4m+3) and the discrete value output subjected to the A/D conversion at the sampling point (4m+1) is exchanged with the discrete value output subjected to the A/D conversion at the sampling point (4m+2), and the exchanged discrete value output is added to the discrete value output subjected to the A/D conversion by the A/D converter means. As a result, only a digital modulated wave is sampled from the addition output with the sampling pulse and provided as a discrete output value subjected to the A/D conversion, whereby an AM modulated wave is eliminated.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1

is a block diagram illustrating a configuration of a first embodiment of the AM modulated wave eliminating circuit according to the present invention;

FIG. 2

is a schematic diagram descriptive of an assumption of a digital modulated signal selecting phase in the first embodiment of the AM modulated wave eliminating circuit according to the present invention;

FIG. 3

is a schematic diagram descriptive of another example of the assumption of a digital modulated signal selecting phase in the first embodiment of the AM modulated wave eliminating circuit according to the present invention;

FIG. 4

is a block diagram illustrating a configuration of a second embodiment of the AM modulated wave eliminating circuit according to the present invention;

FIGS. 5A through 5I

are schematic diagrams descriptive of functions of the second embodiment of the AM modulated wave eliminating circuit according to the present invention;

FIG. 6

is a block diagram illustrating a configuration of an AM data multiplex modulation system; and

FIG. 7

is a schematic diagram descriptive of multiplexing in the AM data multiplex modulation system.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Now, the AM modulated wave eliminating circuit according to the present invention will be described with reference to the preferred embodiments thereof.

FIG. 1

is a block diagram illustrating a configuration of a first embodiment of the AM modulated wave eliminating circuit according to the present invention.

An input AM data multiplex modulated wave is supplied to a multiplier

1

and multiplied by an oscillation output of a local oscillator

3

having an angular frequency of ωc/2 (rad/s), and a multiplication output is supplied to a low pass filter

5

, which eliminates high-pass components from the multiplication output. Similarly, the input AM data multiplex modulated wave is supplied to a multiplier

2

and multiplied by an oscillation output of a local oscillator

4

having an angular frequency of ωc/2 (rad/s), and a multiplication output is supplied to a low pass filter

6

, which eliminates high-pass components from the multiplication output.

Each of the AM data multiplex modulated waves which are supplied to the multipliers

1

and

2

has been subjected to AM stereo modulation of the Motroller-type AM stereo modulation and is expressed by the following Equation (6):

ν(t)={1+κνm(t)}cos(ωct+θ)

+In cos(ωc+ωα)t+Qn sin(ωc+ωα)t

−In cos(ωc−ωα)t+Qn sin(ωc−ωα)t (6)

θ=arc tan{νs(t)/(1+κνm(t))}

The first term of the Equation (6) represents an AM stereo modulated wave, and the second and subsequent terms designate a digital modulated wave. In the Equation (6), the reference symbol κ denotes a degree of AM modulation, the reference symbol νm(t) denotes a monaural modulated wave, the reference symbol νs(t) denotes a stereo modulated wave of the AM stereo modulated wave, the reference symbol ωc denotes an angular frequency (rad/s) of an AM modulated carrier wave, the reference symbols In and Qn denote I and Q complex signal arrays of the QSPK baseband signal, and the reference symbol (a denotes an angular frequency (rad/s) of a difference between a digital modulated carrier wave and an AM modulated carrier wave. The Equation (6) can be transformed into the following Equation (6a):

ν(t)={1+κνm(t)}(cos ωct·cos θ−sin ωct·sin θ)

−In sin ωct·sin ωαt+Qn sin ωct·cos ωαt

=[{1+κνm(t)}cos θ]cos ωct

+[−{1+κνm(t)}sin θ−In sin ωct+Qn cos ωαt]

×sin ωct (6a)

An AM multiplex modulated wave which has not been subjected to the AM stereo modulation is expressed by the Equation (5), which can be transformed into the following Equation (5a):

ν(t)={1+κνm(t)}cos ωct

−In sin ωct·sin ωαt+Qn sin ωct·cos ωαt

={1+κνm(t)}cos ωct

+(−In sin ωct+Qn cos ωαt)·sin ωct (5a)

By comparing the Equation (6a) with the Equation (5a), it will be understood that the modulated waves expressed by these Equations are composed of AM stereo modulated waves and digital modulated waves which have a center frequency fc (Hz) and perpendicularly intersect with each other. However, the AM modulated wave intersects with the digital modulated wave completely perpendicularly in case of the AM multiplex modulated wave expressed by the Equation (5a), whereas the AM stereo phase modulated wave is overlapped with the digital modulated wave in case of the AM data multiplex modulated wave subjected to the AM stereo modulation, that is, the wave expressed by the Equation (6a). Accordingly, the presence of the AM stereo phase modulated wave makes it difficult to take out only the digital modulated wave as already described above.

Speaking again of the AM modulated wave eliminating circuit preferred as the first embodiment of the present invention, the AM data multiplex modulated wave expressed by the above-mentioned Equation (6) is multiplied in the multiplier

1

by the oscillation output cos({fraction (3/2)}) ωct from the local oscillator

3

, the high-pass frequency components are eliminated from a multiplication result and the low pass filter

5

sends out a signal expressed by the following Equation (7):

2{ν(t)cos({fraction (3/2)})ωct}

=(1+M)cos{(−½)ωct+θ}

+In cos{(−½)ωc+ωα}t+Qn sin{(−½)ωc+ωα}t

−In cos{(−½)ωc−ωα}t+Qn sin{(−½)ωc−ωα}t

=(1+M)cos{(½)ωct−θ}

−In cos{(½)ωc+ωα}t−Qn sin{(½)ωc+ωα}t

+In cos{(½)ωc−ωα}t−Qn sin{(½)ωc−ωα}t (7)

wherein M=κνm(t).

Similarly, the AM data multiplex modulated wave expressed by the Equation (6) shown above is multiplied in the multiplier

2

by an oscillation output cos(½)ωct from a local oscillator

4

, high-pass frequency components are eliminated from a multiplication result and the low pass filter

6

sends out a signal which is expressed by the following Equation (8):

2{ν(t)cos(½)ωct}

=(1+M)cos{(½)ωct+θ}

+In cos{(½)ωc+ωα}t+Qn sin{(½)ωc+ωα}t

−In cos{(½)ωc−ωα}t+Qn sin{(½)ωc−ωα}t (8)

An output from the low pass filter

5

and an output from the low pass filter

6

are supplied to an adder

7

for addition. The adder

7

provides an addition output expressed by an Equation (9) shown below. Furthermore, the outputs from the low pass filters

5

and

6

are supplied to an subtracter

8

, which subtracts the output of the low pass filter

6

from the output of the low pass filter

5

. The subtracter

8

provides a subtraction output expressed by an Equation (10) shown below:

(1+M)cos{(½)ωct−θ)}+(1+M)cos{(½)ωct+θ)}

=2(1+M)cos(½)ωct·cos θ (9)

(1+M)cos{(½)ωct−θ)}−(1+M)cos{(½)ωct+θ)}

2In cos{(½)ωc+ωα}t−2Qn sin{(½)ωc+ωα}t

+2In cos{(½)ωc−ωα}t−2Qn sin{(½)ωc−ωα}t

=−2(1+M)sin(½)ωct·sin θ

−2In cos{(½)ωc+ωα}t−2Qn sin{(½)ωc+ωα}t

+2In cos{(½)ωc−ωα}t−2Qn sin{(½)ωc−ωα}t (10)

wherein θ=0 at the monaural modulation, whereby the first term of the Equation (10) disappears and only the digital modulated wave remains.

An output from the adder

7

is supplied to a multiplier

9

, which multiplies the output by an oscillation output of a local oscillator

11

having an angular frequency ωc/2 (rad/s) and a multiplication output is supplied to a low pass filter

13

, which eliminates high-pass frequency components from the multiplication output. Similarly, an output from the subtracter

8

is supplied to a multiplier

10

, which multiplies the output by an oscillation output from a local oscillator

12

having an angular frequency ωc/2 (rad/s) and a phase π/2 (rad) delayed from the oscillation output of the local oscillator

11

and a multiplication output is supplied to a low pass filter

14

, which eliminates high-pass frequency components from the multiplication output.

That is, the output from the adder

7

is multiplied by an output cos(½)ωct from the local oscillator

11

and the low pass filter

13

provides an output which is expressed by an Equation (11) shown below. The output from the low pass filter

13

is supplied to a multiplier

15

and multiplied by an oscillation output from a local oscillator

17

having an angular frequency ωc/2 (rad/s). The multiplier

15

provides an output which is expressed by an Equation (12) which is shown below:

2(1+M)cos(½)ωct·cos θ·cos(½)ωct

=(1+M)cos θ·cos θ

=(1+M)cos θ (11)

(1+M)cos(½)ωct·cos θ (12)

As apparent from the Equation (12), an AM carrier wave in-phase signal is obtained from the multiplier

15

.

On the other hand, the output from the subtracter

8

is multiplied by an output sin(½) ωct from the local oscillator

12

, whereby the low pass filter

14

provides an output expressed by an Equation (13) shown below. The output from the low pass filter

14

is supplied to a multiplier

16

and multiplied by an oscillation output from a local oscillator

18

having an angular frequency ωc/2 (rad/s). The multiplier

16

provides an output expressed by an Equation (14) shown below:

[−2(1+M)sin(½)ωct·sin θ

−2In cos{(½)ωc+ωα}t−2Qn sin{(½)ωc+ωα}t

+2In cos{(½)ωc−ωα}t−2Qn sin{(½)ωc−ωα}t]

×sin(½)ωct

=−(1+M)cos θ·sin θ

+In sin ωαt−Qn cos ωαt−In sin(−ωα)t−Qn cos(−ωα)t

=−(1+M)sin θ+2In sin ωαt−2Qn cos ωαt (13)

νamq(t)={−(1+M)sin θ+2In sin ωαt−2Qn cos ωαt}

×cos(½)ωct

=−(1+M)cos(½)ωct·sin θ+2In cos(½)ωct·sin ωαt

−2Qn cos(½)ωct·cos ωαt

=−(1+M)cos(½)ωct·sin θ+In sin{(½)ωc+ωα}t

−Qn cos{(½)ωc+ωα}t−In sin{(½)ωc−ωα}t

−Qn cos{(½)ωc−ωα}t (14)

As apparent from the formula (14) which represents an AM carrier wave reverse-phase signal by the first term and a digital modulated wave by the second and subsequent terms, the multiplier

16

provides a signal νamq, that is, “AM carrier wave reverse-phase signal+digital modulated signal”.

Then, high-pass frequency components are eliminated by supplying the output from the multiplier

15

, that is, the AM carrier wave in-phase signal, to a low pass filter

19

which is set at a cut-off frequency fc(=ωc/(2π)) and feeding the output from the multiplier

16

, that is, the νamq(t) reverse-phase signal, to a low pass filter

20

which is set at the cut-off frequency fc (=ωc/(2π)). By passing the output through the low pass filter

20

, the term {(½)ωc+ωα}t is eliminated from the Equation (14). In the Equations (12) and (14), variation rates of (1+M), cos θ, sin θ, In and Qn are extremely lower than those of cos(½)ωct and cos{(½)ωc−ωα}t. Therefore, values of former factors can be regarded as constants for a time on the order of a period of cos{(½)ωc−ωα}t.

On the other hand, a reference numeral

21

denotes a phase level generator which outputs a signal of constant ki=sin θi/sin θi (1≦i≦n, −(π/4)≦θ≦(4/π)). (1≦i≦n, −(π/4≦θ≦4/π)) was adopted since the Motroller system adopts {−(π/4)≦θ≦(4/π)} as a maximum phase deviation angle of AM stereo.

The outputs ki from the phase level generator

21

are multiplied by outputs from the low pass filter

19

independently in multipliers

221

,

222

,

223

and

22

n

. Each of the multipliers

221

,

222

,

223

,

22

n

provides an output which is expressed by the following Equation (12a):

(1+M)cos(½)ωct·cos θ·sin θi/cos θi (12a)

Adders

231

,

232

,

233

and

23

n

add the outputs from the multipliers

221

,

222

,

223

and

22

n

to an output from a low pass filter

20

which is expressed by the following Equation (14a):

−(1+M)cos(½)ωct·sin θ−In sin{(½)ωc−ωα}t

−Qn cos{(½)ωc−ωα}t (14a)

In

FIG. 2

, a indicates a waveform of the AM carrier wave in-phase signal having passed through the filter

19

for a period T of cos{(½)ωc−ωα}t, whereas b

1

, b

2

, b

3

and bn in

FIG. 2

indicate waveforms of the outputs from the mulitipliers

221

,

222

,

223

and

22

n

for a period T of cos {(½)ωc−ωα}t which are products of the waveform indicated by a in

FIG. 2

multiplied by ki. A wave form output from the low pass filter

20

which is expressed by the Equation (14a), or a composite waveform consisting of the waveforms indicated by c

1

and c

2

in

FIG. 2

is added to the waveforms indicated by b

1

, b

2

, b

3

and bn in

FIG. 2

by the adders

231

,

232

,

233

and

23

n

respectively. Though the waveforms indicated by c

1

and c

2

in

FIG. 2

are actually composed into a single waveform, the waveforms are shown independently for convenience of description: c

1

indicating a waveform of the AM carrier wave reverse-phase signal and c

2

indicating a waveform of the digital modulated wave.

The adders

231

,

232

,

233

and

23

n

provide outputs having waveforms which are indicated by d

1

, d

2

, d

3

, dn and d

0

in FIG.

2

. Though the waveforms of the outputs from the adders

231

,

232

,

233

and

23

n

are actually composite waveforms which are obtained by synthesizing the waveforms indicated by d

1

, d

2

, d

3

and dn with that indicated by d

0

, the waveforms are shown independently for convenience of description.

The addition outputs from the adders

231

,

232

,

233

and

23

n

are supplied to a phase presumer

24

, which integrates the input waveforms for a period of cos{(½)ωc−ωα}t. Since this integration makes it possible to approximate variations of In and Qn for the time to “0”, the digital modulated wave is cancelled, whereby the phase presumer

24

finds and outputs a point at which a signal having the cancelled digital modulated wave gives θ=θi as a point at which an input has a minimum value out of the inputs in a number of n. The minimum value of the input is apparent from a fact that [a value of the Equation (12a)−a value of the Equation (14a) in which the digital modulated wave is cancelled], that is, (1+M)cos(½)ωct·sin θ−(1+M)cos(½)ωct·sin θ is “0” at θ=θi.

The output from the phase presumer

24

is supplied to a phase level selector

25

, which selects, out of outputs input into the phase level selector

25

from the phase level generator

21

, an optimum phase level, that is, a phase level which is expressed by the following Equation (15):

sin θ/cos θ (15)

An output from the phase level selector

25

and the AM carrier wave in-phase signal are supplied to a multiplier

26

and multiplied by each other. As a result, the multiplier

26

outputs a signal expressed by the following Equation (16):

(1+M)cos(½)ωct·cos θ·sin θ/cos θ

=(1+M)cos(½)ωct·sin θ (16)

The output from the multiplier

26

and (AM carrier wave reverse-phase signal+digital modulated wave) are input into an adder

27

. As a result, the adder

27

outputs a signal expressed by the following Equation (17):

(1+M)cos(½)ωct·sin θ+νamq(t)

=(1+M)cos(½)ωct·sin θ−(1+M)cos(½)ωct·sin θ+2In cos(½)ωct·sin ωα

t−2Qn cos(½)ωct·cos ωαt=2In cos(½)ωct·sin ωαt−2Qn cos(½)ωct·cos ωαt

=+2In cos(½)ωct·sin ωαt−2Qn cos(½)ωct·cos ωαt

=2In cos(½)ωct·sin ωαt−2Qn cos(½)ωct·cos ωαt (17)

Accordingly, the AM stereo modulated wave is cancelled in the output from the adder

27

, whereby the adder

27

outputs the digital modulated wave.

Now, description will be made of a modification of the first embodiment of the AM modulated wave eliminating circuit according to the present invention.

The first embodiment of the present invention adopts the method to find the minimum value by cancelling the digital modulated wave. This method may be replaced with a method to search for a waveform containing a wave which is most likely to be a digital modulated wave during the presumption of a phase.

In

FIG. 3

, e through h indicate waveforms exemplifying waveforms which are actually input into the phase presumer

24

on the basis of the waveforms indicated by d

1

through d

0

in FIG.

2

. Out of the waveforms indicated by e through h in

FIG. 3

, those indicated by e, f and h are sinusoidal waves which do not complete within a period T of cos{(½)ωc−ωα}t, whereas that indicated by g in

FIG. 3

is a clear sinusoidal wave. This sinusoidal wave signifies that only a digital modulated wave remains in the waveform and Ki of this waveform is a phase of an AM stereo to be determined.

Though ({fraction (3/2)})ωc is taken as the oscillation angular frequency of the local oscillator

3

and (½)ωc is taken as the angular frequency of the local oscillators

4

,

11

,

12

,

17

and

18

in the description made above, it is possible to adopt different angular frequencies. When the oscillation angular frequency of ({fraction (3/2)})ωc is represented by ωpf1, the angular frequency of (½)ωc is designated by ωpf2 and ωpf1 is higher than ωpf2, it is sufficient to select a relationship which satisfies ωpf1−ωc=ωc−ωpf2.

As understood from the foregoing description, the AM modulated wave eliminating circuit according to the present invention provides an effect to enable to extract a digital modulated wave by eliminating an AM modulated wave even in a case of an AM stereo modulation, thereby making it possible to demodulate digital modulated waves which could not conventionally be demodulated.

FIG. 4

is a block diagram illustrating a configuration of a second embodiment of the AM modulated wave eliminating circuit according to the present invention.

In the AM modulated wave eliminating circuit preferred as the second embodiment of the present invention, an input AM data multiplex modulated wave is supplied to a carrier reproducer

41

, which reproduces an AM modulated carrier wave, and the AM modulated carrier wave reproduced by the carrier reproducer

41

is supplied to a timing signal generator

42

, which generates a timing signal for A/D conversion. A timing signal which has a frequency four times as high as that of a carrier wave is generated as a sampling pulse. On the other hand, the input AM data multiplex modulated wave is supplied to a delay device

43

, which delays this wave and an output from the delay device

43

is supplied to an A/D converter

44

, which performs A/D conversion at a timing on the basis of the timing signal. An A/D conversion output provided from the A/D converter

43

is supplied, for exchange with another A/D conversion output, to a sampling point exchanging circuit

45

which exchanges an A/D conversion output subjected to the A/D conversion at a sampling point (4m : m=0, 1, 2, 3, . . . ) (a discrete value output) with an A/D conversion output subjected to the A/D conversion at a sampling point (4m+3) and exchanges an A/D conversion output subjected to the A/D conversion at a sampling point (4m+1) with an A/D conversion output subjected to the A/D conversion at a sampling point (4m+2), and the A/D conversion output from the A/D converter

44

and an output from the sampling point exchanging circuit

45

are supplied to an adder

46

, which adds these outputs and sends out an addition output.

The AM data multiplex modulated wave ν(t) expressed by the Equation (5) mentioned above can be transformed as expressed by the following Equation (6):

ν(t)={1+κνm(t)}cos ωct

+In cos(ωc+ωα)t+Qn sin(ωc+ωα)t

−In cos(ωc−ωα)t+Qn sin(ωc−ωα)t

={1+κνm(t)}cos ωct

−2In sin ωct sin ωαt+2Qn cos ωαt sin ωct (6)

The reference symbol ωα denotes an angular frequency (rad/s) of a difference between a digital modulated carrier wave and the AM modulated carrier wave, and the digital modulated carrier wave exists in upper and lower side bands of the AM modulated carrier wave at locations which are symmetrical with each other and apart from each other for a frequency fα. Furthermore, let us assume fc>>fα.

In the AM modulated wave eliminating circuit preferred as the second embodiment of the present invention shown in

FIG. 4

, an AM data multiplex modulated wave expressed by the Equation (6) mentioned above is input into the carrier reproducer

41

and the delay device

43

.

The input AM data multiplex modulated wave signal ν(t) is input into the carrier reproducer

41

and the delay device

43

. Description will be made first of carrier eave reproduction by the carrier reproducer

41

. The carrier reproducer

41

reproduces a carrier wave for AM modulation and provides an output νc(t) which is expressed by the following Equation (7):

νc(t)=cos ωct (7)

The signal νc(t) is input into the timing signal generator

42

, which generates a timing signal for A/D conversion. In the second embodiment, the timing signal is output when the signal νc(t) of a single frequency has phases of (π/4) radian, (3π/4) radian, (5π/4) radian and (7π/4) radian. In other words, the timing signal is generated so that the AM data multiplex modulated signal ν(t) is sampled by the A/D converter

44

at time t expressed by the following Equation (8):

t=1+2m/8fc (8)

wherein m=0, 1, 2, 3, . . . , whereby a sampling frequency is 4fc (Hz).

On the other hand, the delay device

43

sends out the AM data multiplex modulated wave ν(t) with a delay time which is a total sum of a delay time in the carrier reproducer

41

and that in the timing signal generator

42

. Let us not to take the delay time into consideration and take a delay time in the delay device

43

as 0 for simplicity. Accordingly, a signal output from the delay device

43

is equal to the AM data multiplex modulated wave ν(t), and is input into the A/D converter

44

and is sampled with the timing signal generated by the timing signal generator

42

.

Therefore, the AM data multiplex modulated wave signal ν(t) is sampled with the sampling pulse having a sampling frequency of 4fc (Hz) and subjected to A/D conversion by the A/D converter

44

.

Accordingly, a discrete output signal νD(m) which is provided from the A/D converter

44

is as expressed by the following Equation (9):

νD(m)={1+Kνm(t)}cos 2πfc(1+2m)/8fc

−In{sin 2πfc(1+2m)/8fc}{sin 2πfα(1+2m)/8fc}

+Qn{sin 2πfc(1+2m)/8fc}{cos 2πfα(1+2m)/8fc}

={1+Kνm(t)}cos(1+2m)π/4

−In sin(1+2m)π/4·sin(1+2m)πfα/4fc

+Qn sin(1+2m)π/4·cos(1+2m)πfα/4fc

={1+Kνm(t)}cos(1+2m)π/4

+[−In sin(1+2m)πfα/4fc

+Qn cos(1+2m)πfα/4fc]·sin(1+2m)π/4 (9)

Then, description will be made of exchanging processes in the sampling point exchanging circuit

45

. This circuit exchanges sampling points in a manner: m=0→←m=3, m=1→←m=2, m=4→←m=7, m=5→←m=6, . . . . The symbols →← indicate mutual exchange of sampling points. That is, the sampling point exchanging circuit

45

exchanges the sampling points with each other as indicated by (4m)→←(4m+3) and (4m+1)→←(4m+2) at m=0, 1, 2, 3, . . .

Since the condition of fc>>fα makes variations of a value of {1+Kνm(t)} and that of [−In sin(1+2m)πfα/4fc+Qn cos(1+2m)πfα/4fc] in the Equation (9) remarkably slower than those of a value of cos(1+2m)π/4 and a value of sin(1+2m)π/4, it is possible in the second embodiment that the values of the former terms remain unchanged regardless of the exchange of the sampling points.

In the Equation (9), cos(1+2m)π/4=1/{square root over (2)} and sin (1+2m)π/4=1/{square root over (2)} at m=0, 4, 8, . . . ,

cos(1+2m)π/4=1/{square root over (2)} and sin(1+2m)π/4=−1/{square root over (2)} at m=1, 5, 9, . . .

cos(1+2m)π/4=−1/{square root over (2)} and sin(1+2m)π/4=−1/{square root over (2)} at m=2, 6, 10, . . . , and

cos(1+2m)π/4=−1/{square root over (2)} and sin(1+2m)π/4=1/{square root over (2)} at m=3, 7, 11, . . .

Therefore, the exchange of the sampling points described above (m=0→←m=3, m=1→←m=2, m=4→←m=7, m=5→←m=6, . . . ) changes a sign only of cos(1+2m)π/4 but does not change a sign of sin(1+2m)π/4, whereby the sampling point exchanging circuit

45

provides an output expressed by the following Equation (10):

−{1+Kνm(t)}cos(1+2m)π/4

+[−In sin(1+2m)πfα/4fc

+Qn cos(1+2m)πfα/4fc]·sin(1+2m)π/4 (10)

The fact that the exchange of the sampling points does not cause changes of the values will be described with reference to schematic diagrams shown in

FIGS. 5A through 5I

.

{1+kνm(t)} varies as shown in

FIG. 5A

in which a scale in a direction of amplitude is larger than an actual scale and {−In sin(1+2m)πfα/4fc+Qn cos(1+2m)πfα/4fc] varies as shown in

FIG. 5B

in which a scale in the direction of amplitude is larger than the actual scale. In contrast, cos(1+2m)π/4 varies as shown in FIG.

5

C and sin(1+2m)π/4 varies as shown in FIG.

5

D.

FIG. 5E

shows sampling points which are numbered as 0, 1, 2, 3, . . . .

Corresponding to the variations described above, sampling points for the AM data multiplex modulated wave signal ν(t) are exchanged with each other as indicated by →← in FIG.

5

E. Since {1+Kνm(t)} varies little as shown in

FIG. 5A

, its value remains substantially unchanged as shown in

FIG. 5F

regardless of the exchange of the sampling points. Furthermore, since [−In sin(1+2m)πfα/4fc+Qn cos(1+2m)πfα/4fc] varies little as shown in

FIG. 5B

, its value remains substantially unchanged as shown in

FIG. 5G

regardless of the exchange of the sampling points.

In contrast, −cos(1+2m)π/4 varies as shown in FIG.

5

H and sin(1+2m)π/4 varies as shown in FIG.

5

I. The sign of cos(1+2m)π/4 has been made negative as already described above.

Then, the output from the A/D converter

44

expressed by the Equation (8) and the output from the sampling point exchanging circuit

45

expressed by the Equation (10) are added by an adder

46

, which provides an output expressed by the following Equation (11):

2[−In sin(1+2m)πfα/4fc

+Qn cos(1+2m)πfα/4fc]·sin(1+2m)π/4 (11)

The output from the adder

46

expressed by the Equation (11) is a signal consisting only of a digital modulated wave which is sampled at the sampling frequency of 4fc (Hz). Accordingly, the adder

46

outputs a signal prepared by eliminating the AM modulated wave from the AM data multiplex modulated wave and the baseband digital data can be demodulated from this signal.

As understood from the foregoing description, the AM modulated wave eliminating circuit according to the present invention is capable of eliminating an AM modulated wave from an AM data multiplex modulated wave with a simple and small configuration.

FIG. 1

21

Phase level generator

24

Phase presumer

25

Phase level selector

FIG. 2

a AM carrier wave in-phase signal

c

1

AM carrier wave reverse-phase signal

c

2

Digital modulated wave

d

0

Digital modulated wave

FIG. 4

41

Carrier regenerator

42

Timing signal generator

43

Delay device

45

Sampling point exchanging circuit

FIG. 6

31

AM modulator

32

QPSK baseband signal generator

33

D/A modulator

34

Sign inverter

35

Complex conjugater

36

D/A modulator

#

1

Signal

#

2

AM modulator

#

3

AM data multiplex modulated wave

#

4

Digital modulated wave

FIG. 7

#

1

AM modulated wave

#

2

Digital modulated wave

#

3

AM data multiplex modulated wave

Number	Date	Country	Kind
10-060392	Feb 1998	JP
10-151890	May 1998	JP

Number	Name	Date
5633869	Carlin et al.	May 1997
5878089	Dapper et al.	Mar 1999
6128334	Dapper et al.	Oct 2000

AM modulated wave eliminating circuit

Information

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Date Filed

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Inventors

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CPC

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Abstract

Description

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Priority Claims (2)

US Referenced Citations (3)