CROSS-REFERENCE TO RELATED APPLICATIONS
Not applicable.
STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT
Not applicable.
NAMES OF THE PARTIES TO A JOINT RESEARCH AGREEMENT
Not applicable.
INCORPORATION-BY-REFERENCE OF MATERIALS SUBMITTED ON A COMPACT DISC
Not applicable.
BACKGROUND OF THE INVENTION
1. Field of the Invention
The present disclosure relates to an amplifier circuit and a controlling method thereof.
2. Description of Related Art Including Information Disclosed Under 37 CFR 1.97 and 37 CFR 1.98.
With the development of the wireless communication system developing from the second generation (2G) wireless network, to the third generation (3G) wireless network, to the new fourth generation (4G) wireless network, a big challenge that portable system designers encounter is to design a system supporting numerous wireless communication standards including the global system for mobile communications (GPRS), general packet radio service (GPRS), enhanced data rates for global evolution (EDGE), code division multiple access (CDMA), wideband CDMA (WCDMA) and wireless fidelity (Wi-Fi), such as IEEE 802.11 and the related standards. Also, the communication systems using a 4G topology require capability to operate in multiple frequencies and multiple modes and support of a multiple-input multiple-output (MIMO) antenna structure. However, the power source of most portable systems is a battery. Therefore, with the trend toward design of more complicated circuits and increasing number of devices, integrating modules in a system to a single chip and increasing battery lifetime are becoming important issues in portable system design.
In a portable wireless communication system, a baseband processor and a radio frequency receiver can now be integrated into a single chip. However, a radio frequency (RF) power amplifier (PA), which consumes the most power in the chip, is implemented by gallium-arsenide (GaAs) technology, and thus it cannot be integrated in the single chip. The known semi-insulative property of the GaAs material can largely reduce parasitic resistance in the base, making the GaAs material well-suited for the application of high frequency circuits. However, GaAs technology exhibits relatively low yields and high costs in manufacturing. In addition, such technology cannot implement a system-on-chip structure. Therefore, with the recent improvement of the CMOS technology, more and more chip designers are attempting to design RF power amplifiers by the CMOS technology.
The considerations in the design of an RF power amplifier in general are the linearity and the power efficiency. In general, the power efficiency is maximum when the RF power amplifier operates in or near the saturation region. However, in order to modulate an RF signal having an inconsistent envelope, the RF power amplifier usually operates at several dB of back-off out power from 1 dB gain compression point or peak power so as to insure it works at the linear region. The RF power amplifier operated in the linear region has lower power efficiency, and thus suffers from reduced battery life of the portable system.
The factors causing the non-linearity of the power amplifier in general are the even-order harmonic and odd-order harmonic, wherein the even-order harmonic includes a second-order harmonic, which is usually the primary portion of the total harmonic distortion. FIG. 1 shows a block diagram of a conventional RF power amplifier 10 configured to suppress the second-order harmonic. Referring to FIG. 1, an input signal from an input terminal 11 is split through an input balun 12 into two signals S1 and S2, each having equal power and phase difference of 180 degrees with respect to each other. Signals S1 and S2 are transmitted to first and second input matching circuits 13 and 14, respectively, to obtain impedance matching, and then transmitted to first and second amplifiers 15 and 16, respectively. Next, the output signals of the first and second amplifiers 15 and 16 are transmitted to first and second output matching circuits 17 and 18, respectively, to obtain impedance matching, and transmitted to an output balun 19. The split signals S1 and S2 are combined in phase via the output balun 19 and the combined signal is outputted from an output terminal 20. Referring to FIG. 1, the conventional RF power amplifier 10 requires two baluns 12 and 19 to split and combine a signal.
US publication No. 2006/0049876 discloses an active circuit 22 having improved linearity using multiple gated transistors, wherein the active circuit 22 is a common gate circuit. Referring to FIG. 2, the active circuit 22 comprises a main circuit 24 and an assistant circuit 26. The main circuit 24 comprises a transistor M1 and a capacitor CA, and the assistant circuit 26 comprises a transistor M2 and a capacitor CB. The third-order harmonic of the active circuit 22 is suppressed via the assistant circuit 26; however, the second-order harmonic of the active circuit 22 requires an additional balun to be suppressed.
A conventional balun occupies a large chip area, and the power consumption of the balun reduces the overall efficiency of the chip. Therefore, it is desirable to provide an amplifier circuit having improved linearity and to provide a design for improving the power efficiency of the amplifying circuit and for increasing the battery life.
BRIEF SUMMARY OF THE INVENTION
According to one embodiment, providing an amplifier circuit comprises a first unit and a second unit. The first unit comprises a first amplifying unit, wherein the first amplifying unit comprises a first main circuit unit and a first assistant circuit unit, and the first assistant circuit unit is configured for assisting the linearity of the first main circuit unit. The second unit comprises a second amplifying unit, wherein the second amplifying unit comprises a second main circuit unit and a second assistant circuit unit, and the second assistant circuit unit is configured for assisting the linearity of the second main circuit unit. The first amplifying unit is configured for conducting in one half cycle of an input signal, and the second amplifying unit is configured for conducting in the other half cycle of the input signal.
According to one embodiment, providing a method for improving the linearity of an amplifier circuit comprises the steps of: receiving an input signal; conducting a first amplifying unit in one half cycle of the input signal and cutting off a second amplifying unit; cutting off the first amplifying unit in the other half cycle of the input signal and conducting the second amplifying unit; receiving a first control signal for adjusting the linearity of the first amplifying unit when the first amplifying unit conducts; and receiving a second control signal for adjusting the linearity of the second amplifying unit when the second amplifying unit conducts.
BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS
FIG. 1 shows a block diagram of a conventional RF power amplifier configured to suppress the second-order harmonic.
FIG. 2 shows a schematic view of an active circuit having improved linearity using multiple gated transistors.
FIG. 3 shows a block diagram of an amplifier circuit according to one embodiment.
FIG. 4 shows a block diagram of the first amplifying unit according to one embodiment.
FIG. 5 shows a block diagram of the second amplifying unit according to one embodiment.
FIG. 6 shows a graph illustrating first and second order derivatives (gm′ and gm″) of the trans-conductance of the NMOS transistor N1 with respect to voltage Vgs.
FIG. 7 shows a block diagram of the first amplifying unit according to another embodiment.
FIG. 8 shows a block diagram of the first amplifying unit according to yet another embodiment.
FIG. 9 shows a block diagram of the second amplifying unit according to yet another embodiment.
FIG. 10(A) shows a simulation result of the output power Pout, gain, power added efficiency (PAE) of the amplifier circuit according to yet another embodiment.
FIG. 10(B) shows a comparison result of the amplifier circuit with the prior art 1 and prior art 2 arrangements.
DETAILED DESCRIPTION OF THE INVENTION
Exemplary Embodiments will now be described more fully with reference to the accompanying drawings. The embodiments may, however, be embodied in many different forms and should not be construed as limited to the embodiments set forth herein. Rather, these embodiments are provided so that this disclosure will be thorough and complete, and will fully convey the scope of the embodiments to those skilled in the art.
FIG. 3 shows a block diagram of an amplifier circuit 30 according to one embodiment. The amplifier circuit 30 comprises a first unit 32 and a second unit 34. The first unit 32 is configured to conduct in one half cycle of an input signal, and the second unit 34 is configured to conduct in the other half cycle of the input signal. Referring to FIG. 3, the first unit 32 comprises a first input matching circuit 36, a first amplifying unit 38, and a first output matching circuit 40, and the second unit 34 comprises a second input matching circuit 37, a second amplifying unit 39, and a second output matching circuit 41. In one embodiment, the first and second amplifying units 38 and 39 are push-pull architectures.
Referring to FIG. 3, the first and second input matching circuits 36 and 37 are disposed between an input terminal and the first and second amplifying units 38 and 39, and thus the input impedances of the first and second amplifying units 38 and 39 can match impedance with the input terminal and avoid an input signal Vin being reflected to the point of origin, resulting in the loss of the input power. The first and second output matching circuits 40 and 41 are disposed between an output terminal and the first and second amplifying units 38 and 39, and thus the output impedances of the first and second amplifying units 38 and 39 can match impedance with the system load impedance of the output terminal. And the matching circuits 40 and 41 are provided with phase tuning function in order to coherence the output signal from amplifying units 38 and 39. In one embodiment, the matching circuits 36, 37, 40, and 41 comprise at least one lump device. In another embodiment, the matching circuits 36, 37, 40, and 41 comprise at least one transmission line, or comprise at least one balanced to unbalanced transformer.
FIG. 4 shows a block diagram of the first amplifying unit 38 according to one embodiment. The first amplifying unit 38 comprises AC coupling capacitors C1, C2, and C3, a main circuit unit 382, and an assistant circuit unit 384. Referring to FIG. 4, an input signal in1 is applied to the main circuit unit 382 and the assistant circuit unit 384 via the AC coupling capacitors C1 and C2, respectively. In one embodiment, the main circuit unit 382 is an NMOS transistor N1. The drain terminal of the NMOS transistor N1 is connected to a power source VDD and to an output terminal out1 via the AC coupling capacitors C3. The source terminal of the NMOS transistor N1 is connected to a reference voltage potential, and the gate terminal of the NMOS transistor N1 is connected to a bias voltage Vbias1.
As mentioned before, the first amplifying unit 38 conducts during the positive half-cycle of the input signal in1, while the second amplifying unit 39 cuts off. At this time, a current I1 flowing to the output terminal of the amplifier circuit 30 can be expressed as
wherein n represents the harmonic number of the combined current I0.
In contrast, the first amplifying unit 38 cuts off during the negative half-cycle of the input signal in1, while the second amplifying unit 39 conducts. At this time, a current I2 flowing to the output terminal of the amplifier circuit 30 can be expressed as
The even-order harmonic of I0 is suppressed after adding current I1 and I2. Therefore, the even-order harmonic of the amplifier circuit 30 can be suppressed via such architecture. The first and second amplifying units 38 and 39 do not turn on at the same time, and thus the direct power consumption of the chip can be reduced.
Referring to FIG. 4, the source terminal current of the NMOS transistor N1 and the gate terminal to the source terminal voltage (Vgs) can be expressed as
wherein gm represents a trans-conductance of the NMOS transistor N1, and gm′ and gm″ are first and second derivatives of the trans-conductance, respectively.
FIG. 6 shows a graph illustrating first and second order derivatives (gm′ and gm″) of the trans-conductance of the NMOS transistor N1 with respect to voltage Vgs. Referring to FIG. 6, gm″ has a positive value when Vgs is less than a bias point of the class B amplifier (about 0.83V in the figure), and has a negative value when Vgs is larger than the bias point of the class B amplifier. It is known in the art that gm″ affects a 3rd Inter-Modulation Distortion (IMD3).
Accordingly, the assistant circuit unit 384 is configured to assist the linearity of the main circuit unit 382. In one embodiment, the assistant circuit unit 384 is an NMOS transistor N2. The drain terminal of the NMOS transistor N2 is connected to the power source VDD and to the output terminal out 1 via the AC coupling capacitors C3. The source terminal of the NMOS transistor N2 is connected to the reference voltage potential, and the gate terminal of the NMOS transistor N2 is connected to a bias voltage Vth1. The bias voltage Vth1 is used to adjust the peak value of the second order derivative of the assistant circuit unit 384 so as to suppress the peak value of the second order derivative of the main circuit unit 382. In addition, the second order derivative of the assistant circuit unit 384 can be adjusted by varying the width to length ratio (W/L) of the NMOS transistor N2.
FIG. 7 shows a block diagram of the first amplifying unit 38 according to another embodiment. In this embodiment, the assistant circuit unit 384 comprises a plurality of NMOS transistors N2A, N2B, and N2C having the same width to length ratio (W/L) as the NMOS transistor N1. The drain terminal of the NMOS transistors N2A, N2B, and N2C is connected to the power source VDD. The source terminal of the NMOS transistors N2A, N2B, and N2C is connected to the reference voltage potential, and the gate terminal of the NMOS transistors N2A, N2B, and N2C is connected to the bias voltage Vth1. The linearity of the main circuit unit 382 can be adjusted by varying the number of parallel NMOS transistors N2A, N2B, and N2C. In another embodiment, the width to length ratio (W/L) of the parallel NMOS transistors N2A, N2B, and N2C is not the same as the width to length ratio (W/L) of the NMOS transistor N1. The linearity of the main circuit unit 382 can be adjusted by varying the bias voltage Vth1, or by varying the width to length ratio (W/L) of the parallel NMOS transistors N2A, N2B, and N.
FIG. 5 shows a block diagram of the second amplifying unit 39 according to one embodiment. The second amplifying unit 39 comprises AC coupling capacitors C4, C5, and C6, a main circuit unit 392, and an assistant circuit unit 394. Referring to FIG. 5, an input signal in2 is applied to the main circuit unit 392 and the assistant circuit unit 394 via the AC coupling capacitors C4 and C5, respectively. In one embodiment, the main circuit unit 392 is a PMOS transistor P1. The drain terminal of the PMOS transistor P1 is connected to the power source VDD and to the output terminal out2 via the AC coupling capacitors C6. The source terminal of the PMOS transistor P1 is connected to the reference voltage potential, and the gate terminal of the PMOS transistor P1 is connected to a bias voltage Vbias2.
Referring to FIG. 5, the assistant circuit unit 394 is configured to assist the linearity of the main circuit unit 392. In one embodiment, the assistant circuit unit 394 is a PMOS transistor P2. The drain terminal of the PMOS transistor P2 is connected to the power source VDD and to the output terminal out2 via the AC coupling capacitors C6. The source terminal of the PMOS transistor P2 is connected to the reference voltage potential, and the gate terminal of the PMOS transistor P2 is connected to a bias voltage Vth2. The bias voltage Vth2 is used to adjust the peak value of the second order derivative of the assistant circuit unit 394 so as to suppress the peak value of the second order derivative of the main circuit unit 392. In addition, the second order derivative of the assistant circuit unit 394 can be adjusted by varying the width to length ratio (W/L) of the PMOS transistor P2.
In another embodiment, the assistant circuit unit 394 comprises a plurality of PMOS transistors P2A, P2B, and P2C having the same width to length ratio (W/L) as the PMOS transistor P1. The plurality of PMOS transistors P2A, P2B, and P2C are connected in parallel. The linearity of the main circuit unit 392 can be adjusted by varying the number of parallel PMOS transistors. In yet another embodiment, the width to length ratio (W/L) of the parallel PMOS transistors is not the same as the width to length ratio (W/L) of the PMOS transistor P1. The linearity of the main circuit unit 392 can be adjusted by varying the bias voltage Vth2, or by varying the width to length ratio (W/L) of the parallel PMOS transistors.
FIG. 8 shows a block diagram of the first amplifying unit 38 according to yet another embodiment. In this embodiment, the main circuit unit 382 is a PMOS transistor P3. The drain terminal of the PMOS transistor P3 is connected to the power source VDD, the source terminal is connected to the reference voltage potential, and the gate terminal is connected to the bias voltage Vbias1. The assistant circuit unit 384 is a PMOS transistor P4. The drain terminal of the PMOS transistor P4 is connected to the power source VDD, the source terminal is connected to the reference voltage potential, and the gate terminal is connected to the bias voltage Vth1. The linearity of the main circuit unit 382 can be adjusted by varying the bias voltage Vth1, or varying the width to length ratio (W/L) of the PMOS transistor P4.
In another embodiment, the assistant circuit unit 384 comprises a plurality of PMOS transistors having the same width to length ratio (W/L) as the PMOS transistor P3. The plurality of PMOS transistors are connected in parallel. The linearity of the main circuit unit 382 can be adjusted by varying the number of parallel PMOS transistors.
FIG. 9 shows a block diagram of the second amplifying unit 39 according to yet another embodiment. In this embodiment, the main circuit unit 392 is an NMOS transistor N3. The drain terminal of the NMOS transistor N3 is connected to the power source VDD, the source terminal is connected to the reference voltage potential, and the gate terminal is connected to the bias voltage Vbias2. Also, the assistant circuit unit 394 is an NMOS transistor N4. The drain terminal of the NMOS transistor N4 is connected to the power source VDD, the source terminal is connected to the reference voltage potential, and the gate terminal is connected to the bias voltage Vth2. The linearity of the main circuit unit 392 can be adjusted by varying the bias voltage Vth2, or by varying the width to length ratio (W/L) of the NMOS transistor N4.
In another embodiment, the assistant circuit unit 394 comprises a plurality of NMOS transistors having the same width to length ratio (W/L) as the NMOS transistor N3. The plurality of NMOS transistors is connected in parallel. The linearity of the main circuit unit 392 can be adjusted by varying the number of parallel PMOS transistors.
The present disclosure also discloses an exemplary embodiment of method for improving the linearity of the amplifier circuit. In one embodiment, the method first receives an input signal. The first amplifying unit 38 conducts in one half cycle of an input signal, and the second amplifying unit 39 cuts off. The first amplifying unit 38 cuts off in the other half cycle of an input signal, and the second amplifying unit 39 conducts. Next, a first control signal is received to adjust the linearity of the first amplifying unit 38 when the first amplifying unit 38 conducts, and a second control signal is received to adjust the linearity of the second amplifying unit 39 when the second amplifying unit 39 conducts. In one embodiment, the first control signal is the bias voltage Vth1, and the second control signal is the bias voltage Vth2.
FIG. 10(A) shows a simulation result of the output power Pout, gain, power added efficiency (PAE) of the amplifier circuit 30 according to yet another embodiment. FIG. 10(B) shows a comparison result of the amplifier circuit 30 with the prior art 1 and prior art 2 arrangements. The prior art 1 arrangement is a combination of an architecture disclosed in US publication No. 2006/0049876 with an additional resonator circuit. The resonator circuit is used to suppress the second-order harmonic so as to achieve the same linearity as the amplifier circuit 30. The prior art arrangement 2 is a combination of an architecture disclosed in US publication No. 2006/0049876 with two additional input and output baluns. The baluns are used to suppress the second-order harmonic so as to achieve the same linearity as the amplifier circuit 30. Referring to FIG. 10(B), the PAE of the amplifying circuit of an embodiment can increase 10% to 20% compared to the prior art 1 and prior art 2 arrangements.
The above-described exemplary embodiments are intended to be illustrative only. Those skilled in the art may devise numerous alternative embodiments without departing from the scope of the following claims.
Patent Application
20100244963
