Information
-
Patent Grant
-
6703960
-
Patent Number
6,703,960
-
Date Filed
Thursday, June 20, 200222 years ago
-
Date Issued
Tuesday, March 9, 200420 years ago
-
Inventors
-
Original Assignees
-
Examiners
-
CPC
-
US Classifications
Field of Search
US
- 341 155
- 341 156
- 341 158
- 341 159
- 341 160
-
International Classifications
-
Abstract
Disclosed herein is an analog-to-digital converter having first and second comparator stages, a voltage reference stage, a switching stage, and an encoder. The first comparator stage receives an analog signal and a threshold and outputs a control signal. The voltage reference stage receives the control signal and outputs one of two or more sets of reference voltages. The second comparator stage receives the analog signal, as well as the set of reference voltages output from the voltage reference stage, and outputs a thermometer code in response to comparisons of the analog signal to the reference voltages. The switching stage receives the control signal, and in response thereto, variously couples inputs of the encoder to: bits of the thermometer code output from the second comparator stage, a first potential, or a second potential. Methods for converting analog signals to digital signals are also disclosed.
Description
FIELD OF THE INVENTION
The invention pertains to analog-to-digital converters.
BACKGROUND OF THE INVENTION
Analog-to-digital converters (ADCs) have existed for decades and are a key factor in the quality and speed of many test systems. One type of commonly used ADC is the flash ADC. A flash ADC is advantageous in many applications in that 1) it is easy to construct, 2) it has good matching, and 3) it has little intrinsic delay (i.e., it can perform fast A/D conversions).
An exemplary flash ADC is illustrated in FIG.
9
. The ADC comprises a resistor network, a plurality of comparators, and an encoder. The resistor network serves to provide a different reference voltage to each of the comparators. Thus, when an analog voltage signal, V
IN
, is received by each of the comparators, each comparator compares the analog voltage signal to a different reference voltage. If the analog voltage signal is greater than a comparator's reference voltage, the comparator drives its output high. If the analog signal is less than a comparator's reference voltage, the comparator drives its output low. In this manner, the comparators generate a thermometer code output (i.e., an output in which bits are consecutively asserted, beginning with a least significant bit). The thermometer code output generated by the comparators is then converted to a binary digital signal (B
3
, B
2
, B
1
) via the encoder.
Although the flash ADC is often relied on for its simplicity and speed, its advantages must sometimes be weighed against a number of disadvantages. One of its disadvantages is a high component count. Although the 3-bit ADC shown in
FIG. 9
only requires eight resistors, eight comparators, and an encoder, an 8-bit ADC would require 256 resistors, 256 comparators, and an encoder. A flash ADC's resistor and comparator count therefore grows exponentially with respect to the number of bits in its output (i.e., an N-bit flash ADC requires
2
N
resistors and
2
N
comparators).
Another disadvantage of the flash ADC is its high aspect ratio. To maintain good matching, each of a flash ADC's
2
N
resistor and comparator slices is typically stacked end-to-end. Since the number of stages in a flash ADC (i.e., resistor, comparator, and encoder) remains constant regardless of the value of N, a flash ADC with even a modest value of N will have a high aspect ratio. A high aspect ratio is problematic in that it makes a flash ADC difficult to integrate with other components on a die.
Related to the problem of high aspect ratio is the problem of input impedance mismatch. Input impedance mismatch results from the variance in signal route lengths needed to supply the analog voltage signal, V
IN
, to each of a flash ADC's comparators. One can appreciate that this problem is exacerbated by higher values of N.
Other disadvantages of the flash ADC include a high input capacitance and excessive power dissipation. As the value of N is increased, the parasitic capacitance seen by the analog voltage signal, V
IN
, grows exponentially. Likewise, the power consumed by a flash ADC grows exponentially.
SUMMARY OF THE INVENTION
In a first embodiment of the invention, an analog-to-digital converter comprises first and second comparator stages, a voltage reference stage, a switching stage, and an encoder. The first comparator stage receives an analog signal and a threshold and outputs a control signal. The voltage reference stage receives the control signal and outputs one of two or more sets of reference voltages. The second comparator stage receives the analog signal, as well as the set of reference voltages output from the voltage reference stage, and outputs a thermometer code in response to comparisons of the analog signal to the reference voltages. The switching stage receives the control signal, and in response thereto, variously couples inputs of the encoder to: bits of the thermometer code output from the second comparator stage, a first potential, or a second potential.
In a second embodiment of the invention, a method for converting analog signals to digital signals commences with the comparison of an analog signal, V
IN
, to (V
MAX
−V
MIN
)/2, where V
MAX
and V
MIN
define an expected voltage range for V
IN
. In response to V
IN
being greater than (V
MAX
−V
MIN
)/2, the LSB inputs of an encoder are driven to a first potential, and the MSB inputs of the encoder are determined by comparing V
IN
to a number of reference voltages ranging from (V
MAX
−V
MIN
)/2 to V
MAX
. In response to V
IN
being less than (V
MAX
−V
MIN
)/2, the MSB inputs of the encoder are driven to a second potential, and the LSB inputs of the encoder are determined by comparing V
IN
to a number of reference voltages ranging from V
MIN
to (V
MAX
−V
MIN
)/2. A digital signal is then output from the encoder.
In a third embodiment of the invention, a method for converting analog signals to digital signals commences as an analog signal is input to a comparator stage. At or about the same time, the analog signal is compared to at least one threshold. In response thereto, a voltage reference stage is programmed to deliver one of two or more sets of reference voltages to the comparator stage. Also, and in response to the comparison(s) of the analog signal to the threshold(s), different sets of an encoder's inputs are coupled to either: a first potential, a second potential, or outputs of the comparator stage. A digital signal is then output from the encoder.
BRIEF DESCRIPTION OF THE DRAWINGS
Illustrative and presently preferred embodiments of the invention are illustrated in the drawings, in which:
FIG. 1
illustrates an analog-to-digital converter (ADC);
FIG. 2
illustrates an exemplary embodiment of the
FIG. 1
ADC, wherein the ADC is configured to operate in a first mode;
FIG. 3
illustrates the
FIG. 2
ADC configured to operate in a second mode;
FIG. 4
illustrates an exemplary embodiment of the voltage reference stage switches shown in
FIGS. 2 & 3
;
FIG. 5
illustrates an exemplary embodiment of the switching stage switches shown in
FIGS. 2 & 3
;
FIG. 6
illustrates a first exemplary method for converting analog signals to digital signals;
FIG. 7
illustrates a second exemplary method for converting analog signals to digital signals;
FIG. 8
illustrates an ADC that operates in conformance with the method shown in
FIG. 7
; and
FIG. 9
illustrates a flash ADC.
DESCRIPTION OF THE PREFERRED EMBODIMENT
FIG. 1
illustrates an analog-to-digital converter (ADC)
100
comprising first and second comparator stages
102
,
106
, a voltage reference stage
104
, an encoder
110
, and a switching stage
108
. The first comparator stage
102
receives an analog signal (V
IN
) and a threshold (VREF) and outputs a control signal (CTRL). The voltage reference stage
104
receives the control signal and outputs one of two or more sets of reference voltages. The second comparator stage
106
receives the analog signal (V
IN
) and the set of reference voltages output from the voltage reference stage
104
, and in response to comparisons of the analog signal to the reference voltages, outputs a thermometer code. The switching stage
108
receives the control signal (CTRL), and in response thereto, variously couples inputs of the encoder
110
to 1) bits of the thermometer code output from the second comparator stage
106
, 2) a first potential, or 3) a second potential. The encoder's output is a digital signal such as a binary digital signal (B
3
, B
2
, B
1
).
FIGS. 2 & 3
illustrate an exemplary implementation of the
FIG. 1
ADC. In
FIGS. 2 & 3
, the first comparator stage
102
consists of a single comparator
202
. The comparator
202
receives an analog voltage signal (V
IN
) and a voltage threshold (VREF), and in response to a comparison of these two input signals, outputs a control signal (CTRL). In the exemplary embodiment shown, the control signal is driven low when V
IN
<VREF, and HIGH when V
IN
>VREF. The voltage threshold (VREF) is equal to (V
MAX
−V
MIN
)/2, where V
MAX
is greater than or equal to a maximum voltage that V
IN
is expected to assume, and where V
MIN
is less than or equal to a minimum voltage that V
IN
is expected to assume. In this manner, the control signal (CTRL) reflects whether the analog voltage signal (V
IN
) is above or below a midpoint of a voltage range extending from V
MIN
to V
MAX
.
The control signal (CTRL) output from the first comparator stage
102
is provided to a voltage reference stage
104
. In response to receiving the control signal, the voltage reference stage
104
outputs one of two sets of reference voltages. The first set of reference voltages ranges from V
MIN
to VREF (i.e., (V
MAX
−V
MIN
)/2) and is output when the control signal (CTRL) is low (i.e., when V
IN
<VREF). The second set of reference voltages ranges from VREF to V
MAX
and is output when the control signal (CTRL) is high (i.e., when V
IN
>VREF).
Each set of reference voltages may be derived from a resistor network. As shown in
FIG. 2
, the resistor network may comprise a stick of series connected resistors
204
,
206
,
208
,
210
, with each reference voltage being output from an end of one of the series connected resistors
204
-
210
. As defined herein, a stick of series connected resistors includes, but is not limited to, both of the following: 1) discrete resistive components connected in series
204
-
210
, and 2) a continuous resistive component from which a plurality of taps are taken.
As shown in
FIGS. 2 & 3
, ends of the resistor stick
204
-
210
are alternately coupled to differing voltages by a number of switches
212
,
214
, with each of the switches being operated by the control signal (CTRL) provided by the first comparator stage
102
. A first of the switches
212
is configured to alternately couple a first end of the resistor stick
204
-
210
to V
MAX
or VREF, and a second of the switches
214
is configured to alternately couple a second end of the resistor stick
204
-
210
to VREF or V
MIN
. In this manner, the resistor stick
204
-
210
may be coupled between V
MAX
and VREF when the control signal indicates, that V
IN
is greater than VREF, and the resistor stick
204
-
210
may be coupled between VREF and V
MIN
when the control signal indicates that V
IN
is less than VREF. Thus, depending upon the state of the control signal, one of two sets of reference voltages may be output from the resistor stick
204
-
210
.
The reference voltages output from the voltage reference stage
104
are supplied to a second comparator stage
106
. The second comparator stage
106
comprises a plurality of comparators
216
,
218
,
220
,
222
that, taken together, produce a thermometer code output (i.e., an output in which bits are consecutively asserted, beginning with a least significant bit). Each of the comparators
216
-
222
comprises two inputs, one of which is coupled to the analog voltage signal (V
IN
), and one of which is coupled to a unique output from the voltage reference stage
104
. In this manner, each comparator
216
-
222
receives a different output from the voltage reference stage
104
and compares the analog voltage signal to a different reference voltage. The output of each comparator
216
-
222
is coupled to two inputs of an encoder
110
via a switching stage
108
.
The thermometer code that is output from the second comparator stage
106
is provided to a switching stage
108
. The switching stage
108
alternately couples the output of each comparator
216
-
222
in the second comparator stage
106
to a most significant bit (MSB) input and a least significant bit (LSB) input of an encoder
110
. In
FIG. 2
, the outputs of the second comparator stage
106
are shown coupled to the encoder's LSB inputs (I
0
-I
3
). In
FIG. 3
, the outputs of the second comparator stage
106
are shown coupled to the encoder's MSB inputs (i.e., I
4
-I
7
). When an encoder input is not coupled to a comparator
216
-
222
of the second comparator stage
106
, the encoder input is coupled to either a first (LOW) or second (HIGH) potential.
In
FIGS. 2 & 3
, the switching performed by the switching stage
108
is accomplished by eight switches
224
,
226
,
228
,
230
,
232
,
234
,
236
,
238
, each of which is operated by the control signal (CTRL) produced by the first comparator stage
102
. Half of the switches
224
-
230
are configured to alternately couple MSB inputs of the encoder
110
to either 1) outputs of the second comparator stage
106
, or 2) a LOW potential. The other half of the switches
232
-
238
are configured to alternately couple LSB inputs of the encoder to either 1) outputs of the second comparator stage
106
, or 2) a HIGH potential.
The switching stage
108
shown in
FIGS. 2 & 3
operates as follows. When CTRL is pulled low as a result of V
IN
being less than VREF, the switches
224
-
230
coupled to the encoder's MSB inputs couple the encoder's MSB inputs to a LOW potential (see FIG.
2
). At the same time, the switches
232
-
238
coupled to the encoder's LSB inputs couple the encoder's LSB inputs to outputs of the second comparator stage
106
.
In
FIG. 3
, CTRL is pulled high as a result of V
IN
being greater than VREF. When CTRL is pulled high, the switches
224
-
230
coupled to the encoder's MSB inputs couple the encoder's MSB inputs to outputs of the second comparator stage
106
. At the same time, the switches
232
-
238
coupled to the encoder's LSB inputs couple the encoder's LSB inputs to a HIGH potential.
As will be described in greater detail later in this description, the switches
224
-
238
of the switching stage
108
may comprise a plurality of pass gates
500
,
504
that correspond to the inputs of the encoder
110
(see FIG.
5
). Each pass gate
500
,
504
may comprise 1) an input that is coupled to an output of one of the plurality of comparators
216
-
222
, and 2) an output that is coupled to an input of the encoder
110
. Each pass gate
500
,
504
may then be controlled by means of the control signal (CTRL) that is generated by the first comparator stage
102
.
The switches
224
-
238
may also comprise a number of transistors
502
,
506
. A first number of transistors
506
may be coupled between outputs of the pass gates and a first potential (e.g., LOW), and a second number of transistors
502
may be coupled between outputs of the pass gates and a second potential (e.g., HIGH). All of the transistors
502
,
506
may be controlled by means of the control signal (CTRL).
The theory behind the ADC illustrated in
FIGS. 2 & 3
is as follows. In a conventional flash ADC
900
(FIG.
9
), all of an encoder's MSB inputs are driven LOW until V
IN
exceeds a predetermined voltage. Once V
IN
exceeds the predetermined voltage, all of the encoder's LSB inputs are driven HIGH. Although conventional flash ADCs fail to give special meaning to this predetermined voltage, the inventor has recognized its importance and designated it VREF in this description.
As shown in
FIG. 9
, VREF falls at the midpoint of a conventional flash ADC's resistor stick
902
-
916
. Since the structure of a conventional flash ADC
900
is vertically symmetrical (i.e., constructed with an equal number of resistors and comparators above and below VREF), the
2
N
resistors
902
-
916
and comparators
918
-
932
shown in
FIG. 9
may be replaced with the
2
N−1
resistors
204
-
210
and comparators
216
-
222
of the voltage reference and second comparator stages
104
,
106
illustrated in
FIGS. 2 & 3
. An additional comparator
202
and plurality of switches
212
,
214
,
224
-
238
may then be used to switch the
2
N−1
resistors
204
-
210
and comparators
216
-
222
into their
FIG. 2
configuration (i.e., when V
IN
<VREF) or their
FIG. 3
configuration (i.e., when V
IN
>VREF). In this manner, the total number of comparators needed to construct a flash-type ADC may be reduced from
2
N
to
2
N−1
+1, and the total number of resistors may also be reduced from
2
N
to
2
N−1
.
Just like the ADC
900
illustrated in
FIG. 9
, the ADC
200
illustrated in
FIGS. 2 & 3
is easy to construct. However, the ADC
200
illustrated in
FIGS. 2 & 3
also provides many advantages over the ADC illustrated in
FIG. 9
, such as: 1) approximately half the input capacitance, 2) a lower aspect ratio, 3) reduced power dissipation, 4) a reduction in the impedance mismatch at the inputs to the second comparator stage
106
(as a result of there being fewer stacked comparators
216
-
222
in the stage), and 5) a lower probability of component mismatch in the second comparator stage
106
(again, as a result of there being fewer comparators
216
-
222
in the stage).
The ADC
200
illustrated in
FIGS. 2 & 3
may be fabricated using a variety of technologies, including, for example, Complimentary Metal-Oxide Semiconductor (CMOS) and bipolar technologies. However, a currently preferred technology is CMOS Silicon-On-Insulator technology, in that it provides a lower-power, higher-speed alternative as compared to bipolar and bulk CMOS technologies.
FIGS. 4 & 5
illustrate exemplary embodiments of the switching components
212
,
214
,
224
-
238
illustrated in
FIGS. 2 & 3
.
FIG. 4
illustrates an exemplary embodiment of the voltage reference stage switches
212
,
214
. Each switch
212
,
214
comprises a pair of pass gates
400
/
402
,
404
/
406
that have their outputs coupled to one end of the resistor stick
204
-
210
. Each pass gate
400
-
406
receives the control signals CTRL and CNTRL_INV (with CTRL_INV being supplied by an inverter
408
that receives CTRL at its input).
The pass gates
400
,
402
of the upper switch
212
are respectively configured to receive the voltages VREF and V
MAX
at their inputs. The pass gate
400
that receives VREF is configured to 1) pass VREF to its output when CTRL is pulled low (i.e., when V
IN
<VREF), and 2) pass nothing when CTRL is pulled high (i.e., when V
IN
>VREF). The pass gate
402
that receives V
MAX
is configured to 1) pass V
MAX
to its output when CTRL is pulled high, and 2) pass nothing when CTRL is pulled low. In this manner, the upper end of the resistor stick
204
-
210
is tied to VREF when V
IN
<VREF, and to V
MAX
when V
IN
>VREF.
The pass gates
404
,
406
of the lower switch
214
function similarly to those of the upper switch
212
, but with different input voltages being passed. The first pass gate
404
receives V
MIN
at its input and 1) passes V
MIN
to its output when CTRL is pulled low (i.e., when V
IN
<VREF), and 2) passes nothing when CTRL is pulled high (i.e., when V
IN
>VREF). The second pass gate
406
receives VREF at its input and 1) passes VREF to its output when CTRL is pulled high, and 2) passes nothing when CTRL is pulled low. In this manner, the lower end of the resistor stick
204
-
210
is tied to V
MIN
when V
IN
<VREF, and to V
MAX
when V
IN
>VREF.
FIG. 5
illustrates an exemplary embodiment of the switching stage switches
224
-
238
found in
FIGS. 2 & 3
. Specifically,
FIG. 5
illustrates the switches
224
,
232
that are coupled to inputs
13
and
17
of the encoder
110
shown in
FIGS. 2 & 3
.
The switch
232
that is coupled to input
13
of the encoder
110
comprises a pass gate
500
and a pull-up transistor
502
, each of which is coupled to input
13
of the encoder
110
via the node TGATE_LSB. The pull-up transistor
502
is coupled via its source and drain between a HIGH potential and the node TGATE_LSB. The gate of the transistor
502
is driven by CTRL so that the node TGATE_LSB is pulled high when V
IN
>VREF. When V
IN
<VREF, the pull-up transistor
502
ceases to conduct so that the node TGATE_LSB may be driven by the pass gate
500
. The pass gate
500
is controlled by the signals CTRL_DELAY and DELAY_INV (with CTRL_DELAY being supplied by a buffer
508
that receives CTRL at its input, and with DELAY_INV being supplied by inverters
510
,
512
that receive CTRL_DELAY at their inputs). Note that the signals CTRL_DELAY and DELAY_INV are merely delayed versions of the signals CTRL and CTRL_INV. The signals CTRL_DELAY and DELAY_INV are used to control the pass gate
500
so that, in the event that V
IN
>VREF, node TGATE_LSB may be pulled high prior to a drive fight being initiated by the pass gate
500
. The input to the pass gate
500
is coupled to the output of a corresponding comparator
216
so that the pass gate
500
passes the output of the comparator
216
to the node TGATE_LSB when V
IN
<VREF.
The switch
224
that is coupled to input
17
of the encoder
110
comprises a pass gate
504
and a pull-down transistor
506
, each of which is coupled to input
17
of the encoder
110
via the node TGATE_MSB. The pull-down transistor
506
is coupled via its source and drain between a LOW potential and the node TGATE_MSB. The gate of the transistor
506
is driven by CTRL so that the node TGATE_MSB is pulled low when V
IN
<VREF. When V
IN
>VREF, the pull-down transistor
506
ceases to conduct so that the node TGATE_MSB may be driven by the pass gate
504
. The pass gate
504
is controlled by the signals CTRL_DELAY and DELAY_INV. The input to the pass gate
504
is coupled to the output of a corresponding comparator
216
so that the pass gate
504
passes the output of the comparator
216
to the node TGATE_MSB when V
IN
>VREF.
Having completed a description of the analog-to-digital converters
100
,
200
and components thereof illustrated in
FIGS. 1-5
, a number of methods
600
,
700
for making analog-to-digital conversions will now be described.
FIG. 6
illustrates a first exemplary method
600
for converting analog signals to digital signals. The method
600
commences with the comparison
602
of an analog signal, V
IN
, to (V
MAX
−V
MIN
)/2, where V
MAX
and V
MIN
define an expected voltage range for V
IN
. In response to V
IN
being greater than (V
MAX
−V
MIN
)/2, the LSB inputs of an encoder are driven
604
to a first potential, and the MSB inputs of the encoder are determined
604
by comparing V
IN
to a number of reference voltages ranging from (V
MAX
−V
MIN
)/2 to V
MAX
. In response to V
IN
being less than (V
MAX
−V
MIN
)/2, the MSB inputs of the encoder are driven
606
to a second potential, and the LSB inputs of the encoder are determined
606
by comparing V
IN
to a number of reference voltages ranging from V
MIN
to (V
MAX
−V
MIN
)/2. In either case, a digital signal is then output
608
from the encoder.
As is demonstrated in
FIG. 5
, the MSB and LSB inputs of an encoder
110
may be driven to first and second potentials by means of pull-up and pull-down transistors
502
,
506
that are coupled to the encoder's MSB and LSB inputs. The pull-up and pull-down transistors
502
,
506
may then be controlled by means of a control signal (CTRL), wherein the state of the control signal is determined by the comparison of V
IN
to (V
MAX
−V
MIN
)/2.
Although not required, the method
600
illustrated in
FIG. 6
may comprise isolating an encoder's LSB and MSB inputs from logic
216
-
222
(
FIG. 2
) that determines their respective states. Thus, when the LSB inputs of an encoder
110
are driven to a first potential, the encoder's LSB inputs may be isolated from logic
216
-
222
that determines their state when V
IN
is less than (V
MAX
−V
MIN
)/2. Likewise, when the MSB inputs of an encoder
110
are driven to a second potential, the encoder's MSB inputs may be isolated from logic
216
-
222
that determines their state when V
IN
is greater than (V
MAX
−V
MIN
)/2.
The reference voltages referred to in the
FIG. 6
method may be derived in a number of ways. An exemplary way, however, involves coupling
610
a resistor stick
204
-
210
between first and second programmable end voltages. Each of the number of reference voltages are then derived from the resistor stick
204
-
210
by programming
612
the first and second programmable end voltages in response to the comparison of V
IN
to (V
MAX
−V
MIN
)/2.
FIG. 7
illustrates a second exemplary method
700
for converting analog signals to digital signals. The method
700
commences with the comparison
702
of an analog signal to at least one threshold, as well as the input
704
of the analog signal to a comparator stage. In response to comparing the analog signal to the at least one threshold, a voltage reference stage is programmed
706
,
708
to deliver one of two or more sets of reference voltages to the comparator stage. Also in response to comparing the analog signal to the at least one threshold, different sets of an encoder's inputs are then coupled
710
to either a first potential, a second potential, or outputs of the comparator stage. Finally, a digital signal is output
712
from the encoder.
The voltage reference stage may be programmed to deliver one of two or more sets of reference voltages to the comparator stage by 1) switching first and second voltages applied to ends of a resistor stick, in response to the afore-mentioned comparison of an analog signal to at least one threshold, and then 2) utilizing different taps from the resistor stick to deliver a set of reference voltages to the comparator stage.
Each comparison of the analog signal to a threshold may be used to generate a control signal, wherein the control signal is indicative of the result of the comparison. If one or more such control signals are generated, an encoder's inputs may be coupled to first and second potentials by means of pull-up and pull-down transistors, with each transistor being controlled by one or more of the afore-mentioned control signal(s). The encoder's inputs may be coupled to outputs of a comparator stage by means of pass gates that are also controlled by the afore-mentioned control signal(s).
In one embodiment of the
FIG. 7
method
700
, the at least one threshold is only one threshold, and the two or more sets of reference voltages are only two sets of reference voltages. In this embodiment, the first set of reference voltages may range, for example, from V
MIN
to (V
MAX
−V
MIN
)/2, and the second set of reference voltages may range, for example, from (V
MAX
−V
MIN
)/2 to V
MAX
. Comparing the analog signal to the at least one threshold then comprises comparing a voltage level of the analog signal to (V
MAX
−V
MIN
)/2.
In another embodiment of the
FIG. 7
method, an analog signal is compared to a plurality of thresholds. For example, refer to the ADC
800
illustrated in FIG.
8
. The ADC
800
comprises a first comparator stage
802
in which an analog signal, V
IN
, is compared to three different thresholds (REF_A, REF_B, and REF_C). Results of the various comparisons (CTRL_A, CTRL_B, CTRL_C) are used to program a voltage reference stage
804
to deliver one of two different sets of reference voltages to a second comparator stage
806
. The second comparator stage
806
compares each of the reference voltages to the analog signal, V
IN
. Then, in response to various control signals generated as a result of V
IN
being compared to REF_A, REF_B, and REF_C, an encoder's inputs are coupled to either a first potential, a second potential, or outputs of the second comparator stage
808
. The encoder
810
then outputs a digital signal (B
3
, B
2
, B
1
).
Although the ADC
800
shown in
FIG. 8
may provide few advantages over the ADC
200
shown in
FIG. 2
, the teachings illustrated in
FIG. 8
might provide significant advantages for larger values of N (where N is the number of bits output from the ADC
800
). For example, consider an 8-bit ADC that receives only a single reference voltage. In a scenario such as that which is illustrated in
FIG. 2
, an 8-bit ADC would require one comparator in its first comparator stage and 128 comparators in its second comparator stage. This represents a total comparator count of
129
. On the other hand, configuring an 8-bit ADC as shown in
FIG. 8
would raise the number of comparators in the first stage to three, but lower the number of comparators in the second stage to sixty-four. The comparator count is thus reduced to sixty-seven (for a savings of nearly fifty percent).
The
FIG. 8
ADC
800
is advantageous in that it can reduce an ADC's overall component count, as well as improve the aspect ratio of an ADC (for better die usage). However, a disadvantage of the
FIG. 8
ADC
800
is that it can lead to increased switching delay as an input signal is analyzed in the ADC's first comparator stage
802
.
While illustrative and presently preferred embodiments of the invention have been described in detail herein, it is to be understood that the inventive concepts may be otherwise variously embodied and employed, and that the appended claims are intended to be construed to include such variations, except as limited by the prior art.
Claims
- 1. An analog-to-digital converter, comprising:a) a first comparator stage receiving an analog signal and a threshold and outputting a control signal; b) a voltage reference stage receiving said control signal and outputting one of two or more sets of reference voltages; c) a second comparator stage receiving said analog signal and said one of two or more sets of reference voltages, said second comparator stage outputting a thermometer code in response to comparisons of said analog signal to said reference voltages; d) an encoder; and e) a switching stage receiving said control signal, and in response thereto, variously coupling inputs of said encoder to: bits of the thermometer code output from the second comparator stage, a first potential, or a second potential.
- 2. An analog-to-digital converter as in claim 1, wherein:a) said two or more sets of reference voltages are comprised of two sets, a first of which ranges from VMIN to (VMAX−VMIN)/2, and a second of which ranges from (VMAX−VMIN)/2 to VMAX; and b) said at least one threshold comprises the threshold (VMAX−VMIN)/2.
- 3. An analog-to-digital converter as in claim 1, wherein the voltage reference stage comprises:a) a stick of series connected resistors, wherein each of said reference voltages is output from an end of one of the series connected resistors; and b) a pair of switches controlled by said control signal, wherein said pair of switches alternately couple ends of said stick to differing voltages.
- 4. An analog-to-digital converter as in claim 1, wherein the second comparator stage comprises a plurality of comparators producing said thermometer code, and wherein each comparator comprises:a) an input coupled to said voltage reference stage; b) an input coupled to said analog signal; and c) an output coupled to two inputs of said encoder via said switching stage.
- 5. An analog-to-digital converter as in claim 4, wherein the output of each comparator in the second comparator stage is alternately coupled, via said switching stage, to an MSB input and an LSB input of said encoder.
- 6. An analog-to-digital converter as in claim 4, wherein the switching stage comprises a plurality of pass gates corresponding to the inputs of said encoder, wherein each pass gate is controlled by said control signal, and wherein each pass gate comprises:a) an input coupled to an output of one of said plurality of comparators; and b) an output coupled to an input of said encoder.
- 7. An analog-to-digital converter as in claim 6, wherein, for each of said plurality of comparators, one of said pass gates couples the comparator's output to an MSB input of said encoder, and another of said pass gates couples the comparator's output to an LSB input of said encoder.
- 8. An analog-to-digital converter as in claim 1, wherein the switching stage comprises a plurality of pass gates corresponding to the inputs of said encoder, wherein each pass gate is controlled by said control signal, and wherein each pass gate comprises:a) an input coupled to an output of said comparator stage; and b) an output coupled to an input of said encoder.
- 9. An analog-to-digital converter as in claim 8, wherein the switching stage further comprises:a) a first number of transistors coupled between a first number of outputs of said pass gates and said first potential; and b) a second number of transistors coupled between a second number of outputs of said pass gates and said second potential, wherein the first and second numbers of transistors are controlled by said control signal.
- 10. An analog-to-digital converter as in claim 1, wherein the analog-to-digital converter is implemented in CMOS technology.
- 11. An analog-to-digital converter as in claim 1, wherein the analog-to-digital converter is implemented in CMOS Silicon-On-Insulator technology.
- 12. A method for converting analog signals to digital signals, comprising:a) comparing an analog signal, VIN, to (VMAX−VMIN)/2, where VMAX and VMIN define an expected voltage range for VIN; b) in response to said comparison, generating either a first number of reference voltages ranging from (VMAX−VMIN)/2 to VMAX, or a second number of reference voltages ranging from VMIN to (VMAX−VMIN)/2; c) in response to VIN being greater than (VMAX−VMIN)/2: i) driving LSB inputs of an encoder to a first potential; and ii) determining MSB inputs to the encoder by comparing VIN to a the first number of reference voltages; d) in response to VIN being less than (VMAX−VMIN)/2: i) driving the MSB inputs of the encoder to a second potential; and ii) determining the LSB inputs to the encoder by comparing VIN to a the second number of reference voltages; and e) outputting a digital signal from the encoder.
- 13. A method as in claim 12, further comprising:a) driving the MSB and LSB inputs of the encoder to the first and second potentials by means of pull-up and pull-down transistors that are coupled to the MSB and LSB inputs; and b) controlling the pull-up and pull-down transistors by means of a control signal, wherein the state of the control signal is determined by the comparison of VIN to (VMAX−VMIN)/2.
- 14. A method as in claim 12, further comprising:a) when the LSB inputs of the encoder are driven to the first potential, isolating the LSB inputs from logic that determines the state of the LSB inputs when VIN is less than (VMAX−VMIN)/2; and b) when the MSB inputs are driven to the second potential, isolating the MSB inputs from logic that determines the state of the MSB inputs when VIN is greater than (VMAX−VMIN)/2.
- 15. A method as in claim 12, further comprising:a) coupling a resistor stick between first and second programmable end voltages; and b) deriving each of the number of reference voltages from the resistor stick by programming the first and second end voltages in response to the comparison of VIN to (VMAX−VMIN)/2.
- 16. A method for converting analog signals to digital signals, comprising:a) inputting an analog signal to a comparator stage; b) comparing the analog signal to at least one threshold, and in response thereto: i) programming a voltage reference stage to deliver one of two or more sets of reference voltages to the comparator stage; and ii) coupling different sets of an encoder's inputs to either: A) a first potential; B) a second potential; or C) outputs of the comparator stage; and c) outputting a digital signal from the encoder.
- 17. A method as in claim 16, wherein the at least one threshold is only one threshold, and wherein the two or more sets of reference voltages are only two sets of reference voltages.
- 18. A method as in claim 16, wherein:a) the two or more sets of reference voltages comprises two sets, a first of which ranges from VMIN to (VMAX−VMIN)/2, and a second of which ranges from (VMAX−VMIN)/2 to VMAX; and b) comparing the analog signal to the at least one threshold comprises comparing a voltage level of the analog signal to (VMAX−VMIN)/2.
- 19. A method as in claim 16, wherein programming the voltage reference stage to deliver one of two or more sets of reference voltages comprises:a) in response to comparing the analog signal to the at least one threshold, switching first and second voltages that are applied to ends of a resistor stick; b) utilizing different taps from the resistor stick to deliver a set of reference voltages to the comparator stage.
- 20. A method as in claim 16, wherein comparing the analog signal to the at least one threshold produces a control signal, the method further comprising:a) coupling encoder inputs to the first and second potentials by means of pull-up and pull-down transistors controlled by the control signal; and b) coupling encoder inputs to outputs of the comparator stage by means of pass gates controlled by the control signal.
US Referenced Citations (5)