Patent Application
20120105302
The present invention relates to antennas, and, more particularly, to an antenna used for radio communication performed with a high-frequency signal in a UHF or SHF band.
As an antenna in the related art, Japanese Unexamined Patent Application Publication No. 2008-99234 (Patent Literature 1) discloses an example of a known antenna used in a communication system disclosed. The antenna disclosed in Patent Literature 1 will now be described below with reference to the accompanying
The communication system 500 disclosed in Patent Literature 1 is a system capable of achieving large-capacity transmission by transmitting a high-frequency signal through electric field coupling. More specifically, high-volume data communication can be performed using weak radio waves by applying a communication method, such as a UWB (ultra wide band) communication method, using high frequencies and a wide frequency band to electric field coupling. As illustrated in
The electronic apparatus 501 includes a transmission circuit unit 502, a resonating unit 504, and a transmission electrode 506. The transmission circuit unit 502 is a circuit for generating a high-frequency signal such as a UWB signal. The transmission electrode 506 emits the high-frequency signal generated by the transmission circuit unit 502 as a radio wave. The resonating unit 504 performs impedance matching between the transmission circuit unit 502 and the transmission electrode 506.
On the other hand, the electronic apparatus 511 includes a receiving circuit unit 512, a resonating unit 514, and a receiving electrode 516. The receiving electrode 516 is coupled to the transmission electrode 506 by electric field coupling and receives the radio wave emitted from the transmission electrode 506. The receiving circuit unit 512 performs demodulation and decoding on the radio wave received by the receiving electrode 516. The resonating unit 514 is a circuit for performing impedance matching between the receiving circuit unit 512 and the receiving electrode 516.
The transmission electrode 506 will be described in detail. As illustrated in
The substrate 522 is made of an insulating material. The ground electrode 524 is disposed on the entire undersurface of the substrate 522, and a ground potential is applied to the ground electrode 524. The stub 526 is a linear electrode disposed on the surface of the substrate 522, and has a length approximately half (λ/2) of the wavelength of a high-frequency signal transmitted and received in the communication system 500. The substrate 528 is made of an insulating material, and is disposed on the surface of the substrate 522 so that it partly covers the stub 526. The transmission electrode 506 is a rectangular electrode disposed on the surface of the substrate 528. The via-hole conductor 530 connects the transmission electrode 506 and the stub 526. The via-hole conductor 532 connects the stub 526 and the ground electrode 524. As illustrated in
On the other hand, like the transmission electrode 506, the receiving electrode 516 is a part of the antenna 550 illustrated in
In the antennas 520 and 550 having the above-described structure, the transmission electrode 506 and the receiving electrode 516 are close to each other so that a predetermined distance (for example, 3 cm) is set between the transmission electrode 506 and the receiving electrode 516. More specifically, the antenna 520 is designed so that a predetermined capacitance is generated between the transmission electrode 506 and the receiving electrode 516 and the input impedance of the antenna 520 and the output impedance (for example 50Ω) of the transmission circuit unit 502 match (that is, impedance matching) when the distance between the transmission electrode 506 and the receiving electrode 516 becomes a predetermined distance. Similarly, the antenna 550 is designed so that a predetermined capacitance is generated between the transmission electrode 506 and the receiving electrode 516 and the output impedance of the antenna 550 and the input impedance of the receiving circuit unit 512 match (that is, impedance matching) when the distance between the transmission electrode 506 and the receiving electrode 516 becomes a predetermined distance. As a result, the reflectivity of a high-frequency signal output from the transmission circuit unit 502 becomes low, and the high-frequency signal is input into the antenna 520. Since the stub 526 has a length approximately half of the wavelength of the high-frequency signal, a standing wave is generated at the stub 526 as illustrated in
As described previously, the via-hole conductor 530 is connected to the stub 526 at the position apart from the via-hole conductor 532 by a quarter (λ/4) of the wavelength of a high-frequency signal. As illustrated in
The communication system 500 disclosed in Patent Literature 1 has a problem in that it has a low degree of design flexibility. More specifically, as illustrated in
However, when the input-side end portion of the stub 526 and the node of a standing wave exactly match, the input impedance of the stub 526 becomes 0Ω. Accordingly, the impedance matching between a connector 540 connected to the stub 526 and the stub 526 is broken. As a result, a high-frequency signal cannot enter the stub 526. In order to prevent this situation, in the antenna 520, as illustrated in
The present disclosure provides an antenna than can have a high degree of design flexibility.
In accordance with one representative aspect of the disclosure, an antenna includes a ground conductor to which a ground potential is applied, a linear conductor configured to transmit a high-frequency signal, an insulating layer configured to isolate the ground conductor and the linear conductor from each other, and a radiation conductor that is connected between the linear conductor and the ground conductor. The radiation conductor has a line width larger than that of the linear conductor between a point of connection to the linear conductor and a point of connection to the ground conductor, and is configured to emit an electric field.
The inventors realized that in order to meet the above-described design condition for impedance matching, it is necessary to accurately connect the connector 540 to the stub 526. More specifically, the input impedance of the stub 526 whose one end is connected to the ground is low at both ends thereof and high at the center thereof like the standing wave. Furthermore, the rate of change of the input impedance of the stub 526 whose one end is connected to the ground is high at both ends thereof and low at the center thereof like the standing wave. The connector 540 is connected to the end portion of the stub 526. Accordingly, when the point of connection between the connector 540 and the stub 526 is slightly shifted from an original point, the input impedance of the stub 526 is significantly deviated from the output impedance of the connector 540. As a result, the reflectivity of a high-frequency signal cannot be reduced and the high-frequency signal cannot be input from the connector 540 to the stub 526. Because of the above-described reason, it is necessary to accurately connect the connector 540 to the stub 526 in the antenna 520. This leads to low design flexibility. For example, when an RF cable is used as the connector 540 and a characteristic impedance is changed from 50Ω to 35Ω, the point of connection between the connector 540 and the antenna 520 is required to be redesigned. At the time of actual use, the characteristic impedance of a connector or a cable varies from product to product. Therefore, after the antenna 520 has been designed for a specific connector, it is very difficult to change the connector to another connector or a cable. In the antenna 550, a similar problem occurs.
An antenna according to exemplary embodiments that can address the above constraining issues are now described with reference to the accompanying drawings.
The structure of an antenna according to a first exemplary embodiment will now be described with reference to
The antenna 10a can be used in, for example, a communication system such as communication system 500 illustrated in
As illustrated in
As illustrated in
As illustrated in
As illustrated in
As illustrated in
As illustrated in
The via-hole conductor b6 passes through the insulating layer 14b in the z-axis direction, and is connected to the end portion of the connecting conductor 22 in the positive x-axis direction. The via-hole conductors b7 and b8 pass through the insulating layer 14b in the z-axis direction, and are connected to the via-hole conductors b3 and b4, respectively.
As illustrated in
When the insulating layers 14a to 14c having the above-described structures are laminated, the linear conductor 24 and the ground conductor 26 are insulated from each other by the insulating layer 14b. The linear conductor 24 faces the ground conductor 26 via the insulating layer 14b in plan view from the z-axis direction. Accordingly, the linear conductor 24 and the ground conductor 26 provide a microstrip line structure.
The radiation conductor 16 and the ground conductor 26 are insulated from each other by the insulating layers 14a and 14b so that they are not directly connected to each other. The radiation conductor 16 faces the ground conductor 26 via the insulating layers 14a and 14b in plan view from the z-axis direction.
The number of the insulating layers 14 between the radiation conductor 16 and the ground conductor 26 (two layers, the insulating layers 14a and 14b) is larger than that between the linear conductor 24 and the ground conductor 26 (one layer, the insulating layer 14b). Accordingly, a distance d2 between the radiation conductor 16 and the ground conductor 26 in the z-axis direction is larger than a distance d1 between the linear conductor 24 and the ground conductor 26 in the z-axis direction.
As illustrated in
An equivalent circuit diagram of the antenna 10a having the above-described structure is as illustrated in FIG. 3. More specifically, between the terminal conductors 18 and 20, the linear conductor 24, the radiation conductor 16, and the ground conductor 26 are connected in series in this order. A capacitance C1 is generated between the linear conductor 24 and the ground conductor 26. A capacitance C2 is generated between the radiation conductor 16 and the ground conductor 26. The linear conductor 24 generates an inductance L1. The radiation conductor 16 generates an inductance L2. That is, in the antenna 10a, a resonance circuit including the capacitances C1 and C2 and the inductances L1 and L2 is provided.
The antenna 10a is designed so that the capacitances C1 and C2 and the inductances L1 and L2 satisfy the following conditions. More specifically, a relationship represented by equation (1) is established among a center frequency f of a high-frequency signal transmitted in the antenna 10a, the capacitances C1 and C2, and the inductances L1 and L2.
f=2π/√{(L1+L2)×(C1+C2)} (1)
(C2 is substantially zero)
The input impedance Z of the antenna 10a needs to match the output impedance (for example 50Ω) of the transmission circuit unit 502 illustrated in
Z=√/{(L1+L2)/(C1+C2)} (2)
(C2 is substantially zero)
In the antenna 10a, the linear conductor 24 and the radiation conductor 16 are designed so that the capacitances C1 and C2 and the inductances L1 and L2 satisfy equations (1) and (2). Here, the linear conductor 24 and the radiation conductor 16 are preferably designed so that a reactance X1 (|L2/C2|) of the radiation conductor 16 is larger than a reactance X2 (|L1/C1|) of the linear conductor 24.
The antenna 10a having the above-described structure can be used in, for example, the communication system 500 illustrated in
On the other hand, in the antenna 10a used as the resonating unit 514 and the receiving electrode 516, the radiation conductor 16 absorbs the emitted electric field. Subsequently, the high-frequency signal is externally output from the antenna 10a via the linear conductor 24 and the terminal conductor 18.
An exemplary manufacturing method of the antenna 10a will now be described below with reference to
First, the insulating layers 14 that are made of liquid crystal polymer and include copper foil on the entire surfaces thereof are prepared. Subsequently, the radiation conductor 16 and the terminal conductors 18 and 20 illustrated in
Subsequently, the connecting conductor 22 and the linear conductor 24 illustrated in
Subsequently, a laser beam is emitted from the undersurfaces of the insulating layers 14a and 14b to positions at which the via-hole conductors b1 to b8 are to be formed, so that via holes are formed. Subsequently, a conductive paste mainly composed of copper is charged into the via holes formed in the insulating layers 14a and 14b, so that the via-hole conductors b1 to b8 illustrated in
Subsequently, the insulating layers 14a to 14c are laminated in this order. The insulating layers 14a to 14c are press-bonded by applying a force to the insulating layers 14a to 14c from the positive and negative z-axis directions. Consequently, the antenna 10a illustrated in
As will be described later, the antenna 10a having the above-described structure has a high degree of design flexibility. More specifically, in the antenna 520 in the communication system 500 disclosed in Patent Literature 1, a standing wave is generated at the stub 526 and an electric field is emitted from the transmission electrode 506 with the standing wave. In order to generate the standing wave, it is necessary to achieve matching between the input impedance of the stub 526 and the output impedance of the connector 540 by accurately connecting the connector 540 to the stub 526. Therefore, the design flexibility of the antenna 520 is low.
On the other hand, the antenna 10a uses no standing wave to emit an electric field. The antenna 10a includes an LC resonance circuit, and only a high-frequency signal having the center frequency f of the LC resonance circuit is transmitted through the linear conductor 24 and the radiation conductor 16. The line width W2 of the radiation conductor 16 is larger than the line width W1 of the linear conductor 24, and the area of the radiation conductor 16 is larger than that of the linear conductor 24. As a result, the radiation conductor 16 emits an electric field that is changed in accordance with a high-frequency signal. That is, like the antennas 520 and 550, two antennas 10a can communicate with each other by near field radio communication.
In the antenna 10a, the linear conductor 24, the radiation conductor 16, and the ground conductor 26 are connected in series, and an LC resonance circuit is formed between the terminal conductors 18 and 20. Accordingly, the center frequency f of a high-frequency signal transmitted through the antenna 10a is determined by the capacitance C1 and the inductance L1 of the linear conductor 24 and the capacitance C2 and the inductance L2 of the radiation conductor 16 as described previously. The capacitances C1 and C2 and the inductances L1 and L2 can be adjusted by changing the shapes (for example line widths or lengths) of the linear conductor 24 and the radiation conductor 16. That is, in the antenna 10a, impedance matching can be achieved by optionally adjusting a plurality of design factors. On the other hand, in the antenna 520, it is necessary to accurately connect the connector 540 to the stub 526 so that the desired length of the stub 526 is obtained. That is, in the antenna 520, only the length of the stub 526 is used to achieve impedance matching. Thus, the antenna 10a has a higher degree of design flexibility than the antenna 520. By changing the line width or line length of the linear conductor 24 or the presence or absence of a slit portion in a length direction, it is possible to provide a multistage LC resonance circuit that includes the capacitances C1 and the inductances L1 and has a wide band of emission frequencies.
In the antenna 10a, the height in the z-axis direction can be reduced (hereinafter referred to as profile reduction). More specifically, the antenna 520 illustrated in
On the other hand, in the antenna 10a, the radiation conductor 16 included in the LC resonance circuit emits an electric field. Accordingly, dislike the antenna 520, there is no need for the antenna 10a to have the structure of a dipole antenna whose both ends are shorted. The profile reduction of the antenna 10a can be therefore achieved.
In the antenna 10a, as will be described later, the radiation conductor 16 can emit a stronger electric field. More specifically, when the radiation conductor 16 is close to the ground conductor 26, the most part of an electric field emitted from the radiation conductor 16 is directed toward the ground conductor 26 (that is, in the negative z-axis direction) and is consumed by the ground conductor 26. Accordingly, it is difficult for the radiation conductor 16 to emit a strong electric field in the positive z-axis direction.
In the antenna 10a, the distance d2 between the radiation conductor 16 and the ground conductor 26 in the z-axis direction is larger than the distance d1 between the linear conductor 24 and the ground conductor 26 in the z-axis direction. The radiation conductor 16 is therefore apart from the ground conductor 26. As a result, a most part of an electric field emitted from the radiation conductor 16 is directed in the positive z-axis direction. That is, in the antenna 10a, the radiation conductor 16 can emit a stronger electric field.
The ground conductor 26 and the linear conductor 24 form a microstrip line. Therefore, the characteristic impedance (the input impedance and the output impedance) of the linear conductor 24 can easily match the characteristic impedance of the radiation conductor 16 and a characteristic impedance of another component.
In the antennas 10a, even when the distance between two radiation conductors 16 is changed, the transmission characteristic of a high-frequency signal is not deteriorated. More specifically, the antennas 520 and 550 are designed so that, when the distance between the transmission electrode 506 and the receiving electrode 516 becomes a predetermined distance (for example 3 cm), a predetermined capacitance is generated between the transmission electrode 506 and the receiving electrode 516 and the input impedance of the antenna 520 matches the output impedance (for example 50≠) of the transmission circuit unit 502 (that is, impedance matching between them is achieved). Similarly, the antennas 520 and 550 are designed so that, when the distance between the transmission electrode 506 and the receiving electrode 516 becomes a predetermined distance, a predetermined capacitance is generated between the transmission electrode 506 and the receiving electrode 516 and the output impedance of the antenna 550 matches the input impedance of the receiving circuit unit 512 (that is, impedance matching between them is achieved). Accordingly, when the distance between the transmission electrode 506 and the receiving electrode 516 deviates from a predetermined distance, impedance matching is not achieved. In this case, in the antennas 520 and 550, the transmission of a high-frequency signal cannot be performed.
On the other hand, in the antenna 10a, impedance matching with the transmission circuit unit 502 or the receiving circuit unit 512 is achieved with an LC resonance circuit including the linear conductor 24, the ground conductor 26, and the radiation conductor 16. As described previously, the capacitance C2 between the ground conductor 26 and the radiation conductor 16 is substantially zero. Therefore, the impedance of the LC resonance circuit does not depend on the capacitance C2. That is, the impedance is practically determined in accordance with the inductance L1 of the linear conductor 24, the inductance L2 of the radiation conductor 16, and the capacitance C1 between the linear conductor 24 and the ground conductor 26. Even when the distance between the radiation conductors 16 is changed, impedance matching between the antenna 10a and the transmission circuit unit 502 or the receiving circuit unit 512 can be achieved. Accordingly, in the antennas 10a, even when the distance between the radiation conductors 16 is changed, the transmission characteristic of a high-frequency signal is not deteriorated.
Exemplary modifications of the antenna 10a will now be described below with reference to the accompanying drawings.
By making the linear conductor 24′ meander, the inductance L1 of the linear conductor 24′ can be increased. That is, in the antenna 10b, the range of adjustment of the inductance L1 can be increased. As a result, the adjustment of the resonant frequency of the antenna 10b and the impedance matching between the antenna 10b and the transmission circuit unit 502 or the receiving circuit unit 512 can be easily performed.
The linear conductor 24a is connected in parallel to the linear conductor 24. Thus, in the antenna 10c, a plurality of linear conductors connected in parallel, the linear conductors 24 and 24a, may be disposed. As a result, multiple resonances can be obtained and a frequency band can be increased to, for example, 4.48 GHz±200 MHz. The line widths of the linear conductors 24 and 24a may be the same or different. By opening one of the ends of the linear conductors 24 and 24a, an open stub-type linear conductor may be formed.
In the antenna 10d, the point of connection between the radiation conductor 16 and the ground conductor 26 via the via-hole conductor b1 is nearer to the center of the radiation conductor 16 than that in the antenna 10a. Accordingly, in the antenna 10d, the position of the via-hole conductor b1 is farther from the side of the radiation conductor 16 in the positive x-axis direction than that in the antenna 10a. As a result, as illustrated in
In the antenna 10e, since the connecting conductor 22′ is meandering, it functions as an inductive line. Since the via-hole conductor b30 is disposed, the radiation conductor 16 and the ground conductor 26 are connected with a line having two branches as illustrated in
The ground conductor 26′ has the opening O in which no conductor is disposed at a position overlapping the radiation conductor 16 in plan view from the z-axis direction. Therefore, the radiation conductor 16 does not overlap the ground conductor 26′ in plan view from the z-axis direction (a normal direction with respect to the radiation conductor 16 or a stacking direction of the insulating layers 14a to 14c). As a result, little electric field is consumed by the ground conductor 26′. The antenna 10f can therefore emit a stronger electric field from the radiation conductor 16 as compared with the antenna 10a.
In the antenna 10f, since the radiation conductor 16 and the ground conductor 26′ do not face each other, the capacitance C2 generated therebetween is substantially zero. Thus, a capacitance in the antenna 10f is reduced. That is, the input impedance of the antenna 10f as viewed from an input port is practically seen as an inductance, and the output impedance of the input port as viewed from the antenna 10f is seen as 50Ω. By achieving impedance matching at this portion, the reflection characteristic of the input impedance becomes deeper and the favorable reflection characteristic is obtained over a wide frequency band. Thus, when the capacitance in the antenna 10f is reduced, the usable frequency band of the antenna 10f can be increased.
Structure of an antenna according to a second exemplary embodiment will now be described with reference to
As illustrated in
As illustrated in
As illustrated in
As illustrated in
As illustrated in
As illustrated in
The conductor 35 includes a radiation conductor 36a, a connecting conductor 36b, and a linear conductor 36c. As illustrated in
As illustrated in
As illustrated in
The antenna 10g having the above-described structure can obtain an operational effect similar to that of the antenna 10a.
Furthermore, the profile reduction of the antenna 10g can be achieved. More specifically, since the radiation conductor 36a and the ground conductor 38 do not face each other, little electric field emitted from the radiation conductor 36a is consumed by the ground conductor 38 even when the distance between the radiation conductor 36a and the ground conductor 38 in the z-axis direction is reduced. Accordingly, in the antenna 10g, only a single layer, the insulating layer 34a, is needed between the radiation conductor 36a and the ground conductor 38. As a result, the profile reduction of the antenna 10g can be achieved.
The structure of an antenna according to a third exemplary embodiment will now be described below with reference to
As illustrated in
As illustrated in
As illustrated in
As illustrated in
As illustrated in
As illustrated in
As illustrated in
As illustrated in
The leg portion 46b is formed by bending in the negative z-axis direction a protrusion extending from the midpoint of the long side of the radiation portion 46a in the negative x-axis direction toward the negative x-axis direction. The leg portion 46c is formed by bending in the negative z-axis direction a protrusion extending from the midpoint of the long side of the radiation portion 46a in the positive x-axis direction toward the positive x-axis direction. The leg portion 46d is formed by bending in the negative z-axis direction a protrusion extending from the corner of the radiation portion 46a in the negative x-axis direction and the positive y-axis direction toward the negative x-axis direction. The leg portion 46e is formed by bending in the negative z-axis direction a protrusion extending from the corner of the radiation portion 46a in the positive x-axis direction and the positive y-axis direction toward the positive x-axis direction. The leg portion 46f is formed by bending in the negative z-axis direction a protrusion extending from the corner of the radiation portion 46a in the negative x-axis direction and the negative y-axis direction toward the negative x-axis direction. The leg portion 46g is formed by bending in the negative z-axis direction a protrusion extending from the corner of the radiation portion 46a in the positive x-axis direction and the negative y-axis direction toward the positive x-axis direction.
As illustrated in
The antenna 10h having the above-described structure can obtain an operational effect similar to that of the antenna 10a.
In the antenna 10h, the radiation conductor 46 is formed of not copper foil, but a metal plate. Accordingly, in the antenna 10h, by adjusting the lengths of the leg portions 46b to 46g, the capacitance C2 and the inductance L2 of the radiation conductor 46 can be adjusted.
An antenna that is an exemplary modification of the antenna 10h will be described below with reference to
The connecting conductor 56 is disposed on the surface of the insulating layer 44a, and is a linear conductor extending in the y-axis direction. As illustrated in
The radiation conductor 46 further includes the leg portion 46h. The leg portion 46h is formed by bending in the negative z-axis direction a protrusion extending from the midpoint of the short side of the radiation portion 46a in the positive y-axis direction toward the positive y-axis direction. The leg portion 46h is connected to the connecting conductor 56.
Thus, in the antenna 10i, there are two points of connection between the ground conductor 48 and the radiation conductor 46. Accordingly, the capacitance C2 and the inductance L2 of the radiation conductor 46 can be adjusted.
Embodiments consistent with the present disclosure are useful for an antenna, and, in particular, has an advantage in its suitability for providing a high degree of design flexibility.
| Number | Date | Country | Kind |
|---|---|---|---|
| 2009-162740 | Jul 2009 | JP | national |
The present application is a continuation of International Application No. PCT/JP2010/059113 filed May 28, 2010, which claims priority to Japanese Patent Application No. 2009-162740 filed Jul. 9, 2009, the entire contents of each of these applications being incorporated herein by reference in their entirety.
| Number | Date | Country | |
|---|---|---|---|
| Parent | PCT/JP2010/059113 | May 2010 | US |
| Child | 13344243 | US |
