This application claims the benefit under 35 U.S.C. §119(a) of a Korean Patent Application No. 10-2008-0094291, filed on Sep. 25, 2008, in the Korean Intellectual Property Office, the disclosure of which is incorporated herein by reference in its entirety for all purposes.
1. Field
The following description relates to a method and apparatus for transmitting and/or receiving a signal in a multiple input multiple output (MEMO) system.
2. Description of the Related Art
Multiple antennas may be a key for a next generation mobile communication system to improve the throughput without increasing frequency resources. A gain using the multiple antennas may be classified into a diversity gain and a multiplexing gain.
The diversity gain may improve the reliability of a transmission signal and the error performance of a communication system. The multiplexing gain may increase the data transmission rate.
Generally, conventional technologies associated with the multiple antennas may have either the diversity gain or the multiplexing gain. For a space-time block coding (STBC) technology using the multiple antennas, at least two transmit antennas may be needed to reduce the data transmission rate.
In one general aspect, there is provided a method and system for transmitting and receiving a signal that have both a diversity gain and a multiplexing gain regardless of a number of transmit antennas.
In another general aspect, there is provided a multiple antenna multiplexing technology having a transmit diversity gain that is applicable to a multiple antenna technology of a multi-node collaborating technology where network constituent elements collaborate with each other.
In still another general aspect, there is provided a signal transmission apparatus including an input unit to receive NT symbol streams in parallel, where NT is an integer greater than or equal to 2, a multiple signal generating unit to generate NT multiple signals having a transmit diversity gain by applying a cyclic delay of a different pattern to each of the NT symbol streams, and a multiple signal providing unit to provide the generated NT multiple signals to NT transmit antennas, respectively, based on a transmit antenna index of each of the NT transmit antennas.
The input unit may convert a phase of each of the NT symbol streams to provide the NT symbol streams with the converted phase to the multiple signal generating unit.
The input unit may convert a phase of each of the NT symbol streams to satisfy a condition that Δp,i−Δp-1,i>L and a condition that |det(D(k))| has a value of ‘1’. The multiple signal generating unit may apply the cyclic delay of the different pattern to each of the NT symbol streams, provided from the input unit, to satisfy the condition of Δp,i−Δp-1,i>L and the condition that |det(D(k))| has the value of ‘1’. Here, Δp,i denotes a cyclic delay value to be applied to an ith symbol stream that is transmitted via a pth transmit antenna, L denotes a length of a channel response, |det(D(k))| denotes an absolute value of a D(k) matrix determinant corresponding to a phase rotation and the cyclic delay.
The multiple signal generating unit may include a cyclic delay pattern application unit to apply the cyclic delay of the different pattern to each of the NT symbol streams, an adder to add up NT outputs of the cyclic delay pattern application unit, and a power level adjustment unit to adjust a power level of a signal that is output from the adder.
The cyclic delay of the different pattern may be expressed by Δi(p−NT)N
Here, Δi denotes a cyclic delay value, i denotes a symbol stream index, and p denotes a transmit antenna index where p=1, 2, . . . , NT.
The power level adjustment unit may adjust the power level of the signal to be 1/√{square root over (NT)}.
The cyclic delay of the different pattern may be expressed by Δi(p−NT)N
The cyclic delay value Δi may be set to be greater than a length of a channel response.
In yet another general aspect, there is provided a signal reception apparatus including NR receive antennas, each to receive a multiple signal having a transmit diversity gain, where NR is an integer greater than or equal to 2, and a cyclic prefix (CP) removal unit to remove a CP in the multiple signals having the transmit diversity gain. The multiple signals having the transmit diversity gain may be generated by applying, by a transmission device, a cyclic delay of a different pattern to each of NT symbol streams, and the generated NT multiple signals may be transmitted via NT transmit antennas, respectively, where NT is an integer greater than or equal to 2.
The channel impulse response between the NT transmit antennas and the NR receive antennas may be defined by the following equation:
where i denotes a symbol stream index, q denotes a receive antenna index, p denotes a transmit antenna index, Δi denotes a cyclic delay value, k denotes a sample index of a symbol stream, and N denotes a block size of the symbol stream.
The cyclic delay of the different pattern may be expressed by Δi(p−NT)N
The NT symbol streams with a converted phase may be defined by the following equation:
where {tilde over (s)}i,p denotes the NT symbol streams with the converted phase, si denotes the NT symbol streams, and i denotes a symbol stream index, and p denotes a transmit antenna index where p=1, 2, . . . , NT.
In a further general aspect, there is provided a signal transmission method including receiving NT symbol streams in parallel, where NT is an integer, greater than or equal to 2, generating NT cyclic delay signals by applying a cyclic delay of a different pattern to each of the NT symbol streams, generating NT multiple signals having a transmit diversity gain by coupling the NT cyclic delay signals, and providing the generated NT multiple signals to NT transmit antennas, respectively, based on a transmit antenna index of each of the NT transmit antennas.
The generating of the NT cyclic delay signals may comprise converting a phase of each of the NT symbol streams and applying the cyclic delay of the different pattern to each of the NT symbol streams with the converted phase.
The generating of the NT cyclic delay signals may comprise converting a phase of each of the NT symbol streams and applying the cyclic delay of the different pattern to each of the phase-converted NT symbol streams to be a condition that Δp,i−Δp-1,i>L and a condition that |det (D(k))| has a value of ‘1’,
where Δp,i denotes a cyclic delay value to be applied to an ith symbol stream that is transmitted via a pth transmit antenna, L denotes a length of a channel response, and |det(D(k))| denotes an absolute value of a D(k) matrix determinant corresponding to a phase rotation and the cyclic delay.
The generating of the NT multiple signals may comprise adding up the NT cyclic delay signals to output an addition signal, and adjusting a power level of the addition signal.
The cyclic delay of the different pattern may be expressed by Δi(p−NT)N
The power level of the addition signal may be adjusted to be 1/√{square root over (NT )}. The cyclic delay value Δi may be set to be greater than a length of a channel response.
Other features will become apparent to those skilled in the art from the following detailed description, which, taken in conjunction with the attached drawings, discloses exemplary embodiments.
Throughout the drawings and the detailed description, unless otherwise described, the same drawing reference numerals will be understood to refer to the same elements, features, and structures. The elements may be exaggerated for clarity and convenience.
The following detailed description is provided to assist the reader in gaining a comprehensive understanding of the methods, apparatuses and/or systems described herein. Accordingly, various changes, modifications, and equivalents of the systems, apparatuses and/or methods described herein will be suggested to those of ordinary skill in the art. Also, descriptions of well-known functions and constructions are omitted to increase clarity and conciseness.
A method and apparatus for transmitting and/or receiving a signal according to example(s) disclosed herein may combine a multi-carrier scheme and a single-carrier scheme that may be used as a mobile communication radio access technology, so as to simultaneously improve an error correction performance and a data transmission rate. The method and apparatus may be used as a key multiple antenna technology for a multi-node collaborating communication. The method and apparatus may be applicable to a multiple input multiple output (MIMO) system having NT transmit antennas and NR receive antennas. Here, NT and NR are integers greater than or equal to 2.
As illustrated in
In the case of a MIMO system using an orthogonal frequency division multiplexing (OFDM) technology, the transmission end 100 may further include an inverse fast Fourier transforming (IFFT) unit 109.
The S/P converter 101 may convert a serial-input information bit to NT parallel streams.
The encoding unit 103 may include NT encoders to encode the NT parallel streams, respectively. The interleaving unit 105 may include NT interleavers to interleave the encoded NT parallel streams, respectively.
The symbol mapping unit 107 may include NT symbol mappers to perform symbol mapping for output signals of the interleaving unit 105, respectively, according to a modulation scheme.
Hereinafter, an output signal {si(n)}n=0N-1 of the symbol mapping unit 107 or the IFFT unit 109 may be referred to as a symbol stream.
A single symbol stream may be expressed by NT layers.
The cyclic delay 111 may apply a cyclic delay of a different pattern to each of NT symbol streams to generate NT multiple signals Xp having a transmit diversity gain. The generated NT multiple signals Xp may be provided to transmit antennas included in the antenna unit 115 via NT CP inserters of the CP insertion unit 113, respectively, based on a transmit antenna index p. Here, p=1, 2, . . . , NT.
The CP insertion unit 113 may insert a CP into each of the NT multiple signals Xp to provide the NT multiple signals Xp with the inserted CP to the antenna unit 115.
The signal transmission apparatus 200 may be applicable to the cyclic delay 111 of
As illustrated in
The input unit 201 may receive NT symbol streams in parallel. Here, NT is an integer greater than or equal to 2.
The multiple signal generating unit 203 may apply a cyclic delay of a different pattern to each of the NT symbol streams that are provided from the input unit 201 to generate NT multiple signals Xp having a transmit diversity gain.
The multiple signal providing unit 205 may provide the NT multiple signals Xp to NT transmit antennas, respectively, based on a transmit antenna index p. Here, p=1, 2, . . . , NT.
For example, the multiple signal providing unit 205 may provide a pth multiple signal to a pth transmit antenna. The pth multiple signal may be inserted with a pth CP and then be provided to the pth transmit antenna.
The cyclic delay of the different pattern may be expressed by Δi(p−NT)N
The cyclic delay value Δi may be set to be greater than a length L of a channel response.
Accordingly, the multiplex signal xp=[xp(0), xp(1), . . . , xp(N-1)] may be obtained from the symbol stream {si(n)}n=0N-1. Here, p and I=1, . . . , NT. In the case of the multiple signal Xp, a symbol stream corresponding to each layer may have a transmit diversity gain and may also be decoupled from a reception end, which may be expressed by the following Equation 1:
where i denotes the symbol stream index, p denotes the transmit antenna index, Δi denotes the cyclic delay value, k denotes a time domain sample index of the symbol stream, and N denotes a block size of the symbol stream.
The input unit 201 may convert a phase of each of the NT symbol streams Si to provide the NT symbol streams {tilde over (s)}i,p with the converted phase to the multiple signal generating unit 203. The NT symbol streams {tilde over (s)}i,p with the converted phase may be expressed by the following Equation 2:
When using the symbol streams {tilde over (s)}i,p with the converted phase, a singularity of an equivalent MIMO channel may be avoided.
The multiple signal Xp with the inserted CP may be transmitted via a radio channel that is modeled via an Lth order finite impulse response (FIR) filter. In this case, a time domain channel impulse response (CIR) between a pth transmit antenna and a qth receive antenna may be expressed by the following Equation 3:
h
q,p
=[h
q,p(0), hq,p(1), . . . , hq,p(L), 0, . . . , 0]T (3).
A frequency domain CIR may be obtained by performing a fast Fourier transform (FFT) for the time domain CIR, and may be expressed by the following Equation 4:
λq,p=[λq,p(0), λq,p(1), . . . , λq,p(N−1]T (4).
A virtual channel response between an ith transmit layer and the qth receive antenna may be expressed by the following Equation 5:
Referring to the above Equation 3 through Equation 5, through a transmission signal mapping using the cyclic delay of the different pattern, it may be understood that the symbol stream {si(n)}n=0N-1 corresponding to each layer has passed through a virtual MIMO channel hq,ieq.
Where the cyclic delay value Δi is greater than the length L of the channel response, each equivalent channel tap hq,ieq(k) may not include a sum of time domain channel taps. Accordingly, a frequency selectivity of each equivalent channel may increase whereby a maximum transmit diversity gain may be obtained.
Zeroth subchannels of all the ith layers where i=1, . . . , NT may have the same frequency domain channel response, as given by the following Equation 6:
Accordingly, a MIMO channel corresponding to the Zeroth subchannel may be a singular matrix and the reception end may not detect a signal of each layer. The above singularity problem of the equivalent MEMO channel may be solved by changing a phase of a symbol stream corresponding to each layer, as described above.
The multiplex signal Xp obtained from the phase-converted symbol stream may be expressed by the following Equation 7:
The virtual channel response between the ith transmit layer and the qth receive antenna may be expressed by the following Equation 8:
The multiple signal generating unit 300 may be applicable to the multiple signal generating unit 203 of
As illustrated in
The adder 303 may add up NT outputs of the cyclic delay pattern application unit 301. The power level adjustment unit 305 may adjust a power level of a signal, output from the adder 303, to be 1/√{square root over (NT)}.
Accordingly, the final signal that is output by the multiple signal generating unit 300 may be expressed by the above Equation 7.
Here, the cyclic delay value Δi may be variable and may also be set to be greater than the length L of the channel response.
As illustrated in
In the case of a MIMO system using a single carrier with frequency domain equalization (SC-FDE), the reception end 400 may further include an IFFT unit 411.
Referring to
The multiple signal having the transmit diversity gain may be generated by applying a cyclic delay Δi(p−NT)N
The FFT unit 405 may convert a time domain signal to a frequency domain signal.
The interference cancellation and decoupling layer ‘i’ 407 may cancel multiple signal interference in the frequency domain signal and decouple a received signal in which the multiple signal interference is cancelled, for each layer.
The equalizer 409 may perform a channel equalization. The MAP detector 413 may perform a MAP detection using the channel equalization result.
The de-interleaver 415 may be a configuration corresponding to an interleaver of a transmission end and thus may perform de-interleaving.
The MAP decoder 417 may decode an output of the de-interleaver 415. The adder 419 may add up an output of the MAP decoder 417 and the output of the de-interleaver 415 to provide an output of the adder 419 to the interleaver 421.
The interleaver 421 may interleave an input signal to provide the interleaved input signal to the soft symbol mapper 423. The soft symbol mapper 423 may provide a mapped signal to the interference cancellation and decoupling layer ‘i’ 407 so that an iterative detection may be performed.
In some implementations, the above method and apparatus for transmitting and/or receiving a signal may have a compatibility to be applicable to a general MIMO signal detection technology.
<MIMO Equalization and Detection>
In a reception end, a received signal r in which a CP is cancelled may be expressed by the following Equation 9:
r=H
eq
s+w (9).
Here, may be expressed by the following Equation 10:
Hq,ieqCircN[hq,ieq(0), hq,ieq(1), hq,ieq(N−1)] denotes an N×N circulant channel matrix between the ith transmit layer having [hq,ieq(0), hq,ieq(1), hq,ieq(N−1)] as a first column vector and the qth receive antenna.
The received signal r is a time domain signal and may be converted to a frequency domain signal through FFT. The frequency domain signal may be expressed by the following Equation 11:
R=D
F
r=Λ
eq
S+W
DF=IN
Λeq=DFHeqDF−1 (11),
where F denotes an N×N FFT matrix and denotes a Kronecker product.
<MIMO SC-FDE>
The equalizer 409 may include a per-ton minimum mean square error (MMSE) equalizer. The equalizer 409 may minimize E{|si(n)−ŝi(n)|2} using the received signal r and a priori information to obtain a frequency domain estimate {Ŝi(k)}k=0N-1 of a transmission signal. The a priori information may be expressed by the following Equation 12:
{LEA(ci(j))}j=02N-1=πi({LD(ci(u))}u=02N-1) (12),
where LD(ci(u)) denotes an extrinsic log-likelihood ratio (LLR) of Ci(U) and ci(u) is a coded bit of the ith transmit layer that is obtained by the MAP detector 413 in a previous iteration.
From {LEA(ci(j))}j=02N-1, a mean vector
In this case, where co-antenna interference (CAI) is cancelled in the received signal, the received signal may be expressed by the following Equation 14:
Here, yi may be expressed by the following Equation 15:
yi=[y1,iT, . . . , yN
Also, Hpeq may be expressed by the following Equation 16:
Hpeq=[H1,peqT, . . . , Hn
Canceling of the CAI may be performed in a frequency domain.
The time domain estimate {si(n)}n=0N-1 may be expressed by the following Equation 17:
ŝ
i(n)=ginHyi+ci(n) (17).
(.)H denotes a complex conjugate transpose. According to an MMSE criterion, an NRN×1 weight vector and ci(n) may be expressed by the following Equation 18:
The NRN×1 weight vector may also be expressed by the following Equation 19:
gin=[g1,inT, . . . , gN
V=Diag(V1, . . . , VN
ŝi(n) may be affected by LEA(ci (2n)), and LEA(ci(2n+1)) via
Due to the matrix V, a calculation process of a frequency domain weight vector may require a high calculation complexity. In order to reduce the calculation complexity, Vp may be approximated to
gin may be expressed by the following Equation 20:
Since Hq,peq is a circulant matrix, it can be known from the above Equation 20 that ginT is an n-point right-shift vector of gi0T. Accordingly, where the affect of
A frequency domain weight vector Gi0 of the per-tone MMSE equalizer may be expressed by the following Equation 22:
Here, Λpeq may be expressed by the following Equation 23:
Λpeq=[Λ1,peqT, . . . , ΛN
Also, Λq,peq may be expressed by the following Equation 24:
Λq,peq=FHq,peqFH (24)
Here, Λq,peq denotes a diagonal matrix having λq,peq(k) as (k, k)th entry. A frequency domain estimate ŝi may be expressed by the following Equation 25:
where (.)* denotes a complex conjugate.
Referring to the above Equation 22 and Equation 25, a per-tone weight vector and a frequency domain estimate corresponding to a kth subchannel may be expressed by the following Equation 26 and Equation 27, respectively:
G
i
0(k)=(Λeq(k)Q(Λeq(k))H+σw2IN
Ŝ
i(k)=(Gi0(k))HYi(k)−((Gi0(k))HΛieq(k)μŝ
Here, Λeq(k) denotes an NR×NT MIMO channel corresponding to the kth subchannel, and Λieq(k) denotes an ith transmit layer of Λeq(k). μŝ
and Q denotes Diag(v1, v2, . . . , vN
where μŝ
The LLR value {LDA(ci(u)}=u=02N-1=πi−1({LE(ci(j))}j=02N-1) may be input into the MAP decoder 417. The MAP decoder 417 may calculate extrinsic information associated with a code bit and a decode bit. The extrinsic information LD(ci(u)) associated with the coded bit may be used as a priori information. Since a signal-to-noise ratio (SNR) of each layer has the same value, the aforementioned MIMO SC-FDE process may be iteratively performed through an unordered successive interference cancellation (SIC) process.
<MIMO OFDM>
OFDM may perform a per-tone equalization in a frequency domain. The per-tone weight vector and the frequency domain estimate corresponding to the kth subchannel may be expressed by the following Equation 29 and Equation 30, respectively:
G
i(k)=(Λieq(k)Vi(k)(Λieq(k))H+σw2IN
Ŝ
i(k)=(Gi(k))HYi(k), i=1, 2, . . . , NT (30).
Here, vi(k) is Diag(v0(k), v1(k), vN
Extrinsic LLRs LE(ci(2k)) and LE(ci(2k+1)) may be expressed by the following Equation 31:
Here, μŜ
Referring to
In some implementations, the operation S505 may include adding up NT cyclic delay signals to output an addition signal, and adjusting a power level of the addition signal.
<Another Example of Phase Rotation and Cyclic Delay>
Hereinafter, an example of applying the phase rotation and the cyclic delay in order to simultaneously obtain both a diversity gain and a multiplexing gain, regardless of a number of transmit antennas, will be described.
In order to simultaneously obtain both the diversity gain and the multiplexing gain, a signal transmission apparatus may apply a different phase rotation and a cyclic delay to each of NT symbol streams and may simultaneously transmit, via NT transmit antennas, the NT symbol streams where the different phase rotation and the cyclic delay are applied. Since the different phase rotation and the cyclic delay are applied to each of the NT symbol streams, it is possible to avoid a singularity of a frequency domain channel matrix where a signal reception apparatus separates each of the NT symbol streams. The phase rotation may be performed by the signal input unit 201 of
Here, θp,i and Δp,i denote a phase rotation value and a cyclic delay value, respectively, to be applied to an ith symbol stream that is transmitted via a pth transmit antenna. A frequency domain signal corresponding to a kth subchannel that is transmitted via a transmit antenna may be expressed by the following Equation 32:
Accordingly, a frequency domain received signal of the kth subchannel in the signal reception apparatus may be expressed by the following Equation 33:
R(k)=Λ(k)D(k)S(k)+W(k)=Λeq(k)S(k)+W(k) (33).
As described above, in order to obtain a maximum multi-path diversity gain, a cyclic delay value Δi may be set to have a value greater than a length of a channel response. According to the above Equation 32, Δp,i may be set to Δp,i−Δp-1,i>L in order to obtain the maximum multi-path diversity gain.
In order to avoid a singularity of an equivalent MIMO channel matrix Δeq(k) corresponding to the kth subchannel, a determinant value of Δeq(k) may have no need to be zero. Determinant det (AB) of an AB matrix that is a multiplication of two matrices A and B may be the same as a multiplication of determinants of each of the matrices A and B. Thus, det(AB)=det(A)det(B). Accordingly, determinant det(Δeq(k)) of Δeq(k)=det(Δ(k))det(D(k). In order to avoid the singularity of Δeq(k), there is a need for a pattern of the phase rotation value and a pattern of the cyclic delay value that satisfy a condition that Δp,i−Δp-1,i>L and also enable |det(D(k))| to have a maximum value.
A sum of absolute square values of elements of a D(k) matrix corresponding to the phase rotation and the cyclic delay may be a square value of Frobenius norm, and may be expressed by the following Equation 34:
Since the square value of Frobenius norm is the same as a sum of square values of singular values of the D(k)matrix. The above Equation 34 may be expressed by the following Equation 35:
Here, σi denotes the singular values of the D(k) matrix. Since an arithmetic mean has a value greater than or equal to a geometric mean at all times, the square values of the singular values of the D(k) matrix may have a relationship as given by the following Equation 36:
Also, since an absolute value |det (D(k))| of the D(k) matrix determinant is the same as a multiplication of singular values of the D(k) matrix, a maximum value of the absolute value |det(D(k)) | may be expressed, using the above Equation 36, by the following Equation 37.
Specifically, it is proved that |det(D(k))| may have a value of maximum ‘1’. Accordingly, the signal transmission apparatus may use the pattern of the phase rotation value and the pattern of the cyclic delay value that satisfy the condition that Δp,i−Δp-1,i>L and the condition that |det(D(k))| has a value of ‘1’. Where the pattern of the phase rotation value and the pattern of the cyclic delay value that satisfy the above conditions are used, it may be possible to enhance a reliability for an initial estimate in an Eb/N0 versus block error rate performance.
For example, where a number of transmit antennas is two, the pattern of the phase rotation value and the pattern of the cyclic delay value that satisfy the above conditions may be obtained as follows. A determinant of the D(k) matrix may be expressed by the following Equation 38:
Accordingly, it is possible to use any of the pattern of the phase rotation value and the pattern of the cyclic devalue value that satisfy a condition that Δ2,1−Δ1,1)=(Δ2,2−Δ1,2)>L and a condition that (θ1,1+θ2,2)−(θ1,2+θ2,1)=π. Where the number of transmit antennas is greater than two, it is possible to obtain the pattern of the phase rotation value and the pattern of the cyclic delay value using a similar scheme.
In the protocol example of
Referring to the above Equation 39, a second layer may not obtain a spatial diversity gain. In the protocol example of
Accordingly, in the protocol example of
As shown in
According to example(s) described above, a method and apparatus for transmitting and/or receiving a signal may be provided that simultaneously obtains both a multiplexing gain and a transmit diversity gain. Both the diversity gain and the multiplexing gain may be obtained regardless of a number of transmit antennas. Here, since a reception end does not require an ordering process for an SIC process, a calculation complexity for each iteration may be reduced in comparison to a conventional H-Blast scheme.
Also, according to example(s) described above, it is possible to simultaneously provide both a multiplexing gain and a transmit diversity gain by applying a different cyclic delay diversity pattern to each independent symbol stream and by transmitting symbol streams with the applied different cyclic delay diversity pattern via all the transmit antennas.
Also, according to example(s) described above, since an equivalent SNR value of a reception end of a layer corresponding to each of a plurality of symbol streams is the same, an ordering process may not be required for an SIC process.
The methods described above may be recorded, stored, or fixed in one or more computer-readable media that includes program instructions to be implemented by a computer to cause a processor to execute or perform the program instructions. The media may also include, alone or in combination with the program instructions, data files, data structures, and the like. Examples of computer-readable media include magnetic media, such as hard disks, floppy disks, and magnetic tape; optical media such as CD ROM disks and DVDs; magneto-optical media, such as optical disks; and hardware devices that are specially configured to store and perform program instructions, such as read-only memory (ROM), random access memory (RAM), flash memory, and the like. Examples of program instructions include machine code, such as produced by a compiler, and files containing higher level code that may be executed by the computer using an interpreter. The described hardware devices may be configured to act as one or more software modules in order to perform the operations and methods described above, or vice versa.
A number of exemplary embodiments have been described above. Nevertheless, it will be understood that various modifications may be made. For example, suitable results may be achieved if the described techniques are performed in a different order and/or if components in a described system, architecture, device, or circuit are combined in a different manner and/or replaced or supplemented by other components or their equivalents. Accordingly, other implementations are within the scope of the following claims.
Number | Date | Country | Kind |
---|---|---|---|
10-2008-0094291 | Sep 2008 | KR | national |