Patent Application
20080170646
1. Field
This invention relates generally to electronic devices using automatic gain control, and more specifically to analog circuits having cascaded controllable amplifying (gain) stages that are controlled with an automatic gain control system.
2. Related Art
In many applications of electronic circuits, a signal can benefit from signal conditioning such as by using an automatic gain control (AGC) system. In particular, for example, the presence of noise signals and other interference signals can deteriorate the quality of a desired signal. This is especially problematic in sensitive analog circuits having high amplification (or gain), such as found in analog receiver circuits. An AGC system is typically used to try to reduce the gain of unwanted signals while maintaining or increasing the gain of the desired signal.
A measure of a signal quality is sometimes stated as a Signal to Noise Ratio (SNR). Due to high amplification (gain) of desired signals in a particular electronic circuit design, such as by using a series of cascaded gain stages in a line-up, the high overall amplification also increases unwanted signals, e.g., noise and interference signals, to high levels. This overall gain increase can detrimentally affect the performance of the circuit, such as by causing clipping of desired signal at an output of the circuit. An AGC system can be used to adjust the overall gain of the amplification (or gain) stages to try to reduce the levels of unwanted signals while still maintaining or increasing the level of the desired signal. However, selectivity in an analog receiver circuit call create some problems for the AGC system as discussed below.
For receivers with high analog selectivity, AGC switch points can occur much earlier than necessary. These early switch points detrimentally impact the on-channel SNR regardless of whether noise or interference signals are present or whether noise or interference signals are strong enough to create clipping of desired signal.
Another receiver performance metric used, for example, for high-speed downlink packet access (HSDPA or 3.5G) transceivers is the receiver error vector magnitude (EVM) performance under both on-channel signal-only and interferer test cases. HSDPA transceivers require a receiver EVM of around 5% to achieve desired network throughput at signal levels of −60 dBm and higher. Receiver EVM performance is typically required to be 5% under on-channel signal-only test cases at antenna signal levels of −60 dBm and higher. Receiver EVM performance is typically required to be approximately 10% for adjacent channel interferer test cases.
To achieve this type of receiver EVM performance, an AGC in such receivers should maintain loop bandwidths of 1-kHz or less. Maintenance of such narrow bandwidths causes the receiver to be unable to quickly track out large gain errors introduced in the receiver whenever the AGC system alters gain settings. This inability leads to degraded receiver performance, such as under fading channel conditions. Running the AGC systems continuously in a medium or high bandwidth mode of operation significantly degrades receiver EVM performance to an unacceptably large degree of more than 15%.
Known receivers lack accurate power detectors at the various analog cascaded gain stages. This forces the AGC to sacrifice on-channel SNR and EVM performance by switching out gain earlier than necessary to prevent noise and interfering signals from compressing signal and/or creating IMD (Inter-modulation Distortion) products.
The present invention is illustrated by way of example and is not limited by the accompanying figures, in which like references indicate similar elements. Elements in the figures are illustrated for simplicity and clarity and have not necessarily been drawn to scale.
The terms “a” or “an”, as used herein, are defined as one or more than one. The term plurality, as used herein, is defined as two or more than two. The term another, as used herein, is defined as at least a second or more. The terms including and/or having, as used herein, are defined as comprising (i.e., open language). The term coupled, as used herein, is defined as connected, although not necessarily directly, and not necessarily mechanically. The terms program, software application, and the like as used herein, are defined as a sequence of instructions designed for execution on a computer system. A program, computer program, or software application may include a subroutine, a function, a procedure, an object method, an object implementation, an executable application, a source code, an object code, and/or other sequence of instructions designed for execution on a computer system.
The present invention, according to an embodiment, overcomes problems with the prior art by accurately estimating the signal at every analog gain stage in an analog line-up and without using analog wideband detectors. In addition, digital equalizers provide accurate estimates of the signal present at each cascaded analog gain stage in a line-up to facilitate a method for a digital AGC to set optimal gain settings for the cascaded analog gain stages. This optimal gain setting method allows for all gain stages to be updated simultaneously. In addition, because the gain steps for each gain stage may not be all equal, one embodiment of the invention provides a dynamic method to set hysteresis and thresholds without complex programming and phasing requirements.
One embodiment of the present invention provides an RF receiver that benefits from optimized SNR performance regardless of whether noise and/or interference signals are present or not. This is a factor to be considered in modern data communication systems using high speed wireless protocols such as for high-speed downlink packet access (HSDPA or 3.5G) transceivers.
Starting at an antenna 104, a wirelessly received signal is coupled from the antenna 104 to a receiver analog line-up 102 that typically comprises an RF front-end section of the receiver 100 that includes an RF duplexer filter (not shown) and one or more low noise amplifiers (LNA) and filters that amplify and condition the received signal from the antenna 104. In this example, these LNA's and filters comprise controllable variable gain amplifier stages 120, 124 and 128, and respective associated filters 122, 126 and 130. The output 108 of the receiver analog line-up 102, according to one embodiment of the present invention, is coupled to an analog-to-digital (A/D) converter 110. The output of the A/D converter 110 provides a digital signal that represents an analog signal sampled by the A/D converter 110 at the output 108 of the receiver analog line-up 102.
The digital signal from the A/D converter 110 is coupled to a digital AGC system 112 that can thereby monitor the signal present at the output 108 of the receiver analog line-up 102, and also can monitor the output of each and every controllable variable gain amplifier stages 120, 124 and 128, and respective associated filters 122, 126 and 130, of the receiver analog line-up 102, as will be discussed in more detail below.
The digital AGC system 112 includes an nth digital equalizer filter fn(t) 132 matched to an nth analog filter 130 in the analog line-up 102. The nth digital equalizer filter fn(t) 132 receives an input digital signal, processes the digital signal, and provides an output digital signal that has been adjusted by substantially the inverse of the filter function of the nth analog filter 130. Thereby, the nth digital equalizer filter fn(t) 132 equalizes the digital signal to substantially cancel the effect of the nth analog filter 130. The resulting output digital signal would be representative of the analog signal at the input of the nth analog filter 130 in the analog line-up 102. This analog signal is also at the output of the nth variable gain stage 128 in the analog line-up 102.
There are many different methods to determine the response of a digital equalizer filter, i.e., the inverse response of an analog filter hn(t) over a desired bandwidth. One can create the filter by using a time domain Minimum Mean Square Error Estimation (MMSE) method. This method creates a filter by minimizing the mean square error between the received signal and a test signal. On example of this type of method is a Recursive Least Squares method which creates a filter using the following equation: fn=R−1g(n). R is the autocorrelation function and g(n) is the cross-correlation function. Alternatively, the response of a digital equalizer filter can be done in the frequency domain as follows:
F
n(t)=[IFFT(Hn−1(f)Kn(f)]/gn(mn)
Note that, in some applications, the equalizer filter function can be simplified by matching it only to the magnitude response of the corresponding analog filter without concern for group delay. This is so because in these applications we would only be interested in the power response over the equalized band and we would not care about the group delay. Note that IFFT stands for inverse Fourier transform function. Kn(f) represents a desired filter response within a bandwidth. Kn(f) could be 1, or it could include, for example, an ideal low pass filter response at a bandwidth of interest. The frequency response of Kn(f) can include a brick wall filter and/or a high pass notch filter to remove all unwanted frequencies that are not relevant for the estimate. Hn−1(f) represents 1 divided by the function of Hn(f) which represents the inverse filter function in the frequency domain for the gain stage of interest. Based on the particular implementation, the compensation for the gain, gn(mn), of a particular gain stage, can occur in the function of the digital equalizer filter, Fn(t), or in the function of the digital estimation device, detn, or in any other place or component in the system 112.
The digital AGC system 112 also includes an nth digital estimation device detn 134 corresponding to an output of the nth variable gain stage 128 in the analog line-up 102. The digital estimation device detn 134 estimates, in one example of the digital AGC system 112, the output power at the output of the corresponding nth variable gain stage 128. The digital estimation device detn 134 can alternatively estimate, in alternative examples of the digital AGC system 112, the peak power or the average power at the output of the corresponding nth variable gain stage 128. A digital estimation device detn 134 can include a digital estimator and also can include an averaging function. This detector can include a digital estimator. One method of creating a digital power estimate is to convert the input signal (the output of the filter fn) to a power (i.e., sum the square of the real and imaginary parts: Î2+Q̂2) then convert the power to a dB (i.e., take 10*log(power)) now we can pass this through a low pass filter (or a simple integrator) to average the power estimate. Note that this detector (detn) can be a power, peak, etc., estimator.
In similar fashion as discussed above, a second digital equalizer filter f2(t) 138 is matched to a second analog filter 126 in the analog line-up 102, and a first digital equalizer filter f1(t) 144 is matched to a first analog filter 122 in the analog line-up 102. The digital AGC system 112 also includes a second digital estimation device det2 140 corresponding to an output of the second variable gain stage 124 in the analog line-up 102, and a first digital estimation device det1 146 corresponding to an output of the first variable gain stage 120 in the analog line-up 102.
The respective outputs 136, 142, and 148, of the digital estimation devices detn 134, det2 140, and det1 146, are monitored by an AGC controller 152. The AGC controller 152 can he implemented as an AGC algorithm or process in a state machine, and/or in code executable by a processor, or in any other digital signal processing controller architecture as should be obvious to those of ordinary skill in the art in view of the present discussion. The AGC controller 152 will be discussed in more detail below with reference to
In this example, the AGC controller 152 has a number of digital control outputs 150, several such outputs 160, 162 and 164, being shown, to digitally control the gain state of each and every one of the controllable variable gain amplifier stages 120, 124 and 128 in the analog line-up 102. Each of the controllable variable gain amplifier stages 120, 124 and 128, is operable at one gain of a plurality of gains. A digital value is provided from the AGC controller 152 to any one of the controllable variable gain amplifier stages 120, 124 and 128, to select a gain state for the particular controllable gain stage. Within a short time (e.g., typically in microseconds) of having the new state selected at its input, the particular controllable gain stage will adjust its gain to the new gain state. In this way, the digital AGC 112 controls the various gains of the various controllable gain amplifier stages 120, 124 and 128 in the analog line-up 102.
A dynamic hysteresis controller 156, in this example, uses a pre-defined set of input threshold values 158 to provide, as an input to the AGC controller 152, the one or more digital threshold values 154 used to determine the gain states to set for the controllable variable gain amplifier stages 120, 124 and 128 in the analog line-up 102. The dynamic hysteresis controller 156 will be discussed in more detail below.
To determine the upper and lower thresholds to use for a particular application one should consider the various gain states of each controllable gain analog amplifier stage. Also, because the gain step sizes can vary, one alternative embodiment of the invention utilizes a dynamic method for selecting hysteresis for setting thresholds for the various gain states of each controllable gain analog amplifier stage.
If a controllable gain analog amplifier stage has variable step sizes, one can either define a hysteresis for each step or one can use the following method to create a scaled version of the hysteresis that is a function of the minimum step size. This method significantly reduces the amount of hardware and software used in a hysteresis controller 156 to provide the thresholds to the AGC controller 152 because separate upper and lower hysteresis programmable registers for each non-uniform gain state in the analog lineup 102 are not needed. For each gain stage gn:
stepsizeUn=(gn(m−1)−gn(m))
stepsizeLn=(gn(m)−gn(m+1))
upperhysteresis=(stepsizeUn−minstepn)scalen+minhysteresisn
lowerhysteresis=(stepsizeLn−minstepn)scalen+minhysteresisn
The above method can be extended to all m gain states, thus creating a precalculated table of upper and lower hysteresis using the following equations. (Note that these minhysteresis, upperhysteresis, and lowerhysteresis, are all two-sided hysteresis. To have the actual upper and lower thresholds we would use upperhysteresis/2 and lowerhysteresis/2).
Upperhysteresis(m)=(stepsizeUn(m)−minstepn)scalen+minhysteresisn
Lowerhysteresis(m)=(stepsizeLn(m)−minstepn)scalen+minhysteresisn
where stepsizeU/Ln(m) indicates the upper and lower step size for the mth gain state of gain stage gn.
In this example, the AGC controller 152, at step 202, compares the output of the first digital estimation device det1 146 to a predefined upper threshold and to a predefined lower threshold that define an operating range for the desired gain of the first variable gain stage 120 in the analog line-up 102. For example, these upper and lower thresholds could be set to maintain the output of the first variable gain stage 120 away from the 1-db compression points of the gain stage 120. The output of the first digital estimation device det1 146 represents an accurate estimate of the power of the analog signal at the output of the first variable gain stage 120 in the analog line-up 102. If the AGC controller 152, at step 202, determines that the digital signal at the output 148 of the first digital estimation device det1 146 is greater than or equal to the predefined upper threshold, then the AGC controller 152, at step 204, decreases the gain of the first variable gain stage 120. The AGC controller 152, at step 203, also decreases the digital estimate at the output 142 of the second digital estimation device det2 140 by the reduction of gain of the first variable gain stage 120. This immediate adjustment down of the digital estimate at the output 142 of the second digital estimation device det2 140 thereby immediately accounts at the next (second) gain stage 124 for the reduced gain being applied to the previous (first) gain stage 120. This quick adjustment being applied to the next (second) gain stage 124 (and in similar fashion to all of the subsequent gain stages) reduces the amount of delay for all adjustments of gain to take effect and reach a steady state gain at all of the gain stages 120, 124 and 128, in the analog line-up 102.
Alternatively, if the AGC controller 152, at step 202, determines that the digital signal at the output 148 of the first digital estimation device det1 146 is less than the predefined lower threshold, then the AGC controller 152, at step 206, increases the gain of the first variable gain stage 120. The AGC controller 152, at step 205, also increases the digital estimate at the output 142 of the second digital estimation device det2 140 by the increase of gain of the first variable gain stage 120. This immediate adjustment up of the digital estimate at the output 142 of the second digital estimation device det2 140 thereby immediately accounts at the next (second) gain stage 124 for the increased gain being applied to the previous (first) gain stage 120. Note that the adjustment in gain, in steps 203 and 205, can be a function based on the nonlinear effects of the amplifier gn. It does not necessarily need to be the change in gain since when the device is in strong compression the gain is no longer linear. For simplicity, in this example, we assume it is linear and thus use the change in gain to update the next estimate an+1.
If the AGC controller 152, at step 202, determines that the digital signal at the output 148 of the first digital estimation device det1 146 is not greater than or equal to the predefined upper threshold and also is not less than the predefined lower threshold, then the AGC controller 152, immediately proceeds to the next comparison step 208.
The AGC controller 152, at step 208, compares the output 142 of the second digital estimation device det2 140 to a predefined upper threshold and to a predefined lower threshold that define an operating range for the desired gain of the second variable gain stage 124 in the analog line-up 102. As discussed above, these upper and lower thresholds could be set, for example, to maintain the output of the second variable gain stage 124 away from the 1 db compression points of the gain stage 124. The output 142 of the second digital estimation device det2 140 represents an accurate estimate of the power of the analog signal at the output of the second variable gain stage 124 in the analog line-up 102. If the AGC controller 152, at step 208, determines that the digital signal at the output 142 of the second digital estimation device det2 140 is greater than or equal to the predefined upper threshold, then the AGC controller 152, at step 210, decreases the gain of the second variable gain stage 124. In similar fashions as has already been discussed above, the AGC controller 152, at step 211, also decreases the digital estimate at the output of the next digital estimation device by the reduction of gain of the second variable gain stage 124. This immediate adjustment down of the digital estimate at the output of the next digital estimation device thereby immediately accounts at the next gain stage for the reduced gain being applied to the previous (second) gain stage 124. This quick adjustment being applied to the next gain stage (and in similar fashion to all of the subsequent gain stages up to the final gain stage n 128) reduces the amount of delay for all adjustments of gain to take effect and reach steady state at all of the gain stages 120, 124 and 128, in the analog line-up 102.
Continuing with the operational flow sequence, if the AGC controller 152, at step 208, determines that the digital signal at the output 142 of the second digital estimation device det2 140 is less than the predefined lower threshold, then the AGC controller 152, at step 212, increases the gain of the second variable gain stage 124. The AGC controller 152, at step 213, also increases the digital estimate at the output of the next digital estimation device by the increase of gain of the second variable gain stage 124. Similar to as has been discussed above, this immediate adjustment up of the digital estimate at the output of the next digital estimation device thereby immediately accounts at the next gain stage for the increased gain being applied to the previous (second) gain stage 124.
If the AGC controller 152, at step 208, determines that the digital signal at the output 142 of the second digital estimation device det2 140 is not greater than or equal to the predefined upper threshold and also is not less than or equal to the predefined lower threshold, then the AGC controller 152, immediately proceeds to the next comparison step.
The AGC controller 152 continues with this operational flow sequence comparing digital estimates to predefined upper and lower thresholds for the remaining outputs of the digital estimation devices, and possibly adjusting gain for respective controllable gain stages in the analog line-up 102, until the output 136 of the nth digital estimation device detn 134 is compared to predefined upper and lower thresholds and the gain of the nth variable gain stage 128 is possibly adjusted, at steps 214, 216 and 218.
After the AGC controller 152 does all of the series of comparisons and calculations to arrive at a set of digital control outputs 150, several such outputs 160, 162 and 164, being shown in
This continuous and autonomous process, as has been described above, allows real time monitoring and gain adjustment for each and every output of the controllable gain stages 120, 124 and 128, in the analog line-up 102. The digital AGC system 112 uses multiple equalized estimates with dynamic hysteresis that allow for more optimal gain settings in the analog line-up 102 and improved SNR because the signal power is estimated at each gain stage. This innovative wireless receiver 100 benefits from the novel digital AGC system 112 controlling the analog line-up 102 to maximize SNR (Signal to Noise Ratio) and EVM (Error Vector Magnitude) performance of the wireless receiver 100 with one or more cascaded analog filtering stages 120, 124 and 128. This applies to both on-channel and interference scenarios for a wireless receiver 100. Such an implementation can achieve >10-dB SNR and <10% EVM performance for high speed wireless communications such under the HSDPA (3.5G) protocol under all adjacent channel interference test cases.
Similar to the previous discussion with reference to
The digital AGC system 312 includes a digital equalizer filter f1(t) 344 matched to the first (and only) analog filter 322 in the analog line-up 302. The analog filter 322 comprises a three-pole analog low-pass filter. Accordingly, the digital equalizer filter f1(t) 344 provides a frequency response function that is substantially the inverse of the three-pole analog low-pass filter 322 in the analog line-up 302. Such inverse function of the digital equalizer filter f1(t) 344 is shown in
More specifically, the digital AGC system 312 includes the digital equalizer filter f1(t) 344 that is matched to the analog filter 322 in the analog line-up 302. The digital equalizer filter f1(t) 344 receives a digital signal from the A/D converter 310, processes the digital signal, and provides an output digital signal that has been adjusted by substantially the inverse of the filter function of the analog filter 322. Thereby, the digital equalizer filter f1(t) 344 equalizes the digital signal to substantially cancel the effect of the analog filter 322. The resulting output digital signal would be representative of the analog signal at the input of the analog filter 322 in the analog line-up 302. This analog signal is also at the output of the first controllable analog gain stage 320 in the analog line-up 302.
By utilizing the digital equalizer filter f1(t) 344 matched to the analog filter 322, the AGC controller 352 monitors the two digital estimates provided at the two outputs 348 and 342 of the two digital estimation devices det1 346 and det2 340, where these two digital estimates represent the analog signals present at the outputs 320 and 308 of the first controllable analog gain stage 320 and the second controllable analog gain stage 324. The AGC controller 352 utilizes the input threshold values 354 to make comparisons and possibly to increase or decrease the gain state of each of the first and second controllable analog gain stages 320 and 324 in the analog line-up 302. This comparison and possible gain adjustment process followed by the AGC controller 352 is similar to the discussion above of such a process with reference to
Additionally, the dynamic hysteresis controller 356, in this example, uses a pre-defined set of input threshold values 358 to provide, as an input to the AGC controller 352, the one or more digital threshold values 354 used to determine the gain states to set for the controllable variable gain amplifier stages 320 and 324 in the analog line-up 302. Methods useful for a hysteresis threshold setting process for the dynamic hysteresis controller 356 have been discussed above with reference to the dynamic hysteresis controller 156 of
It should be clear in view of the specific example discussed immediately above, and further in view of the more generic example discussed further above with reference to
Issues being addressed by this embodiment of the invention include the following: 1) Computing thresholds for the control of multiple gain stages when the gain stages have variable step size, thus allowing for nonlinear (non-monotonic) gain stages. 2) Incorporating scalable hysteresis at each threshold point (can be scaled in proportion to the individual step size). 3) Having control over the number and order of the steps being executed for each gain stage.
This embodiment of the invention allows multiple gain stages to work together to regulate the gain in a closed-loop AGC system. When controlling multiple variable gain stages, where each stage contains gain steps of varying size, it would be desirable to be able to limit the number of thresholds that have to be initialized, and allow the system to self-generate individual thresholds at spacings proportional to the non-monotonic step sizes. These thresholds would generate a lookup table against which the incoming detected power would be compared in order to determine the direction and magnitude of gain update in each stage.
Features of such an implementation include the following: 1) The gain steps within each stage are combined together to form one table containing gain stages at varying deltas. 2) The user can program the individual gains corresponding to each gain step. 3) The user can rank the states in the table such that none, some, or all of one gain stage's steps can be traversed before beginning to update the other stages. For an increasing input signal, this would allow a user or system designer to specify which gain stage should decrement first (for optimal SNR) and for how long before starting to decrement other stages. 4) Each state in the table is represented by one or both of an increment and decrement threshold to be applied to the incoming detected power. The table is created by concatenating the steps for each stage, in the order specified by the user in (3) above, so that the increment and decrement thresholds represent the combination of the stages. 5) Because each state has its own set of gains and increment/decrement steps, the user need only program three items: the initial threshold, the number of steps of one stage to traverse before the other stages starts changing, and the total number of steps to use for the other gain stages. Together, this allows for great freedom in organizing the gain structure of the system. The flow diagram in
In this exemplary implementation, there are two gain stages, Stage A and Stage B. With reference to
A depiction of an example increment and decrement series, for use with the flow diagram 600 of
In general,
staten
state(n+1)
where:
staten
staten
agc_thresh=the programmable threshold that is used to compute the individual thresholds in the concatenation of the steps from all the gain stages.
dn=a running cumulative total of the individual deltas representing the total gain change between Staten and State0 (thus, d0=0). Depending upon the user's selection of the number of states in Stage A to occur first, d1 would equal (d0+(gain0(StageA)−gain1(StageA)), if Stage A decrements first, or (d0+(gain0(StageB)−gain1(StageB)) if Stage B decrements first. The value of d2 then equals the sum of d1 plus the next delta.
hystn=the hysteresis for state transitions between n and (n−1).
An example is given below:
state0
state1
state1
In our exemplary implementation, Stage A has a fixed number (e.g., 8) of gain states with fixed but unequal gain values; thus, 7 deltas are available. For stage B, the user may select a variable number of states with programmable gain values to use in computing deltas (e.g., from 2 to 7 total states), so anywhere from 1 to 6 deltas could be used. For this example, this results in a total number of 7+6=13 deltas. For State A, the user can decide to have anywhere from 0 states first (i.e., all the states in Stage B occur first before Stage A starts updating) to all the Stage A states first. The states in Stage B occur together in series and are not broken up. The number of Stage B gain deltas that can be used in the dn equations will depend upon the user's choice of the total number of states to use for Stage B. The point at which Stage A gain deltas begin being included in the running total is determined by the user's selection of the number of states in Stage A to occur first. Once the total number of Stage A and B states are used up, further delta values are undefined and are not used.
In the exemplary embodiments discussed above, the entire portion of the wireless receiver 100 and 300, including the analog line-up 102 and 302, and the digital AGC system 112 and 312, circuitry is implemented and disposed on a circuit supporting substrate supporting the above mentioned circuits. An example of such a circuit supporting substrate implementing the portion of the wireless receiver 100 and 300 is a single integrated circuit that can be manufactured using complementary metal oxide semiconductor (CMOS) technology. However, it should be obvious to those of ordinary skill in the art in view of the present discussion that other circuit manufacturing technologies, or combination of technologies, can alternatively be used to implement alternative embodiments of the invention. In the exemplary embodiment, the operating frequency range of the receiver 102 and 302 is approximately 800-2000 MHz.
It should be understood that all circuitry described herein may be implemented either in silicon or another semiconductor material or alternatively by software code representation of silicon or another semiconductor material.
While the principles of the invention have been described above in connection with specific apparatus, it is to be clearly understood that this description is made only by way of example and not as a limitation on the scope of the invention. For instance, although the exemplary embodiment is for use with a zero IF receiver, the exemplary embodiment can also be used with a very low IF receiver, a low IF receiver or with any receiver. Additionally, the novel aspects of the digital AGC system can be applied to devices and circuits other than receiver circuits, as should be obvious to those of ordinary skill in the art in view of the present discussion.
Although the exemplary embodiment is shown on a single integrated circuit manufactured using CMOS technology, the invention can also be used on a single integrated circuit manufactured using other manufacturing technologies. Although the exemplary embodiment is shown on a single integrated circuit the invention, is equally applicable when portions of the embodiment are on more than one integrated circuit. Although the exemplary embodiment is shown on an integrated circuit, the invention is equally applicable when the embodiment is on a circuit in a form other than an integrated circuit.
Accordingly, the specification and figures are to be regarded in an illustrative rather than in a restrictive sense, and all such modifications are intended to be included within the scope of the present invention. Any benefits, advantages, or solutions to problems that are described herein with regard to specific embodiments are not intended to be construed as a critical, required, or essential feature or element of any or all the claims.
Unless stated otherwise, terms such as “first” and “second” are used to arbitrarily distinguish between the elements such terms describe. Thus, these terms are not necessarily intended to indicate temporal or other prioritization of such elements.
