BACKGROUND OF THE INVENTION
1. Field of the Invention
The invention relates to a multi-lamps LCD back-light control circuit. More particularly, the invention provides a control circuit that can simplify the circuitry of dimensionally larger LCD devices.
2. Description of the Related Art
Compared to traditional white thermal lamps, cold cathode fluorescent lamps (CCFL) have many advantages such as higher efficiency and longer service life. Therefore, an important number of liquid crystal display (LCD) presently uses CCFL as light source. To achieve a stable operation of the CCFL, the power frequency needed is about 30 KHz through 80 KHz without the stringed wave from the DC current part while the operating voltage is approximately constant. The illumination of the lamp is determined according to the tube current there through. The voltage needed to turn on the lamp is higher than the normal stable operating voltage 2 to 2.5 times. The turn on voltage and operating voltage of the CCFL are determined from the size of the CCFL. Traditional 14″, 15″ LCD screens incorporate CCFL that require a turn-on voltage of about 1400 Vrms, and an operating voltage of about 650 Vrms at the highest normal current of 7 mA. To regulate the CCFL, a common control method is the use of an electrical stabilizer such as a typical fixed frequency operation full bridge phase shift converter that can convert direct current to alternating current.
As shown in FIG. 1, a typical fixed frequency operation full bridge phase shift converter comprises a resonance inductor 105, a capacitor 106 circuit, a lamp 107, NMOS switches 101, 102, 103, 104, and DC current 108. The turn-on or turn-off of NMOS switches 101, 102, 103, 104 are respectively controlled via the gate voltages VG1, VG2, VG3, and VG4. Typical control signals are four similar constant frequencies with a same duty cycle slightly smaller than 50% and different square waves. A typical fixed frequency operation full bridge phase shift converter moves the phase of the control voltage so as to use different sizes of phase retardations to generate different output powers. FIG. 2A and FIG. 2B schematically illustrate the operation time/sequence of a typical fixed frequency operation full bridge phase shift converter. To prevent the NMOS switches 101, 102 (and 103, 104), directly connected to each other, to be simultaneously turned off, which causes a power loss, the control signals VG1, VG2 (and VG3, VG4) must maintain a phase retardation of 180.degree. The phase retardation of VG1, VG3 in FIG. 2A is smaller than that of FIG. 2B, which generates a higher duty cycle VAB and more power outputs.
However, the present use of traditional fixed frequency operation full bridge phase shift converter in LCD screens presents several problems. Within present LCD devices, the DC voltage provided by the circuit is only about 10 to 20 volts. The electrical stabilizer of the CCFL of FIG. 1 needs a direct voltage of hundreds of volts to operate. Moreover, this stabilizer uses an NMOS as power switch. As a result, when it is driven, the voltage VG1 (respectively VG2) at the point A (respectively B) must be cautiously increased. Additional step-up circuits thus must be included within the driving circuits of the NMOSFET power switches 101 and 104.
SUMMARY OF THE INVENTION
It is therefore a first object of the invention to provide a CCFL control circuit that is adapted to a dimensional increase of the LCD devices.
It is a second object of the invention to provide a CCFL control circuit that incorporates a step-up voltage transformer so that the number of high-voltage resistant elements can be reduced within the control circuit.
Furthermore, it is third object of the invention to provide a CCFL control circuit that incorporates PMOSFET as power switches so that additional step-up circuits are not needed to directly drive the switches.
Still, it is a fourth object of the invention to provide a CCFL control circuit in which the cycle of ground switches is fixed so as to change the cycle of the power switches, thereby the voltage conversion is more efficient.
Furthermore, it is a fifth object of the invention to provide a CCFL control circuit in which the cycle of ground switches is fixed so as to change the cycle of the power switches, thereby most of the circuit current flows through the ground switches. Loss increase due to higher resistivity of PMOSFET power switches is therefore favorably reduced.
Still, it is a sixth object of the invention to provide a CCFL control circuit in which stabilization of the lamp current is achieved via pulse width modulation (PWM) feedback control.
Yet, it is a seventh object of the invention to provide a CCFL control circuit in which constant frequency and frequency synchronization are implemented to reduce frequency retardation interference within the multi-lamp circuit, caused by the use of different driving circuits.
Furthermore, it is an eighth object of the invention to provide a CCFL control circuit in which constant frequency and phase synchronization are implemented to reduce phase retardation interference within the multi-lamp circuit, caused by the use of different driving circuits.
Still, it is a ninth object of the invention to provide a CCFL control circuit in which the principal control elements can be fabricated on a same integrated circuit.
To provide a further understanding of the invention, the following detailed description illustrates embodiments and examples of the invention, this detailed description being provided only for illustration of the invention.
BRIEF DESCRIPTION OF THE DRAWINGS
The drawings included herein provide a further understanding of the invention. A brief introduction of the drawings is as follows:
FIG. 1 is a schematic view illustrating a traditional fixed frequency operation full bridge phase shift converter circuit;
FIG. 2A and FIG. 2B are time/sequence charts of the operation of the fixed frequency operation full bridge phase shift converter;
FIG. 3 is a schematic view illustrating a CCFL control integrated circuit according to an embodiment of the invention;
FIG. 4 is a schematic view of a resonance network circuit according to an embodiment of the invention;
FIG. 5 is a schematic view illustrating a multi-lamp applying circuit according to an embodiment of the invention;
FIG. 6 is a schematic view illustrating the relationship between the control signals of the control circuits according to an embodiment of the invention;
FIG. 7A and FIG. 7B are two schematic view illustrating two lamp current phase configurations under similar frequency operation of multi-lamps according to an embodiment of the invention;
FIG. 8A is a schematic view illustrating a triangular wave generator circuit according to an embodiment of the invention;
FIG. 8B is a wave time/sequence chart corresponding to the circuit of FIG. 8A; and
FIG. 9 is a schematic view of a phase synchronization ½ frequency divider circuit according another embodiment of the invention.
DESCRIPTION OF THE PREFERRED EMBODIMENTS
Wherever possible in the following description, like reference numerals will refer to like elements and parts unless otherwise illustrated.
FIG. 3, FIG. 4, and FIG. 5 are schematic views respectively illustrating a cold cathode fluorescent lamp (CCFL) control circuit, a resonance network circuit, and a multi-lamps applying circuit according to an embodiment of the invention. The multi-lamp LCD back-light control circuit of the invention uses a constant operating frequency and PWM (pulse width modulation) feedback to control the CCFL current.
As shown in FIG. 3, the multi-lamp LCD back-light control circuit comprises full bridge switches 317, 318, 319, 320. The resonance network circuit 321, comprised of a step-up transformer 324, inductor 322, and capacitors 323, 326, converts a direct current (DC) from a power source 335 to an alternating current (AC) needed by the CCFL circuit 325. The inductor 322 and capacitors 323, 326 of the resonance network circuit 321 can be disposed on a primary side of the transformer as shown in FIG. 3, or on secondary sides 403, 404 of the transformer as shown in the resonance network circuit 405 of FIG. 4. The inductors 322, 403 of the resonance network circuits 321, 405 as described above may be either independent elements separate from the transformer, or leakage inductors generated by the transformer. Similarly, the secondary side capacitor of the resonance network circuits 321, 405 can be either independent electric element or parasitic capacitor generated between the CCFL and the LCD.
Among the full bridge switches, the switches 317, 320 connected to the power source 335 (also called “power switches”) are PMOSFET switches, while the switches 318, 319 connected to the ground (also called “ground switches”) are NMOSFET switches. Within the full bridge switches, the change and output of the duty cycle of the switches 317, 320 are controlled via a PWM controller 302. In turn, the duty cycle of the switches 318, 319 is constant and further must be controllable above 50%. Furthermore, the phase relationship between the control signal of the NMOSFET ground switches 318, 319 and the control signal of the PMOSFET power switches 317, 320 is invariant. More particularly with respect to the above NMOSFET and PMOSFET having a common drain connection, when said NMOSFET is turned on, said PMOSFET is oppositely turned off With respect to the NMOSFET and PMOSFET without common drain connection, said PMOSFET is turned on only after a preset delay after the turn on of the NMOSFET.
A triangular wave generator 336 is further connected to an input of the PWM controller 302. The operating frequency of the triangular wave generator 336 can be controlled through an external synchronous signal delivered through a control terminal FSYN. A ½ divider circuit 306 is further used to generate a time sequence as driving input of the ground switches 318, 319. The phase of the ½ divider circuit 306 can be synchronized via an external synchronous signal delivered through the control terminal PSYN.
Referring to FIG. 5, the multi-lamps applying circuit is formed via coupling between one another a plurality of integrated circuits (IC) 501, 502, 503, each of which is respectively formed by assembly of the control unit 301 as shown in FIG. 3. The ICs 501, 502, 503 usually have their respective frequency synchronous signal control terminal 504, 505, 506 connected together so that all the ICs are operated with a same frequency. Similarly, respective phase synchronous signal control terminals 507, 508, 509 of the ICs may be connected together to operate all the ICs with a same phase.
With reference to FIG. 3 in conjunction with FIG. 6, now is described the relationship of the control signals between the CCFL control circuit and each control circuit. As illustrated, the CCFL control circuit of the invention is therefore provided with traditional full bridge switch structures and traditional elements of a PWM controller to control the switch of the CCFL and the lamp currents in a feedback control manner. In addition, NMOSFET and PMOSFET are used as switch elements to achieve a more effective switching operation, which favorably increases the efficiency of the whole circuit. Furthermore, the CCFL control circuit of the invention includes different ICs that are connected to one another in such a manner that their respective operating frequency and phase can be synchronously controlled, thereby the perturbations due to frequency and phase differences are prevented during the operation of the lamps.
Within the CCFL control circuit as shown in FIG. 3, the control unit 301 comprises a PWM controller 302, a triangular wave/clock generator 336, a ½ frequency divider 306, and a logic circuit. The PWM feedback control circuit typically measures the AC output 346 of the lamp 325. After AC from the lamp 325 is commutated and filtered through the feedback network 347, the resulting DC is delivered to the inverter input INN of the error amplifier 303 within the PWM controller 302 to be compared with a reference voltage VREF inputted to the not-inverter input INP.
The triangular generator 336 generates a triangular wave output 344 which configuration is shown in FIG. 6 as reference numeral 601. The voltage level (reference numeral 602 in FIG. 6) of the output 345 of the error amplifier 303 is compared with the triangular wave output 344 by means of a comparator 304 to obtain a PWM output wave (reference numeral 603 in FIG. 6) at the output terminal 341. According to the current intensity of the lamp 325, the PWM feedback control circuit can typically modify the voltage level 602 at the output 345 of the error amplifier 303 along which the duty cycle of the output wave 603 from the comparator 304 is also changed. The output of the lamp 325 is thereby automatically regulated.
The full bridge circuit of the invention hence is driven via driving signals, formed from a set of constant duty cycles greater than 50%, that are further accompanied with the output of the PWM controller generating an appropriate change of the duty cycle. In the invention, the control signals of constant cycles drive the NMOSFET 318, 319 as described below.
By means of the ½ frequency divider 306, the clock triangular signal 601 from the triangular wave generator is transformed to a clock signal 604 (also called “half clock signal”) having a frequency equal to half the frequency of the triangular signal 601. The inverter 334 then inverts the half clock signal 604 to an inverted half clock signal 605. Both clock signals 604, 605 are delivered through outputs 339, 340 to delays 312, 311 and OR logic 316, 315 to generate signals 606, 608 of duty cycle greater than 50%, delivered through the outputs NOUT1, NOUT2. The signals 606, 608 have a duty cycle that is delayed a delay time 610 behind the half clock signals 604, 605. If needed, this delay time can be adjusted by means of a time delay controller element 333.
To drive the full bridge switches, the changed duty cycle from the PWM controller is combined with the driving signals of constant cycle in the manner described hereafter. A Boolean AND is applied to the half clock signals 340, 339 and the output 341 of the PWM controller 302 by means of the AND logic 307, 308, so that the PWM output 341 is in outputting configuration only when the half clock signals are in the logic state “1”. The time delays of the delays 309, 310 can be controlled via the controller elements 333. AND logic 348, 349 enable the PWM output 341 to be turned on only after a delay time 611 behind the turn on of the NMOSFET. Because the PMOSFET and NMOSFET respectively are driven via low and high voltages, inverters 313, 314 therefore invert the PWM output to infer the PMOSFET. Within the above circuitry, the control signal 607 driving the PMOSFET 320 at the output POUT2 is in turn-on state (logic “0”) only if the control signal 606 driving the NMOSFET 318 at the output NOUT1 is in turn-on state (logic “1”). Similarly, the control signal 609 driving the PMOSFET 317 at the output POUT1 is in turn-on state (logic “0”) only if the control signal 608 driving the NMOSFET 319 at the output NOUT2 is in turn-on state (logic “1”).
As described above, another characteristic of the invention is a synchronous operation of the frequency and phase. As shown in FIG. 3, the oscillating frequency of the triangular wave generator 336 can be synchronized by means of externally added signals. Processing of the phase synchronous signals is included in the frequency divider circuit 306. As shown in FIG. 5, while operating a plurality of lamps, all the non-inverter input terminals 510, 511, 512 of the error amplifier are connected to a same reference voltage 515 so that the current of the lamps are balanced. Meanwhile, by coupling all the frequency inputs 504, 505, 506 of the controller ICs to one another in a synchronizing configuration, the lamps can be operated with a same frequency. Similarly, by coupling all the phase inputs 504, 505, 506 to one another in a synchronizing configuration, the lamps can be operated with a same phase. The requisite condition of operation with a same phase is that the circuitry must be operated with a same frequency. To obtain a highly effective operation of the circuitry, the operating frequency of the controller ICs can be chosen higher than the resonance frequency of the resonance network.
FIG. 7A and FIG. 7B schematically illustrate two possible phase variations of the lamp current in an operation of multiple lamps with a same frequency. More particularly, FIG. 7A illustrates a lamp current phase in inversion configuration within an operation of multiple lamps with a same frequency. FIG. 7B illustrates a lamp current phase in similar configuration within an operation of multiple lamps with a same frequency. When the phases are inverted, the flow of AC currents through the strayed capacitors between the lamps may generate current leakage. Via synchronous regulation, the invention favorably eliminates the occurrence of current leakage and increases the performance of the entire circuit.
FIG. 8A is a schematic view of the triangular wave generator circuit, and FIG. 8B is a corresponding time/sequence chart. As shown in FIG. 8A, the triangular generator circuit 801 includes a raising edge detector circuit 802. It should be noted that the traditional triangular wave generator without synchronous function has the output 811 of the comparator 810 short-circuited with the input 813 of the NAND gate 814. However, in the embodiment of the invention, an NMOSFET 804 with open drain is added to synchronize the triangular generator with the external signal FSYN. If the connection scheme of the NMOSFET 804 and the external pull-up capacitor 805 is as shown in FIG. 5, a wired NOR logic then is formed from NMOSFET 804, 815 of different ICs, the inverter 803 then can change this NOR logic into an OR logic.
The raising edge detector circuit 802 further includes an AND gate 812 which inputs are connected to the output 811 of the comparator 810 and the output 815 of the raising edge detector circuit 802. The output 813 of the AND gate 812 is inputted to the NAND gate 814 to control the operation of the triangular generator. FIG. 8B is a time/sequence of the entire circuit according to an embodiment of the invention. In FIG. 8B, reference numeral 807 is the external synchronous control triangular signal taken at the point A of FIG. 8A. Reference numeral 808 is the external synchronous control signal. Reference numeral 809 is the external synchronous control signal received by the triangular wave generator clock.
FIG. 9 schematically illustrates a phase synchronous ½ frequency divider circuit according to an embodiment of the invention. As illustrated, to achieve a phase synchronous ½ frequency divider circuit, a common ½ divider circuit 902 is further provided with a D-type inverser 902 that has an output QN directly connected to the input D. Hence, if the phase synchronous terminal PSYN of different IC are connected to one another and further to the Pull up capacitor 907, as traditionally achieved and illustrated in FIG. 5, a wired NOR is formed from NMOSFET 906, 908 of different ICs with a Pull up capacitor 907. The inverter 904 thus inverts the result of the NOR to an OR. When the entire circuit is operated with a same operating frequency, the phase of the output CLK/2 is therefore decided by the first IC switching to a logic state “1” (high level).
It should be apparent to those skilled in the art that the above description is only illustrative of specific embodiments and examples of the invention. The invention should therefore cover various modifications and variations made to the herein-described structure and operations of the invention, provided they fall within the scope of the invention as defined in the following appended claims.
Patent Application
20110084991
