The present invention generally relates to digital communication systems and, more particularly, to an architecture for carrier recovery and equalization for use with communication system receivers.
The recovery of data from modulated signals conveying digital information in symbol form usually requires three functions at a receiver: timing recovery for symbol synchronization, carrier recovery (frequency demodulation to baseband), and channel equalization. Timing recovery is a process by which a receiver clock (timebase) is synchronized to a transmitter clock. This permits a received signal to be sampled at optimum points in time to reduce slicing errors associated with decision-directed processing of received symbol values. Carrier recovery is a process by which a received radio frequency (RF) signal, after being frequency down converted to a lower intermediate frequency (IF) passband (e.g., near baseband), is frequency shifted to baseband to permit recovery of the modulating baseband information. Adaptive channel equalization is a process of compensating for the effects of changing conditions and disturbances in the signal transmission channel. This process typically employs filters that remove amplitude and phase distortions resulting from frequency dependent time variant characteristics of the transmission channel.
Many digital data communication systems employ adaptive equalization to compensate for the effects of changing channel conditions and disturbances on the signal transmission channel. Equalization removes baseband intersymbol interference (ISI) caused by transmission channel disturbances including the low pass filtering effect of the transmission channel. ISI causes the value of a given symbol to be distorted by the values of preceding and following symbols, and essentially represents symbol “ghosts”.
An adaptive equalizer is essentially an adaptive filter. In systems using an adaptive equalizer, it is necessary to provide a method of adapting the filter response so as to adequately compensate for channel distortions. Several algorithms are available for adapting the filter coefficients, thereby changing the filter response. One widely used method employs the Least Mean Squares (LMS) algorithm, which varies filter coefficient values as a function of an error signal. The error signal is formed by subtracting the equalizer output signal from a reference data sequence. As the error signal approaches zero, the equalizer approaches convergence.
When the equalizer operation is initiated, the filter coefficient values (filter tap weights) are usually not set at values which produce adequate compensation of channel distortions. In order to force initial convergence of the filter coefficients, a known “training” signal may be used as the reference signal. The training signal is transmitted to the receiver. The error signal is formed at the receiver by subtracting a locally generated copy of the training signal from the output of the adaptive equalizer, which represents the received training signal. The use of a known signal helps to open the initially occluded “eye”, as known in the art.
After adaptation with the training signal, the “eye” has opened considerably and the equalizer is switched to a decision-directed operating mode for receiving symbols representing data. In this mode, final convergence of the filter tap weights is achieved by using the actual values of data symbols from the output of the equalizer for adapting the filter coefficients instead of using the training signal. The decision directed equalizing mode is capable of tracking and canceling time varying channel distortions more rapidly than methods using training signals. In order for decision directed equalization to provide reliable convergence and stable coefficient values, approximately 90% of the decisions must be correct. The use of a training signal to initially adapt the filter coefficients helps the equalizer achieve this 90% correct decision level.
In practice, however, a training signal is not always available. In such cases “blind’ equalization is often used to provide initial convergence of the equalizer coefficient values and to force the eye to open. Blind equalization has been extensively studied and used for quadrature amplitude modulation (QAM) systems, for example. Among the most popular blind equalization algorithms are the Constant Modulus Algorithm (CMA) and the Reduced Constellation Algorithm (RCA). These algorithms are discussed, for example, in Proakis, Digital Communications, McGraw-Hill: New York, 1989 and in Godard, Self-Recovering Equalization and Carrier Tracking in Two Dimensional Data Communication Systems,” IEEE Transactions on Communications, November 1980.
Briefly, the CMA relies on the fact that, at the decision instants, the modulus of the detected data symbols should lie on a locus of points defining one of several (constellation) circles of different diameters. The RCA relies on forming “super constellations” within the main transmitted constellation. The data signal is first forced to fit into a super constellation. Then the super constellation is subdivided to include the entire constellation.
Typically, within digital communication system receivers, equalization and carrier recovery are entangled within a single loop. Both are included within the same loop because, for optimal reception, an equalizer must have at its input a constellation that is not spinning after the residual carrier frequency offset has been removed. This is particularly true with respect to decision-directed equalization. The carrier error detector, which also is commonly decision-directed, functions more efficiently and accurately with an equalized signal.
However, this architecture has an inherent shortcoming resulting from the excessive multi-symbol delay that is introduced into the carrier tracking loop (CTL) by the equalizer's feed-forward (Non-Causal) section. This excessive delay is known to limit the maximum allowable loop gain as well as to limit the carrier acquisition range. One common solution has been to keep the loop gain sufficiently low thereby ensuring stability. This approach also circumvents the limited acquisition range by “stepping” through the desired acquisition range while allowing sufficient time for the CTL to acquire at each step.
It would be beneficial to reduce the multi-symbol delay that is introduced into the CTL by the equalizer while also preserving the advantages associated with the inclusion of an equalizer within a CTL.
A method and system for carrier recovery and equalization for use within a receiver is provided. In accordance with the principles of the invention, a receiver comprises a carrier tracking loop (CTL) and an equalizer that is selectively configured either external to the CTL or within the CTL. Illustratively, the equalizer can be positioned according to a measure of convergence for the CTL. The measure of convergence can be met when the CTL converges to about a true residual offset value. Notably, the equalizer can operate in a blind mode when located external to the CTL and can be switched to a decision-directed mode when within the CTL.
In one embodiment, the CTL comprises a derotator configured to derotate receive symbols and a slicer configured to correct errors in the received symbols. The CTL also includes an error detector configured to compare symbol output of the slicer with the received symbols to determine an error signal, a loop filter for processing the error signal, and a numerically controlled oscillator. The numerically controlled oscillator can be driven by an output from the loop filter to provide a signal to the derotator for use in derotating the received symbols.
When located external to the CTL, the equalizer is illustratively located prior to the derotator with respect to the signal path. When located within the CTL, the equalizer is illustratively located between the derorator and the slicer. The equalizer can adapt to the incoming symbols and responsively suspend adaptation.
Preferred embodiments of the present invention will be described below in more detail, with reference to the accompanying drawings.
Other than the inventive concept, the elements shown in the figures are well known and will not be described in detail. For example, other than the inventive concept, a set-top box or digital television (DTV) and the components thereof, such as a front-end, Hilbert filter, carrier tracking loop, video processor, remote control, etc., are well known and not described in detail herein. In addition, the inventive concept may be implemented using conventional programming techniques, which, as such, will not be described herein. Finally, like-numbers on the figures represent similar elements.
The present invention provides a carrier tracking loop (CTL) configuration having a variable architecture. This is illustrated in
The preliminary demodulation brings the signal closer to baseband so that the subsequent circuits do not have to operate on the intermediate frequency (IF) signal. The locally generated carrier frequency used for this purpose may not precisely match the transmitter carrier frequency, whereby phase errors result from this demodulation. These phase errors are corrected by a further demodulation process involving the CTL 100.
CTL 100 comprises a derotator 105, an equalizer 110, a slicer 115, an error detector 120, a loop filter 125, and a numerically controlled oscillator (NCO) 130. The derotator 105, as is known, removes the residual carrier frequency from the incoming signal, effectively derotating the signal back to baseband. The derotator 105 can be a complex multiplier that multiplies the incoming signal by a difference or error sinusoidal signal generated by the NCO 130.
The equalizer 110 is an adaptive equalizer that can switch between operating in blind and decision-directed modes. As known, the equalizer 110 generally removes baseband intersymbol interference (ISI) caused by transmission channel disturbances. In one embodiment, and as shown, the equalizer 110 is located within the CTL 100. That is, the equalizer 110 is disposed between the derotator 105 and the slicer 115.
The slicer 115 is illustratively implemented as a decision directed component that processes a current received symbol and makes a decision as to what the transmitted symbol is believed to be. The slicer 115 makes a decision by quantizing the received sample to a nearest constellation point. The quantized symbol is used as an estimate of the actual transmitted symbol. For each current received symbol, the slicer 115 selects, from a look-up table, the constellation point that is closest in Euclidean distance to the input symbol sample as its decision (the quantized symbol).
The error detector 120 receives an input from the equalizer 110 and the slicer 115. Generally, the error detector 120 produces an error signal that represents a phase difference between the symbol output of the equalizer 110 and the symbol output of the slicer 115. The loop filter 125 processes the error signal from the error detector 120 to provide a higher quality signal to the NCO 130. The error signal generated by the error detector 120 can include an error term and a noise term. For example, the noise term can include high frequency components. The loop filter 125 processes the error signal to generate a useful error signal while suppressing the effect of the noise.
The NCO 130 is an electronic system for synthesizing a range of frequencies from a fixed timebase. The NCO 130 can include a digital waveform generator that increments a phase counter by a per-sample increment. This phase can be looked up in a waveform table to create a sine waveform. The NCO 130, however, is phase and frequency-agile. Accordingly, the NCO 130 can be modified to produce phase-modulated or frequency-modulated outputs, or quadrature outputs.
As before, signals that have been preliminarily demodulated and digitized are provided to equalizer 110 prior to being provided to the derotator 105. By switching the architecture of the CTL 100 between those of
According to one embodiment, the CTL 100 can be implemented as hardware using one or more discrete components, integrated circuits, or as an application specific circuit. The location of the equalizer 110 can be effectively changed through the use of one or more multiplexers. That is, the location of the equalizer 110, with respect to the signal path, can be varied using multiplexers. This is illustrated in
In another embodiment, the CTL 100 can be implemented within software. For example, modules of program code can be used to implement each of the components discussed with reference to
In step 310, the architecture of the CTL can be initialized or switched to that illustrated in
The equalizer begins functioning and adapting in blind mode such that the equalizer coefficients begin to converge. While the equalizer continues to adapt, the CTL may be inactive. That is, the CTL may not be adapting. In step 320, a decision can be made as to whether the equalizer has adapted sufficiently. The method 300 can continue to loop as shown until the equalizer has adequately adapted. Any one of a number of known techniques can be used to determine if the equalizer has sufficiently adapted. For example, over a period of time, a count can be determined of the number of equalized received symbols falling within a predefined portion of the signal space. The equalizer can be determined to have sufficiently adapted when this count exceeds a predetermined number. In step 325, once adaptation has occurred and the eye is sufficiently open to allow reasonably accurate slicing decisions on average from the slicer, further adaptation of the equalizer can be stopped.
In step 330, CTL operation can commence and begin adaptation. The integrator is no longer held at a fixed value. While the CTL functions or adapts, the equalizer can remain in a suspended state to prevent the CTL from having to adapt to a changing signal resulting from further adaptation of the equalizer. It can be assumed that the channel is quasi-static and will not change appreciably while the CTL is converging. This is likely with respect to applications of the inventive concept pertaining to cable receivers. Notably, as the equalizer is located outside of the CTL, the delay through the CTL is reduced, thereby allowing a wider pull in range.
In step 335, a determination can be made as to whether the CTL has converged to about a true residual offset value. This occurs when the constellation output of the derotator is virtually static. The method can continue to loop until such time when the CTL converges. At that point the method can proceed to step 340. In step 340, CTL adaptation is suspended. That is, the integrator value can be frozen. In step 345, the CTL architecture can be switched to that shown in
In step 355, the CTL integrator is released such that the CTL continues to adapt again. At this point, both the CTL and the equalizer are functioning and continuing to adapt and, in step 370, an output signal is provided.
Besides the flow chart shown in
The present invention can be realized in hardware, software, or a combination of hardware and software. Aspects of the present invention also can be embedded in a computer program product, which comprises all the features enabling the implementation of the methods described herein, and which when loaded in a computer system is able to carry out these methods. Computer program or application in the present context means any expression, in any language, code or notation, of a set of instructions intended to cause a system having an information processing capability to perform a particular function either directly or after either or both of the following: a) conversion to another language, code or notation; b) reproduction in a different material form.
This invention can be embodied in other forms without departing from the spirit or essential attributes thereof. Accordingly, reference should be made to the following claims, rather than to the foregoing specification, as indicating the scope of the invention.
Filing Document | Filing Date | Country | Kind | 371c Date |
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PCT/US05/02867 | 2/2/2005 | WO | 11/8/2006 |
Number | Date | Country | |
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60570294 | May 2004 | US |