The present disclosure relates generally to object detection and vehicle position determination, for example, in an application for inductive wireless charging of electric vehicles. In particular, the present disclosure is directed to a multi-purpose detection circuit configurable for detecting foreign metallic objects, living objects located near an inductive wireless power transmitter as well as for detecting a vehicle above the wireless power transmitter and for determining a position of the vehicle relative to the inductive wireless power transmitter.
Inductive wireless power transfer (WPT) systems provide one example of wireless transfer of energy. In an inductive WPT system, a primary power device (or wireless power transmitter) transmits power wirelessly to a secondary power device (or wireless power receiver). Each of the wireless power transmitter and wireless power receiver includes an inductive power transfer structure, typically a single or multi-coil arrangement of windings comprising electric current conveying materials (e.g., copper Litz wire). An alternating current passing through the coil e.g., of a primary wireless power transfer structure produces an alternating magnetic field. When a secondary wireless power transfer structure is placed in proximity to the primary wireless power transfer structure, the alternating magnetic field induces an electromotive force (EMF) into the secondary wireless power transfer structure according to Faraday's law, thereby wirelessly transferring power to the wireless power receiver if a resistive load is connected to the wireless power receiver. To improve a power transfer efficiency, some implementations use a wireless power transfer structure that is part of a resonant structure (resonator). The resonant structure may comprise a capacitively loaded inductor forming a resonance substantially at a fundamental operating frequency of the inductive WPT system (e.g., in the range from 80 kHz to 90 kHz).
Inductive wireless power transfer to electrically chargeable vehicles at power levels of several kilowatts in both domestic and public parking zones may require special protective measures for safety of persons and equipment. Such measures may include detection of foreign objects in an inductive power region of the inductive WPT system where electromagnetic field exposure levels exceed certain limits. This may be particularly true for systems where the inductive power region is open and accessible. Such measures may include detection of electrically conducting (metallic) objects and living objects, (e.g., humans, extremities of humans, or animals) that may be present within or near the inductive power region.
In certain applications for inductive wireless charging of electric vehicles, it may be useful to be able to detect foreign objects that may be present in the inductive power region and that could be susceptible to induction heating due to the high magnetic field strength in that region. In an inductive wireless power transfer system for electric vehicle charging operating at a fundamental frequency in the range from 80 kHz to 90 kHz, magnetic flux densities in the inductive power region (e.g., above a primary wireless power transfer structure) can reach relatively high levels (e.g., above 2 mT) to allow for sufficient power transfer (e.g., 3.3 kW, 7 kW, 11 kW, and the like). Therefore, metallic objects or other objects present in the magnetic field can experience undesirable induction heating. For this reason, foreign object detection (FOD) may be implemented to detect metallic objects or other objects that are affected by the magnetic field generated by the primary and/or the secondary wireless power transfer structure of the inductive WPT system.
In certain applications for inductive wireless charging of electric vehicles, it may also be useful to be able to detect living objects that may be present within or near an inductive power region where the level of electromagnetic field exposure exceeds certain limits (e.g., as defined by the International Commission on Non-Ionizing Radiation Protection (ICNIRP) recommendation). For this reason, living object detection (LOD) may be implemented to detect living objects (e.g., human extremities, animals), or other objects that may be exposed to the magnetic field generated by the primary and/or the secondary wireless power transfer structure of the inductive WPT system.
In further applications for inductive wireless charging of electric vehicles, it may also be useful to be able to detect a vehicle or the type of vehicle that may be present above the wireless power transmitter (e.g., above the primary wireless power transfer structure). For this reason, vehicle detection (VD) may be implemented. In yet another application for inductive wireless charging of electric vehicles, it may also be useful to be able to transmit data (e.g., a vehicle identifier or the like) from the vehicle-based secondary device to the ground-based primary device. For this reason, vehicle detection (VD) may be extended for receiving low rate signaling from the vehicle.
Efficiency of an inductive WPT system for electric vehicle charging depends at least in part on achieving sufficient alignment between the ground-based primary wireless power transfer structure and the secondary wireless power transfer structure. Therefore, in certain applications for inductive wireless charging of electric vehicles, it may be useful to be able to determine a position of the vehicle relative to the wireless power transmitter for purposes of guidance and alignment. More specifically, it may be useful to be able to determine a position of the vehicle-based wireless power transfer structure (e.g., the secondary wireless power transfer structure) relative to the ground-based wireless power transfer structure (e.g., the primary wireless power transfer structure). For this reason, position determination (PD) may be implemented.
In an aspect of hardware complexity reduction and cost saving, it may be useful and desirable to provide FOD, LOD, VD, and PD by a common multi-purpose detection circuit.
In one aspect of the disclosure, an apparatus for determining at least one of a presence of a metallic object, living object, vehicle, type of vehicle, and a vehicle position is provided. The apparatus includes a plurality of inductive sense circuits and a plurality of capacitive sense circuits. Each of the plurality of inductive sense circuits includes at least one inductive sense element (e.g., a sense coil) and an associated capacitive element to compensate for the gross reactance as presented at the terminals of the at least one inductive sense element at an operating frequency herein referred to as the sense frequency. Each of the plurality of capacitive sense circuits includes at least one capacitive sense element (e.g., a sense electrode) and an associated inductive element to compensate for the gross reactance as presented at the terminals of the at least one capacitive sense element at the sense frequency. At least one of the plurality of inductive and capacitive sense circuits also includes an impedance matching element (e.g., a transformer) for transforming the impedance of the sense circuit to match with an operating impedance range of the apparatus. The apparatus further includes a measurement circuit for selectively and sequentially measuring an electrical characteristic (e.g., an impedance) in each of the plurality of inductive and capacitive sense circuits according to a predetermined time multiplexing scheme. More specifically, the measurement circuit includes a driver circuit including multiplexing (input multiplexing) electrically connected to the plurality of inductive and capacitive sense circuits for selectively and sequentially driving each of the plurality of sense circuits with a drive signal (e.g., a current signal) at the sense frequency based on a driver input signal. The measurement circuit further includes a measurement amplifier circuit including multiplexing (output multiplexing) electrically connected to the plurality of inductive and capacitive sense circuits for selectively and sequentially amplifying a measurement signal (e.g., a voltage signal) in each of the plurality of sense circuits and for providing a measurement amplifier output signal indicative of the measurement signal in each of the plurality of sense circuits. The measurement circuit also includes a signal generator circuit electrically connected to the input of the driver circuit for generating the driver input signal. The measurement circuit further includes a signal processing circuit electrically connected to the output of the measurement amplifier circuit for receiving and processing the measurement amplifier output signal and for determining the electrical characteristic in each of the plurality of inductive and capacitive sense circuits based on the driver input signal and the measurement amplifier output signal. The apparatus further includes a control and evaluation circuit electrically connected to the measurement circuit for controlling the signal generator circuit, for controlling the input and output multiplexing according to the predetermined time multiplexing scheme, for evaluating the electrical characteristic as measured in each of the inductive and capacitive sense circuits, and for determining at least one of a presence of a metallic object, living object, vehicle, type of vehicle, and a vehicle position based on changes in the measured electrical characteristics.
In another aspect of the disclosure, a method for determining at least one of a presence of a metallic object, living object, vehicle, type of vehicle, and a vehicle position is provided. The method includes selectively and sequentially measuring, in a measurement circuit, an electrical characteristic (e.g., an impedance) in each of the plurality of inductive and capacitive sense circuits according to a predetermined time multiplexing scheme. More specifically, the method includes selectively and sequentially applying, from a driver circuit as part of the measurement circuit and including input multiplexing, a drive signal (e.g., a current signal) at a sense frequency to each of the plurality of inductive and capacitive sense circuits according to the predetermined time multiplexing scheme. The method further includes selectively and sequentially amplifying, in a measurement amplifier circuit as part of the measurement circuit, and including output multiplexing, a measurement signal (e.g., a voltage signal) in each of the plurality of inductive and capacitive sense circuits according to the predetermined time multiplexing scheme, and providing a measurement amplifier output signal indicative for the measurement signal. The method further includes applying, from a signal generator circuit as part of the measurement circuit, a driver input signal to the driver circuit. The method further includes receiving and processing, in a signal processing circuit as part of the measurement circuit, the measurement amplifier output signal, and determining the electrical characteristic in each of the plurality of inductive and capacitive sense circuits based on the driver input signal and the measurement amplifier output signal. The method further includes controlling, in a control and evaluation circuit, the signal generator circuit and the input and output multiplexing according to the time multiplexing scheme. The method further includes evaluating the electrical characteristic as measured in each of the inductive and capacitive sense circuits and determining at least one of a presence of a metallic object, living object, vehicle, type of vehicle, and a vehicle position based on changes in the measured electrical characteristics.
In the figures, the third and fourth digit of a reference number identify the figure in which the reference number first appears. The use of the same reference numbers in different instances in the description or the figures indicates like elements.
The detailed description set forth below in connection with the appended drawings is intended as a description of example implementations and is not intended to represent the only implementations in which the techniques described herein may be practiced. The term “example” used throughout this description means “serving as an example, instance, or illustration,” and should not necessarily be construed as preferred or advantageous over other example implementations. The detailed description includes specific details for the purpose of providing a thorough understanding of the example implementations. In some instances, some devices are shown in block diagram form. Drawing elements that are common among the following figures may be identified using the same reference numerals.
As mentioned above foreign object detection (FOD) (and particularly metal object detection) may be valuable for a variety of applications. For detection in a predetermined region, a FOD system may include a plurality of inductive sense circuits each including an inductive sense element (e.g., a sense coil) distributed across a predetermined area (e.g., a planar array of sense coils integrated into the ground-based wireless power transfer structure). The predetermined region may be defined by the space where metal objects may be found and where the magnetic flux density exceeds certain limits (e.g., a threshold determined based on what levels of temperature a metal object might be heated up). This is generally a three-dimensional space above the plurality of indictive sense elements. The number of the inductive sense elements may be proportional or related to the minimum size of objects that are desirable to be detected. For a system that is configured to detect small objects (e.g., a paper clip), the number of sense elements may be relatively high (e.g., in the order of 100). An example FOD system is described in U.S. Pat. No. 10,627,257, titled Systems, Methods, and Apparatus for Detection of Metal Objects in a Predetermined Space, the entire contents of which are hereby incorporated by reference.
As mentioned above living object detection (LOD) (e.g., human extremities, animals) may be valuable for a variety of applications. For detection in a predetermined region, a LOD system may include a plurality of capacitive sense circuits each including a capacitive sense element (e.g., a sense electrode) e.g., disposed along the periphery of a ground-based wireless power transfer structure of a WPT system. The predetermined region may be defined by the space accessible for living objects and where living objects may be located and where the exposure magnetic field strength exceeds certain limits (e.g., as recommended by ICNIRP). This is generally a three-dimensional space. The number of the capacitive sense elements may be proportional or related to the minimum size of living objects that are desirable to be detected. For a system that is configured to detect human extremities (e., a hand) and animals (e.g., a cat), the number of sense elements may be relatively low (e.g., in the order of 4). A measurement drive circuitry for applying drive signals to each of the plurality of capacitive sense circuits each including a capacitive sense element and additional elements for conditioning, as well as corresponding measurement circuitry as needed for measuring an electrical characteristic in each of the plurality of capacitive sense circuits and for looking for changes in the electrical characteristics that may correspond to the presence of a living object. An example LOD system is described in U.S. Pat. No. 9,952,266, titled Object Detection for Wireless Energy Transfer Systems, the entire contents of which are hereby incorporated by reference.
As mentioned above vehicle detection (VD) or detection of the type of vehicle above the ground-based wireless power transfer structure of a WPT system may be valuable for a variety of applications. For detection of a vehicle or the type of vehicle, a VD system may include a plurality of inductive sense circuits each including an inductive sense element (e.g., a sense coil) distributed across an area defined by the ground-based wireless power transfer structure (e.g., a planar array of sense coils) and a plurality of capacitive sense circuits each including a capacitive sense element (e.g., a sense electrode) disposed in an area defined by the ground-based wireless power transfer structure. Drive circuitry for applying drive signals to each of the inductive and capacitive sense circuits, each including an inductive and capacitive sense element, respectively and additional elements for conditioning, as well as corresponding measurement circuitry as needed for measuring an electrical characteristic in each of the plurality of capacitive sense circuits and for looking for changes in the electrical characteristics that may correspond to the presence of a vehicle.
As mentioned above determination of a position (PD) of a vehicle (e.g., the position of the vehicle-based wireless power transfer structure relative to the ground-based wireless power transfer structure of a WPT system) may be valuable for a variety of applications. For determination of a vehicle position, a PD system may include a plurality of inductive sense circuits each including an inductive sense element (e.g., a sense coil) distributed across an area defined by the ground-based wireless power transfer structure (e.g., a planar array of sense coils) and a plurality of capacitive sense circuits each including a capacitive sense element (e.g., a sense electrode) disposed in an area defined by the ground-based wireless power transfer structure.
In some implementations, the PD system is configured to support a passive beacon PD technique. Passive beacon PD uses at least one passive beacon transponder that may be integrated into the vehicle-based wireless power transfer structure or that may be mounted elsewhere at the vehicle underbody. When positioned above the inductive and capacitive sense element array of the multi-purpose detection circuit, the passive beacon transponder produces a distinct time-varying change (a modulated response) in the electrical characteristic of at least one of the plurality of inductive sense circuits and capacitive sense circuits. This modulated response may be used for determining a position of the at least one passive beacon transponder relative to the array of sense elements, which is related to the position of the vehicle-based wireless power transfer structure relative to the ground-based wireless power transfer structure. The at least one passive beacon transponder may also be used for determining presence of a vehicle (VD) or the type of vehicle e.g., by means of a modulation that is characteristic for the type of vehicle. Further, the at least one passive beacon transponder may be used to transmit data (e.g., at a low data rate) to the primary device by means of the passive modulation technique.
In some implementations, the at least one passive beacon transponder includes an inductive passive beacon transponder configured to mainly interact with the inductive sense circuits. In other implementations, the at least one passive beacon transponder includes a capacitive passive beacon transponder configured to mainly interact with the capacitive sense circuits. In further implementations, the at least one passive beacon transponder is configured to interact with both the inductive and capacitive sense circuits. An example inductive passive beacon PD system is described in U.S. patent application Ser. No. 16/052,445, titled Hybrid Foreign Object Detection and Positioning System, the entire contents of which are hereby incorporated by reference.
Circuitry for applying drive signals to each of the plurality of inductive and/or capacitive sense circuits each including a sense element and additional elements for conditioning, as well as corresponding measurement, control and evaluation circuitry as needed for measuring an electrical characteristic in each of the plurality of inductive sense circuits and detecting changes in the electrical characteristics that may be indicative of one of the presence of a metal object, a living object, a vehicle, the type of vehicle, and a vehicle position may be complex and costly as the number of sense elements increases. Therefore, in an aspect of hardware complexity reduction and cost saving, it may be useful and desirable to combine the various functions such as FOD, LOD, VD, data signaling, and PD in a single system referred to herein as the multi-purpose detection circuit.
An electric vehicle is used herein to describe a remote system, an example of which is a vehicle that includes, as part of its locomotion capabilities, electrical power derived from a chargeable energy storage device (e.g., one or more rechargeable electrochemical cells or other type of battery). As non-limiting examples, some electric vehicles may be hybrid electric vehicles that include, besides electric motors, a traditional combustion engine for direct locomotion or to charge the vehicle's battery. Other electric vehicles may draw all locomotion ability from electrical power. An electric vehicle is not limited to an automobile and may include motorcycles, carts, scooters, and the like.
A foreign object is used herein to describe an object that does not naturally belong to the WPT system. A foreign object may include a metallic object, a non-living dielectric (substantially nonconductive) object, a living object (e.g., an animal, a human extremity), a vehicle, or a combination thereof. It may describe an object that needs to be detected for purposes of safety of equipment or persons, but it may also refer to an object of no harm that is potential to produce a false positive detection in a multi-purpose detection system.
The inductive sense elements 107a, 107b, . . . , 107n and capacitive sense elements 109a, 109b, . . . , 109n are configured to sense at least one of a presence of a foreign object (e.g., object 110) in proximity to at least one of the plurality of inductive sense elements 107a, 107b, . . . , 107n, a living object (e.g., object 114) in proximity to at least one of the plurality of capacitive sense elements 109a, 109b, . . . , 10n, a vehicle or type of vehicle (not shown in
Each of the plurality of inductive sense elements 107a, 107b, . . . , 107n is shown in
Each of the plurality of capacitive sense elements 109a, 109b, . . . , 109n is shown in
Each of the plurality of inductive sense circuits 106 and the plurality of capacitive sense circuits 108 including a corresponding sense element of the plurality of inductive sense elements 107a-107n and the plurality of capacitive sense elements 109a-109n are operably connected to a measurement circuit 104. The measurement circuit 104, including multiplexing (not shown in
The measurement circuit 104 is configured to cause each of the plurality of inductive sense elements (e.g., sense coils) 107a, 107b, . . . , 107n to selectively and sequentially generate an alternating magnetic field at the sense frequency, e.g., by selectively and sequentially applying a sense signal (e.g., a current) to each of the plurality of inductive sense circuits 106a, 106b, . . . , 106n. If a metallic object (e.g., object 110) is present in the alternating magnetic field, eddy currents will be generated in the object. According to Lentz' law, the eddy currents in the object will generate another (secondary) magnetic field that interacts with the primary magnetic field as generated by the respective sense element, and a mutual coupling is developed. This may cause a change in an electrical characteristic (e.g., an impedance) as measured by the measurement circuit 104 in the respective inductive sense circuit (e.g., inductive sense circuit 106a). A change in a measured electrical characteristic may also be caused by a substantially non-conductive but ferromagnetic object (not shown in
The measurement circuit 104 is further configured to cause each of the plurality of capacitive sense elements (e.g., sense electrodes) 109a, 109b, . . . , 109n to selectively and sequentially generate an alternating electric field at the sense frequency, e.g., by selectively and sequentially applying a sense signal (e.g., a current) to each of the plurality of capacitive sense circuits 108a, 108b, . . . , 108n. If a substantially non-conductive, dielectric object (e.g., living object 114 or non-living object 112) with a relative permittivity εr>1 is present in the alternating electric field, it will interact with the electric field. This may cause a change in an electrical characteristic (e.g., an impedance) as measured by the measurement circuit 104 in the respective capacitive sense circuit (e.g., capacitive sense circuit 108a). A change in a measured electrical characteristic may also be caused by a metallic object (e.g., object 110) as it will also interact with the alternating electric field as generated by the respective capacitive sense element. Applying a sense signal (e.g., current) to a capacitive sense circuit (e.g., sense circuit 106a) may also cause the respective capacitive sense element to generate an alternating magnetic field that may interact with a metallic object (e.g., object 110) causing a change in the electrical characteristic as measured in the respective capacitive sense circuit (inductive sensing effect). However, this effect may be orders of magnitude weaker than the electric field effect.
The control and evaluation circuit 102 is configured to control the measurement circuit 104 (e.g., the multiplexing) and to evaluate the outputs of the measurement circuit 104, to determine at least one of a presence of a foreign object (e.g., object 110), living object (e.g., object 114), a presence of a vehicle with reference to
In an example implementation or operation of the control and evaluation circuit 102, the evaluation is based on a time-differential detection scheme that is sensitive e.g., to a fast (e.g., abrupt) change in a sequence (time-series) of consecutive outputs of the measurement circuit 104 (e.g., the measured electrical characteristics), each associated to the same at least one sense circuit (e.g., sense circuit 106a) of the plurality of sense circuits 106 and 108. The measured electrical characteristics (e.g., an impedance) may refer to outputs as obtained after processing (e.g., filtering, combining, averaging, etc.) in the control and evaluation circuit 102. In some implementations or operations based on a time-differential detection scheme, presence of an object (e.g., object 110) is assumed, if at least one difference between a first measurement output associated to a sense circuit (e.g., sense circuit 106a) and at least one first time and a second measurement output associated to the same sense circuit and at least one second time exceeds a threshold. Using time-differential detection, an object can be detected when it enters or leaves the proximity of the at least one sense element (e.g., sense element 107a) or generally when it moves in the proximity of the at least one sense element.
In a further example implementation or operation of the control and evaluation circuit 102, the evaluation is based on a sense circuit-differential detection scheme that is sensitive to differences between measurement outputs associated to different sense circuits of the plurality of sense circuits (e.g., 106). This detection scheme may be also referred to as space-differential detection. In some implementations or operations based on a space-differential detection scheme, presence of an object (e.g., object 110) is assumed if at least one difference between a first measurement output associated to at least one first sense circuit (e.g., sense circuit 106a) and a second measurement output associated to at least one second sense circuit (e.g., sense circuit 106b) exceeds a threshold. In some implementations or operations of a true space-differential detection scheme, the plurality of measurement outputs used to determine a difference refer to substantially the same time. It may be appreciated that in certain cases, space-differential detection may be less sensitive and reliable than time-differential detection because sense circuits of the plurality of sense circuits 104a, 104b, . . . , 104n may be at least partially differently (individually) affected by temperature, mechanical impacts, and aging.
In some aspects, time-differential detection may be sensitive to movements of metallic structures in the environment of the sense element array (e.g., array 107). Such environmental effects may include movements of the metallic vehicle underbody structure when a vehicle is parked over the wireless power transfer structure. These movements may cause false detections in certain implementations of the multi-purpose detection circuit 100 that is solely based on a time-differential approach. Therefore, in some aspects, it may be desirable to mitigate such disturbance effects.
Combining the time-differential scheme with a space-differential scheme on top may be an approach to effectively discriminate such disturbance effects. In certain implementations or operations of a space-differential detection scheme, presence of an object is determined by evaluating at least one difference between a measurement output associated to at least one sense circuit (e.g., sense circuit 106a) and a reference value that is determined based on a plurality of measurement outputs each associated to a different sense circuit of the plurality of sense circuits (e.g., 106). This reference value may be e.g., a mean value, a median value (50th percentile), or any other percentile value that is derived from a histogram built upon the plurality of measurement outputs. It may be appreciated that this special detection scheme has the potential to discriminate environmental effects e.g., from a moving vehicle underbody that may produce changes in an electrical characteristic (e.g., an impedance) in a majority (cluster) of sense circuits. This special scheme may be considered as a mechanism that automatically adapts the detection threshold used by the control and evaluation circuit 102 for determining presence of an object (e.g., object 110). More specifically, in some implementations or operations, the control and evaluation circuit 102 automatically adjusts the reference value as described above. When the vehicle underbody is moving, the reference value may increase as well as the detection threshold. Increasing the detection threshold reduces the false detection rate but also the detection sensitivity (detection probability) to some extent. Therefore, a somewhat lower sensitivity may be accepted for an object entering the predetermined space while the vehicle is moving. As soon as the vehicle underbody comes to rest, the detection threshold is readjusted automatically, and the object detection circuit 100 may return to its ordinary sensitivity (detection probability) maintaining a specified false detection rate.
The wireless power transfer structure 200 includes a coil 202 (e.g., a Litz wire coil) also referred to as the WPT coil that is configured to generate an alternating magnetic field when driven with a current by a power conversion circuit (not shown herein). The wireless power transfer structure 200 may further include a ferrite 204 structure configured to channel and/or provide a path for magnetic flux (e.g., may be arranged in one or more ferrite tiles). The wireless power transfer structure 200 may also include a metal shield 206 (also sometimes referred to as a back plate). The metal shield 206 is configured to prevent the magnetic field or associated electromagnetic emissions from extending far beyond a boundary determined by the shield 206 or at least to attenuate the magnetic field extending beyond that boundary. As an example, the shield 206 may be formed from aluminum.
The vehicle-based wireless power transfer structure 310 includes a WPT coil 312, a layer of ferrite 315, and a shield 316 made of an electrically conductive material. In some implementations, the shield 316 may be formed from a portion of the apparatus that the ferrite 315 and the WPT coil 312 are affixed to the metallic underbody of a vehicle 330. In this case, a housing 318 configured to house the WPT coil 312 and ferrite 315 is provided but that may not house the shield 316. However other implementations are possible where a conductive back plate is included in the housing 318. A power conversion circuit (not shown herein) may be electrically connected to the WPT coil 312 or a portion or all may also be housed in the housing 318.
As mentioned above and as illustrated in
The ground-based (e.g., transmit) wireless power transfer structure 200 may be configured to generate a magnetic field 232. The vehicle-based wireless power transfer structure 310 may be configured to inductively receive power via the magnetic field. Furthermore, as the ground-based wireless power transfer structure 200 may be positioned on a ground or other top facing surface, an object (e.g., object 110 or 112) may come to rest at the top surface of the housing 328 as illustrated in
Each of the plurality of inductive sense circuits 106 may also include an associated capacitive element (not shown herein) to compensate for the gross reactance as presented at the terminals of the at least one inductive sense element at the sense frequency. Each of the plurality of capacitive sense circuits 108 may also include an associated inductive element (not shown herein) to compensate for the gross reactance as presented at the terminals of the at least one capacitive sense element at the sense frequency. At least one of the plurality of inductive and capacitive sense circuits also includes an impedance matching element (e.g., a transformer) for transforming the impedance of the sense circuit (e.g., sense circuit 108a) to match with an operating impedance range of the multi-purpose detection circuit 100. In an example implementation, each of the plurality of inductive sense circuits 106 is naturally matched with an operating impedance range without using an additional impedance matching element. However, the plurality of capacitive sense circuits 108 is not naturally matched, and therefore an additional impedance matching element (e.g., a transformer) is used. In another example implementation, it is vice-versa. In a further example implementation, both the plurality of inductive and capacitive sense circuits 106 and 108, respectively, include an additional impedance matching element.
The measurement circuit 104 is electrically connected to the plurality of inductive and capacitive sense circuits and configured for selectively and sequentially measuring one or more electrical characteristics (e.g., an impedance) in each of the plurality of inductive and capacitive sense circuits according to a predetermined time multiplexing scheme.
The control and evaluation circuit 102 is electrically connected to the measurement circuit 104 and configured to control time multiplexing (input multiplexer (MUX) control and output MUX control in
The measurement circuit 104 further includes a driver circuit 402, a measurement amplifier circuit 404, a signal generator circuit 406, and a signal processing circuit 408.
The driver circuit 402 including multiplexing (input multiplexing) is electrically connected to the plurality of inductive sense circuits 106 and the plurality of capacitive sense circuits 108 and configured to selectively and sequentially apply a drive signal (e.g., a current signal) at the sense frequency to each of the plurality of inductive sense circuits 106 and the plurality of capacitive sense circuits 108 based on a driver input signal generated by the signal generator circuit 406.
The measurement amplifier circuit 404 including multiplexing (output multiplexing) is electrically connected to the plurality of inductive sense circuits 106 and the plurality of capacitive sense circuits 108 and configured to selectively and sequentially amplify a measurement signal (e.g., a voltage signal) in each the plurality of inductive sense circuits 106 and the plurality of capacitive sense circuits 108 and to provide a measurement amplifier output signal indicative of the measurement signal in each of the plurality of sense circuits.
The signal generator circuit 406 electrically connected to the input of the driver circuit 402 is configured to generate the driver input signal.
The signal processing circuit 408 electrically connected to the output of the measurement amplifier circuit 404 is configured to receive and process the measurement amplifier output signal and to determine the one or more electrical characteristics in each of the plurality of inductive and capacitive sense circuits based on the driver input signal and the measurement amplifier output signal.
The dashed lines used in
Example implementations of the measurement circuit 104 and the control and evaluation circuit 102 are described in U.S. Pat. No. 9,726,518, titled Systems, Methods, and Apparatus for Detection of Metal Objects in a Predetermined Space, U.S. Pat. No. 9,921,045, titled Systems, Methods, and Apparatus for Increased Foreign Object Detection Loop Array Sensitivity, in U.S. Pat. No. 10,295,693, titled Systems, Methods, and Apparatus for Foreign Object Detection Loop Based on Inductive Thermal Sensing, in U.S. Pat. No. 10,302,795, titled Systems, Methods, and Apparatus for Detecting Ferromagnetic Objects in a Predetermined Space, in U.S. Pat. No. 10,298,049, titled Systems, Methods, and Apparatus for Detecting Metallic Objects in a Predetermined Space via inductive kinematic Sensing, in U.S. patent application Ser. No. 16/226,156, titled Foreign Object Detection Circuit Using Current Measurement, in U.S. patent application Ser. No. 16/392,464, titled Extended Foreign Object Detection Signal Processing, and in U.S. patent application Ser. No. 16/358,534, titled Foreign Object Detection Circuit Using Mutual Impedance Sensing, the entire contents of which are hereby incorporated by reference.
In an example operation of the multi-purpose detection circuit 100, the sense signal is selectively and sequentially applied to each of the plurality of inductive sense circuits 106 and to each of the plurality of the capacitive sense circuits 108 according to a time division multiplexing scheme and in a round robin fashion. The sense signal for driving an inductive sense circuit (e.g., inductive sense circuit 106a) is applied in a time interval (time slot) allocated to that sense circuit and has a maximum duration equal or shorter than the duration of the time slot. The time frame corresponding to the sum of time slots allocated to the plurality of inductive sense circuits 106 and capacitive sense circuits 108 is also referred herein as to the scan cycle or to the repetition period.
In an aspect to reduce the duration of the scan cycle, a first sense signal is selectively and sequentially applied to each of a portion of the plurality of inductive sense circuits 106 and capacitive sense circuits 108 and a second sense signal is concurrently, selectively and sequentially applied to each of the remaining portions of inductive and capacitive sense circuits. Concurrently applying two or more sense signals reduces the scan cycle and may result in a reduced detection latency with respect to FOD and LOD and in an increased position update rate with respect to PD (e.g., using the passive beaconing approach as previously described).
In an example implementation and operation of the multi-purpose detection circuit 100, the first and the at least one concurrently applied second sense signal are sinusoidal signals of the same frequency.
In another example implementation and operation of the multi-purpose detection circuit 100, the first and the at least one concurrently applied second sense signal are sinusoidal signals but differ in frequency.
In a further example implementation and operation of the multi-purpose detection circuit 100, each of the first and the at least one concurrent second sinusoidal sense signals as applied in time slots allocated to the same sense circuit (e.g., sense circuit 106a) start with the same phase (e.g., zero-phase). In some implementations using more than two current sense signals, starting sense signals in time slots allocated to the same sense circuit with the same phase may help to mitigate interference caused by intermodulation effects as described in U.S. patent application Ser. No. 16/392,464 titled Extended Foreign Object Detection Signal Processing, the entire contents of which are hereby incorporated by reference.
In some implementations and operations of the multi-purpose detection circuit 100, time slots of a scan cycle are reallocated based on some conditions (e.g., whether WPT is active or inactive). In an aspect, it may be desirable to reduce the detection latency with respect to LOD when WPT is active. Therefore, in an example operation, two or more time slots of a scan cycle are allocated to each of the capacitive sense circuits 108 when WPT is active. Conversely, the LOD function may not be required when WPT is inactive. Therefore, in an example operation, time slots of a scan cycle are only allocated to inductive sense circuits (e.g., to the plurality of inductive sense circuits 106) when WPT is inactive. In another example operation, two or more time slots of a scan cycle are allocated to each of the plurality of inductive sense circuits (e.g., inductive sense circuits 106) and one time slot is allocated to each of the plurality of capacitive sense circuits when WPT is inactive. This mode of operation may allow maintaining a limited LOD function when WPT is inactive (e.g., for purposes of monitoring proper functioning of the multi-purpose detection circuit 100 with respect to LOD). Moreover, the time spacing between time slots allocated to the same sense circuit in any of the scanning modes described above is maximized.
The descriptions of the circuits 500, 520, and 540 of
In some implementations, the sense signal is a high frequency signal with a spectrum substantially in the megahertz (MHz) range (e.g., in a frequency range from 2.5 MHz to 3.5 MHz). In other implementations, the sense signal is constraint to the frequency range from 3.155 MHz to 3.400 MHz for frequency regulatory reasons. In some geographic regions or countries, this frequency range may permit higher emission levels e.g., a magnetic field strength H<13.5 dBμA/m at 10 m from the radiating parts of the multi-purpose detection circuit 100 (e.g., from the inductive sense element array 107).
The ground symbol shown in the schematic diagrams of
The circuit 500 of
The sense circuit 501 comprises a single-coil sense element (e.g., sense coil 502) having an inductance L and an equivalent series resistance R, a series capacitor 504 having a capacitance Cs and an equivalent series resistance RCs electrically connected in series to the sense coil 502, and a parallel inductor 506 having an inductance Lp and an equivalent series resistance RLp electrically connected to the capacitor 504 in parallel to the measurement port 508. The circuit 500 further illustrates the sense signal current source 512 and the voltage measurement circuit 510 both electrically connected to the sense circuit 501 at the measurement port 508.
The equivalent series resistance R includes all electrical losses intrinsic to the sense coil 502 and extraneous losses as they may occur in its surrounding materials (e.g., the Litz wire of the WPT coil 202 and the ferrite of the wireless power transfer structure 200 where the sense coil 502 may be integrated). These materials may interact with the magnetic field as generated by the sense coil 502 causing losses.
The circuit 500 of
The sense circuit 501 may be configured to provide a local minimum in the impedance magnitude function |Z11,0(ω)| substantially at a nominal sense frequency, where Z11,0 refers to the impedance as presented by the sense circuit 501 at the measurement port 508 in absence of a foreign object, and w to the angular frequency. The minimum of the impedance magnitude is also referred to herein as the series resonance by definition and applies to the inductive sense circuits with reference to
In an example series resonant configuration of the sense circuit 501, the reactance of the series capacitor 504 substantially compensates for the reactance of the sense coil 502 at the nominal sense frequency providing an impedance Z11,0 that is substantially real (resistive). In this configuration, the inductance Lp of the parallel inductor 506 may be similar or larger than the inductance L of the sense coil 502. In other terms, the impedance magnitude of the parallel inductor 506 may be substantially (e.g., 10 times) higher than the impedance magnitude |Z11,0| as presented at the nominal sense frequency. In this configuration, the parallel inductor 506 may exert a negligible impact on the impedance |Z11,0| at the nominal sense frequency.
In an example parallel resonant configuration of the sense circuit 501, the reactance of the series capacitor 504 overcompensates for the reactance of the sense coil 502 at the nominal sense frequency. The residual capacitive susceptance of the series connection of the capacitor 504 and the sense coil 502 is substantially compensated for by the susceptance of the parallel inductor 506 providing an admittance Y11,0 that is substantially real (resistive). In this configuration, the inductance Lp of the parallel inductor 506 may be smaller, similar, or larger than the inductance L of the sense coil 502. Stated in other terms, the admittance magnitude of the parallel inductor 506 may be substantially (e.g., 20 times) higher than the admittance magnitude |Y11,0| as presented at the nominal sense frequency. In this configuration, the parallel inductor 506 exerts a significant impact on the admittance Y11,0 at the nominal sense frequency.
In some implementations, the parallel inductor 506 together with the series capacitor 504 are used for purposes of resonance tuning and impedance transformation e.g., to transform the impedance Z11 to match the sense circuit 501 with an operating impedance range as previously mentioned with reference to
Impedance transformation may be particularly effective, if the sense circuit 501 is configured for parallel resonance. More specifically, increasing the inductance ratio L/Lp, while maintaining series resonance at the nominal sense frequency, may substantially increase the admittance magnitude |Y11,0| at the nominal sense frequency. Therefore, in an aspect, the sense circuit 501 in the parallel resonant configuration may be considered as an alternative to the sense circuit 521 illustrated in
Increasing the inductance ratio L/Lp, while maintaining resonance at the nominal sense frequency, may also somewhat decrease the impedance magnitude |Z11,0| as presented at the nominal sense frequency in the series resonant configuration of the sense circuit 501. However, impedance transformation may be limited and far less effective than that of the series resonant configuration.
In another aspect of resonance tuning, the series capacitor 504 may include a variable capacitor whose capacitance Cs can be electronically controlled (e.g., a direct current (DC) controlled capacitor) forming a variable capacitor 504. In some implementations of the circuit 500, a variable capacitor 504 is used to compensate for a temperature drift, an aging, or a detuning of the sense circuit 701 caused by an external impact and to maintain its resonance substantially at the nominal sense frequency. Similarly, the parallel inductor 506 may include a variable inductor whose inductance Lp can be electronically controlled (e.g., a DC controlled inductor) forming a variable inductor 506. In a further aspect, the variable capacitor 504 and variable inductor 506 in combination are used to vary the impedance |Z11,0| of the sense circuit 501.
In yet another aspect, the series capacitor 504 in combination with the parallel inductor 506 form a 2nd order high pass filter to attenuate a low frequency disturbance component in the voltage V emanating from the voltage inductively coupled into the sense coil 502 by the magnetic and electric field as generated during wireless power transfer. This high pass filter may reduce dynamic range requirements of the voltage measurement circuit 510 and may also protect the voltage measurement circuit 510 and the current source 512 from being overloaded. Stated in other words, it may reduce non-linear distortion effects (e.g., signal clipping) in a voltage measurement circuit 510 with a limited dynamic range.
With reference to
In an aspect and for sinusoidal signals, a current source (e.g., current source 512) may be characterized by a quasi-ideal current source providing a source admittance magnitude |Ycs| substantially (e.g., at least 10 times) lower than the admittance magnitude |Y11| of the sense circuit 501 as presented at the measurement port 508 at the sense frequency. Analogously, the voltage measurement circuit 510 may be characterized by a quasi-ideal voltage measurement circuit with an admittance magnitude |Yvm| substantially (e.g., at least 10 times) lower than |Y11| at the sense frequency.
In a further aspect and for sinusoidal signals, a measurement circuit (e.g., measurement circuit 104 of
Ymc≈Ycs+Yvm (1)
Conversely, the quality of a measurement circuit (e.g., measurement circuit 104 of
Qmc≈|Y11|/|Ymc| (2)
Equation (2) may be used to assess the quality of a measurement circuit (e.g., measurement circuit 104 of
A more general definition of the quality of a measurement circuit (e.g., measurement circuit 104 of
Qmc≈|ΔV/V0|/|ΔI/I0| (3)
Above characterizations of the current source 512, the voltage measurement circuit 510, and the measurement circuit 104 may be generalized to non-sinusoidal sense signals, where the notions of complex impedance and complex amplitude may not directly apply. This may be accomplished by approximating the signal by a complex Fourier series and applying above characterizations to the individual frequency components of the complex Fourier series.
Other impedance measurement techniques may also be contemplated e.g., by applying a sinusoidal voltage, from a voltage source (e.g., voltage source 552 with reference to
Analogously to the current source voltage measurement technique, the voltage source 552 (sense signal voltage source 552) may be characterized by a quasi-ideal voltage source with a source impedance magnitude |Zvs| substantially (e.g., at least 10 times) lower than the impedance magnitude |Z11| of the sense circuit 501 as presented at the sense frequency. Analogously, the current measurement circuit 550 may be characterized by a quasi-ideal current measurement circuit with an impedance magnitude |Zcm| substantially (e.g., at least 10 times) lower than |Z11| at the sense frequency.
In a further aspect, a measurement circuit (e.g., measurement circuit 104 of
Zmc≈Zvs+Zcm (4)
Conversely, the quality of a measurement circuit (e.g., measurement circuit 104 of
Qmc≈|Z11|/|Zmc| (5)
Equation (5) may be used to assess the quality of a measurement circuit (e.g., measurement circuit 104 of
Other impedance measurement techniques may also include approaches where the sense circuit 501 is driven by a non-ideal source and the voltage V and the current I are measured e.g., using a quasi-ideal voltage measurement circuit and a quasi-ideal current measurement circuit, respectively.
Further, in some implementations, measurement of the voltage V and thus of the impedance Z11 may be affected by noise and other disturbance signals reducing a detection sensitivity of the multi-purpose detection circuit 100. The noise may include circuit intrinsic noise as generated in active and passive components of the circuit 500 of
Moreover, in implementations employing a selective voltage measurement circuit 510 as discussed above, the sense signal waveform as generated by the current source 512 and the corresponding filter of the voltage measurement circuit 510 are adapted e.g., to improve the SNR and consequently to improve the detection sensitivity. Therefore, in some implementations, the voltage measurement circuit 510 also includes a noise analyzer (e.g., included in the signal processing circuit 408 with reference to
With reference to
Presence of an object (e.g., object 110) may be determined if ΔZ satisfies certain criteria (e.g., magnitude |ΔZ| exceeding a detection threshold, angle arg{ΔZ} being within a certain range). Though not shown in
In an implementation of the circuit 500 based on measuring the admittance Y11, presence of an object (e.g., object 110, 112, 114, or vehicle 330) may cause a change ΔY with respect to the admittance Y11,0 as measured in absence of a foreign object. Analogously, presence of an object (e.g., object 110) may be determined if ΔY satisfies certain criteria (e.g., magnitude |ΔY| exceeding a detection threshold, angle arg{ΔY}) being within a certain range).
Using a quasi-ideal current source (e.g., the current source 512), a change ΔZ in the impedance Z11 (e.g., due to the presence of the object 110) manifests in a change ΔV in the voltage V while the current I0 remains substantially unaffected. Therefore, measuring the complex voltage V may be equivalent to measuring the complex impedance Z11. In other words, the complex voltage V may be indicative of the complex impedance Z11 and there may be no requirement for additionally measuring the current I0 thus reducing complexity of the measurement circuit (e.g., measurement circuit 104 of
In an aspect, it may be useful to define the normalized reflected impedance of an object (e.g., object 110) in the sense coil's 502 having a reactance ω L as:
ΔZ′r=(Z−j ω L)/(ω L)=ΔZr/(ω L) (6)
where Z defines the sense coil's 502 impedance in presence of an object (e.g., object 110). Analogously, the normalized reflected admittance ΔY′r may be defined as:
ΔYr′=(Y−(1/(j ω L))ω L=ΔYr ω L (7)
where Y and ΔYr denote the sense coil's 502 admittance in presence of an object (e.g., object 110) and the reflected admittance of the object, respectively. The normalized reflected impedance ΔZ′r or the normalized reflected admittance ΔY′r determine the impact of an object (e.g., object 110) on the sense coil's 502 impedance or admittance, respectively. Its magnitude |ΔZ′r| or |ΔY′r| may be related to the size, the position, and orientation of the object relative to the sense coil 502.
In a further aspect, it may be useful to define the normalized impedance change of a one-port sense circuit (e.g., sense circuit 501 of
ΔZ′=(Z11−Z11,0)/|Z11,0|=ΔZ/|Z11,0| (8)
and analogously, the normalized admittance change as:
ΔY′=(Y11−Y11,0)/|Y11,0|=ΔY/|Y11,0| (9)
also referred herein as to the fractional change ΔZ′ (or ΔY′). The fractional change ΔZ′ (or ΔY′) caused by a defined test object (e.g., object 110) placed at a defined position relative to the sense coil 502 may relate to the detection sensitivity of an object detection circuit (e.g., the multi-purpose detection circuit 100 of
ΔSNR=|ΔV|/Vn (10)
with Vn (not indicated in
As non-limiting examples, the normalized reflected impedance ΔZ′r of an object (e.g., object 110) and thus the related fractional change ΔZ′ may be increased by optimizing the design of the sense coil 502 with respect to its geometry and its integration into the wireless power transfer structure (e.g., wireless power transfer structure 200 with reference to
As further analyzed and discussed below with reference to
In a further aspect of the multi-purpose detection circuit 100, variations in temperature e.g., of the sense coil 502 may result in thermal drift of the impedance Z11 as measured at the measurement port 508. In some implementations, the sense coil's inductance L and equivalent series resistance R, the series capacitor's 504 capacitance, and the parallel inductor's 506 inductance Lp and equivalent series resistance RLp may be subjected to thermal drift. Thermal drift effects may deteriorate the detection sensitivity of the multi-purpose detection circuit 100. Considering the physical nature of temperature drifts in a tuned sense circuit (e.g., sense circuit 501), it may be meaningful to define a temperature sensitivity Sϑ for the real and imaginary part, separately, as the ratios:
Re{S
ϑ
}=Re{ΔZ
ϑ
′}/Re{ΔZ′} (11)
Im{S
ϑ
}=Im{ΔZ
ϑ
′}/Im{ΔZ′} (12)
where ΔZϑ′ denotes the fractional impedance change due to a defined temperature change Δϑ and ΔZ′ the fractional impedance change due to presence of a test object (e.g., object 110) at a defined position relative to the sense coil 502. The fractional change ΔZϑ′ may be considered the complex temperature coefficient of a sense circuit (e.g., sense circuit 501). The temperature sensitivity Sϑ may also be expressed in terms of the fractional admittance changes ΔYϑ′ and ΔY′.
In yet another aspect of the multi-purpose detection circuit 100, it may be desirable to discriminate between certain categories of objects e.g., between foreign metallic objects (e.g., object 110), non-living non-conductive objects (e.g., object 112), and living objects (e.g., object 114). In another aspect, it may also be desirable to discriminate e.g., between foreign metallic objects (e.g., object 110) and the vehicle 330 with reference to
In some implementations and configurations of the circuit 500 of
In an aspect of reducing an error in the measurement of the angle arg{ΔZ}, some implementations of a multipurpose detection circuit 100 employ a phase calibration of the analog circuitry (e.g., the analog front end portion of the measurement circuit 104 with reference to
Reactance compensation (resonance tuning) in the sense circuit 501 produces a local extremum (minimum or maximum) in the impedance magnitude function |Z11,0(ω)| and hence in the voltage magnitude |V| across the measurement port 508. Therefore, reactance compensation provides a mean to calibrate the voltage measurement circuit 510 and hence the impedance measurement with respect to the angle arg{ΔZ}.
In a first step of an example calibration procedure applicable to the series resonant configuration of the circuit 500 of
V
cal
=V
uncal exp(−j arg{Vuncal}) (13)
where Vcal refers to the complex voltage value as determined by the calibrated voltage measurement circuit 510.
Applying the angle correction of Equation (13), an object (e.g., object 110) reflecting an impedance ΔZr that is imaginary (reactive) may cause a measured voltage change ΔVcal that is substantially imaginary. Nevertheless, a small residual error may remain in the angle arg{ΔVcal} due to the impact of the parallel inductor 506 and the electrical losses in the sense circuit 501. The residual angle error of an example series resonant configuration of the circuit 500 and for an example object 110 is provided in TABLE 2.
In some implementations, the residual error described above is reduced by configuring the parallel inductor 506 with an inductance Lp whose impedance ZLp is substantially larger (e.g., 10 times larger) than the series resonant resistance of the sense circuit 501. In other implementations, the residual error is reduced by measuring the impedance Z11,0 at two or more substantially different frequencies and by determining the elements of an equivalent circuit model of the sense circuit 501 (e.g., the equivalent circuit model illustrated in
In an implementation of the multipurpose detection circuit 100 using a plurality of inductive sense circuits (e.g., inductive sense circuits 106a, 106b, . . . , 106n), each including a respective inductive sense element (e.g., inductive sense element 107a, 107b, . . . , 107n) of an array (e.g., array 107), a further residual error may be caused by a parasitic resonance effect of sense circuits associated to adjacent inductive sense elements. More precisely, a residual error in a first sense circuit (e.g., inductive sense circuit 106a) including a first inductive sense element (e.g., inductive sense element 107a) may be caused by a parasitic resonance effect of at least one second inductive sense circuit (e.g., inductive sense circuit 106b) including a second inductive sense element (e.g., inductive sense element 107b) that is located adjacent to the first inductive sense element.
Therefore, in some implementations of the multipurpose detection circuit 100, the measurement accuracy of the angle arg{ΔZ} and thus of the angle arg{ΔZr} is increased by an optimized design of the sense coil 502 and by introducing some spacing between adjacent sense coils 502 of an array (e.g., array 107).
In an implementation configured for parallel resonance as defined above, the circuit 500 may be configured to measure the admittance Y11 and corresponding changes ΔY of Y11 as caused by the object 110, 112, 114, or vehicle 330. In this case, the admittance change ΔY may be indicative of the reflected impedance ΔZr as previously introduced. As discussed above with reference to the series resonant configuration, the angle arg{ΔY} may be subjected to an error and therefore may require calibration to reduce an error in the measurement of the angle arg{ΔY} and thus of the angle arg{ΔZr}.
In an implementation configured for parallel resonance, the circuit 500 may be calibrated analogously to the series resonant configuration using the local minimum of the admittance function |Y11,0(ω)| where susceptance compensation occurs.
In a first step of an example calibration procedure applicable to the parallel resonant configuration of the circuit 500 of
Applying the angle correction of Equation (13), an object (e.g., object 110) reflecting an impedance ΔZr that is imaginary (reactive) may result in a measured voltage change ΔVcal that is substantially imaginary. A residual error may remain in the angle arg{ΔVcal} due to the transformation of ΔZr to ΔY in the lossy sense circuit 501. The residual angle error of an example parallel resonant configuration of the circuit 500 and for example reflected impedance ΔZr is provided in TABLE 2.
In an example implementation, the residual error due to the transformation of ΔZr to ΔY is reduced by measuring the admittance Y11,0 at two or more substantially different frequencies, supposing absence of a foreign object, and by determining the elements of an equivalent circuit model (e.g., the equivalent circuit model of
The series and the parallel resonant configuration of the circuit 500 of
The circuit 520 of
The sense circuit 521 comprises the single-coil inductive sense element (e.g., sense coil 502) having the inductance L with reference to
Though not indicated in
The sense circuit 521 may be configured to provide a local minimum in the impedance magnitude function |Z11,0(ω)| (series resonance) substantially at the nominal sense frequency. Alternatively, it may be configured to provide a local minimum in the admittance magnitude function |Y11,0(ω)| (parallel resonance) substantially at the nominal sense frequency using the transformer's 526 secondary referred main inductance Lm in a manner similar to using the inductance Lp as described above with reference to
In an example series resonant configuration of the sense circuit 521, the reactance of the series capacitor 504 substantially compensates for the reactance of the sense coil 502 at the nominal sense frequency providing an impedance Z11,0 at the measurement port 528 that is substantially real (resistive). The reactance of the series capacitor 524 also compensates for the reactance of the transformer's 526 secondary referred leakage inductance Lσ with reference to
In an example parallel resonant configuration of the sense circuit 521, the reactance of the series capacitor 524 overcompensates for the sum reactance of the sense coil 502 and the transformer's 526 leakage inductance Lσ at the nominal sense frequency. The residual capacitive susceptance of the series connection of the capacitor 524, the sense coil 502 and the transformer's leakage inductance Lσ is substantially compensated for by the susceptance of the transformer's 526 secondary referred inductance Lm providing an admittance Y11,0 that is substantially real (resistive). In this configuration, the inductance Lm may be smaller, similar, or larger than the inductance L of the sense coil 502. Stated in other terms, the primary referred open-circuit admittance of the transformer 526 may be substantially (e.g., 20 times) higher than the admittance magnitude |Y11,0| as presented at the nominal sense frequency. In this configuration and apart from the admittance transformation, the transformer 526 exerts a significant impact on the admittance Y11,0 at the nominal sense frequency.
The transformer 526 may serve for various purposes. In some implementations, the transformer 526 is a nT:1 transformer with nT≠1 used at least for impedance transformation e.g., to match the impedance magnitude |Z11| of the sense circuit 521 with an operating impedance range as previously mentioned with reference to
Apart from the transformation ratio nT:1, the inductance ratio L/Lm may be an additional parameter to match the admittance magnitude |Y11,0| of the parallel resonant configuration with an operating admittance range of the multi-purpose detection circuit 100 in a manner similar to the parameter L/Lp in the circuit 500 of
In a further aspect, the series capacitor 524 in combination with the transformer's 526 main inductance Lm form a 2nd order high pass filter to attenuate a low frequency disturbance component in the voltage V for purposes as previously discussed in connection with
The losses of the transformer 526 and its leakage inductance Lσ may somewhat reduce the fractional change ΔZ (or ΔY) of the sense circuit 521 if compared to the transformerless sense circuit 501 of
The circuit 540 of
The circuit 540 may be considered an electrically dual circuit of the circuit 500 of
In another aspect, the sense circuit 541 may also include a transformer (not shown herein) e.g., electrically connected between the measurement port 548 and the capacitor 546 e.g., for purposes of balancing.
Though not indicated in
As with the circuit 500 of
In an example parallel resonant configuration of the sense circuit 541, the susceptance of the parallel capacitor 544 substantially compensates for the susceptance of the sense coil 502 at the nominal sense frequency providing an admittance Y11,0 that is substantially real (resistive). In this configuration, the capacitance Cs of the series capacitor 546 may be similar or larger than the capacitance Cp of the parallel capacitor 544. Stated in other terms, the admittance magnitude of the series capacitor 546 may be substantially (e.g., 10 times) higher than the admittance magnitude |Y11,0| as presented at the nominal sense frequency. In this configuration, the series capacitor 546 may exert a negligible impact on the admittance |Y11,0| at the nominal sense frequency.
In an example series resonant configuration of the sense circuit 541, the susceptance of the parallel capacitor 544 undercompensates for the susceptance of the sense coil 502 at the nominal sense frequency. The residual inductive reactance of the parallel connection of the capacitor 544 and the sense coil 502 is substantially compensated for by the reactance of the series capacitor 546 providing an impedance Z11,0 that is substantially real (resistive). In this configuration, the capacitance Cs of the series capacitor 546 may be smaller, similar, or larger than the capacitance Cp of the parallel capacitor 544. Stated in other terms, the impedance magnitude of the series capacitor 546 may be substantially (e.g., 20 times) higher than the impedance magnitude |Z11,0| as presented at the nominal sense frequency. In this configuration, the series capacitor 546 exerts a significant impact on the impedance Z11,0 at the nominal sense frequency.
In some implementations, the series capacitor 546 together with the parallel capacitor 544 are used for purposes of resonance tuning and impedance transformation e.g., to transform the impedance Z11 to match the sense circuit 541 with an operating impedance range as previously mentioned with reference to
Impedance transformation may be particularly effective, if the sense circuit 541 is configured for series resonance. More specifically, increasing the capacitance ratio Cp/Cs, while maintaining series resonance at the nominal sense frequency, may substantially increase the impedance magnitude |Z11,0| at the nominal sense frequency. Therefore, in an aspect, the sense circuit 541 in the series resonant configuration may be considered as an alternative to the sense circuit 521 of
Increasing the capacitance ratio Cp/Cs, while maintaining resonance at the nominal sense frequency, may also somewhat decrease the admittance magnitude |Y11,0| as presented at the nominal sense frequency in the parallel resonant configuration of the sense circuit 541. However, impedance transformation may be limited and far less effective than that of the series resonant configuration.
In a further aspect, the sense circuit 541 due to the series capacitor 546 in conjunction with the voltage source current measurement technique provides a high pass filter characteristic to attenuate a low frequency disturbance component in the current/emanating from the voltage inductively coupled into the sense coil 502 by the magnetic and electric field as generated during wireless power transfer. This high pass filter may reduce dynamic range requirements of the current measurement circuit 550 and may also protect the current measurement circuit 550 and the voltage source 552 from being overloaded. Stated in other terms, it may reduce non-linear distortion effects (e.g., signal clipping) in a current measurement circuit 550 with a limited dynamic range.
With reference to
In some implementations, the voltage source 552 may be characterized by a quasi-ideal voltage source providing a source impedance whose magnitude is substantially (e.g., 10 times) lower than the magnitude of the impedance |Z11| of the sense circuit 541 as presented at the sense frequency. Analogously, the current measurement circuit 550 may be characterized by a quasi-ideal current measurement circuit with an impedance magnitude substantially (e.g., 10 times) lower than the impedance magnitude |Z11| at the sense frequency.
Above characterizations of the voltage source 552 and the current measurement circuit 550 may be generalized to non-sinusoidal sense signals as previously discussed with reference to
Other impedance measurement techniques may also be contemplated e.g., by applying a sinusoidal current, from the current source 512, with a defined current I0 (amplitude and phase) to the sense circuit 541 and by measuring the complex voltage V (amplitude and phase) at the measurement port 548 using the voltage measurement circuit 510 as previously discussed with reference to
Further, in some implementations, measurement of the current I and thus of the impedance Z11 may be affected by noise and other disturbance signals reducing a detection sensitivity of the multi-purpose detection circuit 100 as previously discussed with reference to
With reference to
Presence of an object (e.g., object 110) may be determined if ΔY satisfies certain criteria (e.g., magnitude |ΔY| exceeding a detection threshold, angle arg{ΔY} being within a certain range). In an implementation of the circuit 540 where the impedance Z11 is measured as previously mentioned in connection with the series resonance, presence of an object (e.g., object 110) may cause a change ΔZ with respect to the impedance Z11,0.
Using a quasi-ideal voltage source 552, a change ΔY in the admittance Y11 (e.g., due to presence of the object 110) manifests in a change ΔI in the current I while the voltage V0 remains substantially unaffected. Therefore, measuring the complex current I may be equivalent to measuring the complex admittance Y11. In other words, the complex current I may be indicative of the complex admittance Y11 and there may be no requirement for additionally measuring the voltage V0 thus reducing complexity of the measurement circuit (e.g., measurement circuit 104 of
The fractional change ΔY′ (or ΔZ′) as defined by Equations (8) and (9) and with respect to a defined test object (e.g., object 110) placed at a defined position relative to the sense coil 502 may relate to the detection sensitivity of an object detection circuit (e.g., the multi-purpose detection circuit 100 of
ΔSNR=|ΔI|/In (14)
with In referring to the noise component in the current I. In another aspect, increasing the fractional change may reduce dynamic range requirements of the current measurement circuit 550.
As non-limiting examples, the fractional change may be increased by optimizing the design of the sense coil 502 with respect to its geometry and its integration into the wireless power transfer structure (e.g., wireless power transfer structure 200 with reference to
As previously discussed with reference to the circuit 500 of
In some implementations and configurations of the circuit 540 of
Susceptance compensation in the sense circuit 541 exhibiting a local extremum (minimum or maximum) in the admittance magnitude function |Y11,0(ω)| and hence in the resulting current magnitude |I| at the measurement port 548 provides a mean to calibrate the current measurement circuit 550 and hence the admittance measurement with respect to the angle arg{ΔY}.
In a first step of an example calibration procedure applicable to the parallel resonant configuration of the circuit 540 of
I
cal
=I
uncal exp(−j arg{Iuncal}) (15)
where Ical refers to the complex current value as determined by the calibrated current measurement circuit 550.
Applying the angle correction of Equation (15), an object (e.g., object 110) reflecting an admittance ΔYr that is imaginary (reactive) may result in a measured current change ΔIcal that is substantially imaginary. Nevertheless, a residual error may remain in the angle arg{ΔIcal} due to the impact of the series capacitor 546 and the electrical losses in the sense circuit 541. The residual angle error of an example parallel resonant configuration of the circuit 540 and for an example object 110 is provided in TABLE 2.
In some implementations, the residual error is reduced by configuring the series capacitor 546 with a capacitance Cs whose admittance YCs is substantially larger (e.g., 10 times larger) than the parallel resonant conductance of the sense circuit 541. In other implementations, the residual error is reduced by computing the error in the measured angle arg{ΔY} by estimating parameters of the sense circuit 541 (e.g., the Q-factor) at the actual sense frequency. In further implementations, the residual error is reduced by measuring the admittance Y11,0 at two or more substantially different frequencies and by determining the elements of an equivalent circuit model of the sense circuit 541 (e.g., the equivalent circuit model illustrated in
In an implementation configured for series resonance as defined above, the circuit 540 may be configured to measure the impedance Z11 and corresponding changes ΔZ of Z11 as caused by the object 110, 112, 114, or vehicle 330. In this case, the impedance change ΔZ may be indicative of the reflected admittance ΔYr as previously introduced. As discussed above with reference to the parallel resonant configuration, the angle arg{ΔZ} may be subjected to an error and therefore may require calibration to reduce an error in the measurement of the angle arg{ΔZ} and thus of the angle arg{ΔYr}.
In an implementation configured for series resonance, the circuit 540 may be calibrated analogously to the parallel resonant configuration however using the local minimum of the impedance function |Z11,0(ω)| where reactance compensation occurs.
In a first step of an example calibration procedure applicable to the series resonant configuration of the circuit 540 of
Applying the angle correction of Equation (15), an object (e.g., object 110) reflecting an admittance ΔYr that is imaginary (reactive) may result in a measured current change ΔIcal that is substantially imaginary. Nevertheless, a residual error may remain in the angle arg{ΔIcal} due to the transformation of ΔYr to ΔZ in the lossy sense circuit 541. The residual angle error of an example series resonant configuration of the circuit 540 and for an example object 110 is provided in TABLE 2.
In an example implementation, the residual error due to the transformation of ΔYr to ΔZ is reduced by measuring the impedance Z11,0 at two or more substantially different frequencies, supposing absence of a foreign object, and by determining the elements of an equivalent circuit model (e.g., the equivalent circuit model of
The series and the parallel resonant configuration of the circuit 540 of
The circuit 560 of
The sense circuit 561 of
Though not indicated in
An inductive coupling factor:
k
L
=L
M(L1 L2)−1/2 (16)
may be defined for the two-port inductive sense element 562. Further, a two-port inductive sense element (e.g., inductive sense element 562 of
Vind,1=jω LM I2 (17)
Vind,2=jω LM I1 (18)
representing the voltage induced into the first and second sense coil, respectively as indicated in
In some implementations, the reactance of Cs,1 substantially compensates for the reactance of L1 providing a local impedance minimum |Z11| (series resonance) substantially at the nominal sense frequency, while the reactance of Cs,2 substantially compensates for the reactance of L2 providing a local impedance minimum |Z22| (series resonance) substantially at the nominal sense frequency.
In another implementation, the sense circuit 561 is configured to provide a local minimum of the admittance magnitude functions |Y11(ω)| and |Y22(ω)| (parallel resonance) substantially at the nominal sense frequency.
In a further implementation, the sense circuit 561 is configured to provide a local minimum of the admittance magnitude function |Y11(ω)| (parallel resonance) and a local minimum of the impedance magnitude function |Z22(ω)| (series resonance) substantially at the nominal sense frequency.
In yet another implementation, the sense circuit 561 is configured to provide a local minimum of the impedance magnitude function |Z11(ω)| (series resonance) and a local minimum of the admittance magnitude function |Y22(ω)| (parallel resonance) substantially at the nominal sense frequency.
In implementations configured for primary-side and secondary-side series resonance, the reactance of the parallel inductors 566 and 567 is substantially higher than the impedance magnitudes |Z11| and |Z22|, respectively, of the sense circuit 561 at the nominal sense frequency.
In a further example implementation, at least one of the series capacitors 564 and 565 is omitted and the sense circuit 561 is operated as a non-resonant or partially resonant circuit.
In a further aspect, the first series capacitor 564 in combination with the first parallel inductor 566 form a 2nd order high pass filter to attenuate a low frequency disturbance component in the voltage V1. Likewise, the second series capacitor 565 in combination with the second parallel inductor 567 form a 2nd order high pass filter to attenuate a low frequency disturbance component in the voltage V2 for purposes as previously discussed in connection with
With reference to
As with the circuit 500 of
Though not shown herein, other transimpedance measurement techniques such as the voltage source current measurement technique or any other combination may apply (e.g., a current source current measurement technique). In some implementations (also not shown herein), at least one of the impedances Z11 and Z22 of the sense circuit 561 is additionally measured to the transimpedance Z21 (e.g., using one or more of the techniques as previously discussed with reference to
Moreover, at least one of an impedance transformation and balancing may apply to at least one of the primary-side and secondary-side of the sense circuit 561 (not shown herein). More specifically, with reference to the circuit 521 of
With reference to
Using a quasi-ideal current source 512, a change ΔZ in the transimpedance Z21 (e.g., due to presence of the object 110) manifests in a change ΔV in the voltage V2 while the current I0,1 remains substantially unaffected. Therefore, measuring the complex voltage V2 may be equivalent to measuring the complex transimpedance Z21. In other words, the complex voltage V2 may be indicative of the complex transimpedance Z21 and there may be no requirement for additionally measuring the current I0,1 thus reducing complexity of the measurement circuit (e.g., measurement circuit 104 of
In an aspect, it may be useful to define the normalized transimpedance change of a two-port sense circuit (e.g., sense circuit 561 of
ΔZ′=(Z21−Z21,0)/|Z21,0|=ΔZ/|Z21,0| (19)
and, correspondingly, the normalized transadmittance change as:
ΔY′=(Y21−Y21,0)/|Y21,0|=ΔY/|Y21,0| (20)
also referred herein as to the fractional change. As with the circuit 501 of
ΔSNRV=|ΔV|/Vn (21)
with Vn referring to the noise component in the voltage V2.
As non-limiting examples, the fractional change may be increased by optimizing the design and arrangement of the sense coils 562a and 562b, their integration into the wireless power transfer structure (e.g., wireless power transfer structure 200 with reference to
In an example implementation, the fractional change ΔZ′ (or ΔY′) is substantially increased by configuring and arranging the sense coils 562a and 562b such that the mutual inductance LM substantially vanishes in absence of a foreign object, resulting in a transimpedance |Z21,0| that is substantially zero. Example implementations of double sense coil arrangements providing a substantially zero mutual inductance LM are described in U.S. patent application Ser. No. 16/358,534, titled Foreign Object Detection Circuit Using Mutual Impedance Sensing, the entire contents of which are hereby incorporated by reference.
The circuit 580 of
The sense circuit 581 of
Though not indicated in
In an example implementation, the sense coils 562a and 562b are tightly coupled resulting in an inductive coupling factor kL as defined by Equation (16) that is near unity (kL≈<1). Example implementations of double-coil inductive sense elements 562 providing an inductive coupling factor kL near unity are described in U.S. patent application Ser. No. 16/358,534, titled Foreign Object Detection Circuit Using Mutual Impedance Sensing, the entire contents of which are hereby incorporated by reference.
The sense circuit 581 may be configured to provide a local minimum in the transimpedance magnitude function |Z21,0(ω)| (series resonance) substantially at a nominal sense frequency. Alternatively, the sense circuit 581 may be configured to provide a local minimum in the transadmittance magnitude function |Y11,0(ω)| substantially at the nominal sense frequency.
In an example series resonant configuration of the sense circuit 581 using an inductive sense element 562 with kL≈<1, the reactance of the series capacitor 584 substantially compensates for the reactance of the mutual inductance LM providing a local minimum in the transimpedance magnitude function |Z21,0(ω)| (series resonance) substantially at the nominal sense frequency. The principle of mutual reactance compensation may become more evident by contemplating
In this series resonant configuration, the inductance Lp,1 and Lp,2 of the parallel inductor 586 and 587, respectively, may be similar or larger than the inductance L1 and L2 of the sense coils 562a and 562b, respectively. Stated in other terms, the impedance magnitude of the parallel inductor 586 and 587 may be substantially higher than the impedance magnitude |Z11| and |Z22|, respectively, of the sense circuit 581 at the nominal sense frequency. In this configuration, the parallel inductors 586 and 587 may exert a negligible impact on the impedances and transimpedance |Z11|, |Z22|, and |Z21|, respectively, at the nominal sense frequency.
In an example parallel resonant configuration of the sense circuit 581 using an inductive sense element 562 with kL≈<1, the reactance of the series capacitor 584 overcompensates for the reactance of the mutual inductance LM at the nominal sense frequency. The residual capacitive susceptance of the series connection of the capacitor 584 and the mutual inductance LM is substantially compensated for by the susceptance of the parallel inductors 586 and 587 providing a transadmittance Y21,0 that is substantially real (resistive). In this configuration, the inductances Lp,1 and Lp,2 of the parallel inductors 586 and 587, respectively, may be smaller, similar, or larger than the inductance L1 and L2 of the sense coils 562a and 562b, respectively. Stated in other terms, the admittance magnitude of each of the parallel inductors 586 and 587 may be substantially (e.g., 20 times) higher than the admittance magnitudes |Y11| and |Y22|, respectively, as presented at the nominal sense frequency. In this configuration, the parallel inductors 586 and 587 exert a significant impact on the admittance and transadmittance magnitudes |Y11|, |Y22|, and |Y21|, respectively, at the nominal sense frequency.
In some implementations, the parallel inductors 586 and 587 together with the series capacitor 584 are used for purposes of resonance tuning and transimpedance transformation, e.g., to transform the transimpedance Z21 to match the sense circuit 581 with an operating transimpedance range as previously mentioned with reference to
Impedance and transimpedance transformation may be particularly effective, if the sense circuit 581 is configured for parallel resonance. More specifically, increasing the inductance ratios L1/Lp,1 and L2/Lp,2, while maintaining parallel resonance at the nominal sense frequency, may substantially increase the admittance magnitudes |Y11,0|, Y22,0|, and |Y21,0| of the parallel resonant configuration at the nominal sense frequency.
increasing the inductance ratios L1/Lp,1 and L2/Lp,2, while maintaining resonance at the nominal sense frequency, may also somewhat decrease the impedance magnitudes |Z11,0|, |Z22,0|, and |Z21,0| as presented at the nominal sense frequency in the series resonant configuration of the sense circuit 581. However, impedance transformation may be limited and far less effective than that of the parallel resonant configuration.
In a further aspect, the series capacitor 584 in combination with the first parallel inductor 586 form a 2nd order high pass filter to attenuate a low frequency disturbance component in the voltage V1. Likewise, the second series capacitor 584 in combination with the second parallel inductor 587 form a 2nd order high pass filter to attenuate a low frequency disturbance component in the voltage V2 for purposes as previously discussed in connection with
With reference to
With reference to
As with the sense circuit 561 of
QM≈ω LM/RM (22)
of the inductive sense element 562 with respect to LM and RM.
Moreover, the impedance change ΔZ may reflect electrical properties of the object 110, 112, or 114 as discussed with reference to the circuit 500 of
The circuit 580 of
For purposes of comparison, an identical sense coil 502 an equal sense coil current level |IL| is assumed for both configurations of the circuits 500, 520, and 540, though practical implementations configured for parallel resonance may prefer a sense coil 502 with a lower inductance L. Comparing SNRs at the same sense coil current level |IL| may be meaningful e.g., if the current level |IL| is emission or power constraint. Further, it is assumed that the circuits in both configurations are adjusted to a common resonant frequency substantially corresponding with the nominal sense frequency that is substantially higher than the WPT operating frequency.
The equivalent circuit model 500-1 as illustrated in
To analyze the series and parallel resonant configuration of the circuit 500 of
ω L>>R (23)
ω Lp>>RLp (24)
|ΔZr|<<R (25)
are made for a frequency range about the resonant frequency.
In an implementation configured for series resonance and with a reactance:
ω Lp>>|Z11| (26)
in a frequency range about the series resonant frequency, the impedance Z11 at the measurement port 508 of the circuit 500 of
Z11≈R+(jω Cs)−1+jω L+ΔZr (27)
In absence of a foreign object, a local minimum of |Z11,0(ω)| (series resonance) occurs substantially at an angular frequency ω satisfying:
(jω Cs)−1+jω L≈0 (28)
yielding the series resonant angular frequency:
ωs≈(L Cs)−1/2 (29)
At this frequency, the impedance Z11,0 becomes substantially real:
Z
11,0
≈Re{Z
11,0
}=R
s
≈R (30)
with Rs denoting the series resonant resistance, while the impedance Z11 in presence of an object (e.g., object 110) is approximately:
Z11≈Rs+ΔZ≈R+ΔZr (31)
with ΔZr referring to the reflected impedance as previously defined with reference to
Applying Equations (30) and (31) to Equation (8), the fractional change ΔZ′ for the series resonant configuration of the circuit 500 of
ΔZ′≈ΔZr/Rs≈ΔZr/R (32)
Using the definition of Equation (6) of the normalized reflected impedance ΔZ′r at ωs and defining a Q-factor of the series resonant configuration of the circuit 500 of
Qs≈ωs L/Rs (33)
which approximately equals the Q-factor of the sense coil 502 at the series resonant frequency:
Q=ω
s
L/R≈Q
s (34)
the fractional change may also be written in terms of ΔZ′r and Qs as:
ΔZ′≈Qs ΔZ′r (35)
Equation (35) shows that reactance compensation in the series resonant configuration of the circuit 500 of
To analyze the parallel resonant configuration of the circuit 500 of
|ω L−(ω Cs)−1|>>R (36)
is made for a frequency range about the resonant frequency. The admittance Y11 at the measurement port 508 in presence of an object (e.g., object 110) may be expressed as:
Y
11=(RLp+jω Lp)−1+(R+jω L+(jω Cs)−1+ΔZr)−1 (37)
Using Equations (23), (24), (25), (36) and the approximation:
1/(1+x)≈1−x (38)
valid for |x|<<1, where x may be a complex number, and neglecting insignificant terms, the admittance Y11 of Equation (37) may be approximated as:
Y11≈(jω Lp)−1+RLp(ω Lp)−2+(jω L+(jω Cs)−1)−1+(R+ΔZr)(ω L−(ω Cs)−1)−2 (39)
In absence of a foreign object, a local minimum of |Y11,0(ω)| (parallel resonance) occurs substantially at an angular frequency ω satisfying:
(jω Cs)−1+jω(L+Lp)≈0 (40)
yielding for the parallel resonant angular frequency:
ωp≈(Cs(L+Lp))−1/2 (41)
At this frequency, the admittance Y11,0 becomes substantially real:
Y
11,0
≈Re{Y
11,0
}=G
p=(R+RLp)/(ωp Lp)2 (42)
with Gp denoting the parallel resonant conductance, while the admittance Y11 in presence of an object (e.g., object 110) is approximately:
Y11≈Gp+ΔY≈(R+RLp+ΔZr)/(ωp Lp)2 (43)
where:
ΔY≈ΔZr/(ωp Lp)2 (44)
defines the admittance change due to the object.
Further, defining the Q-factor of the sense coil 502:
Q=ω
p
L/R (45)
and the Q-factor of the parallel inductor 506:
Q
Lp=ωp Lp/RLp (46)
the inductance ratio:
n
L
=L/L
p (47)
the admittance Y11,0 of Equation (42) at ωp may be expressed as:
Y11,0≈Gp≈nL((Q/QLp)+nL)/(Q ωp L) (48)
For the case QLp=Q and nL>>1, the parallel resonant conductance Gp becomes approximately:
Gp≈nL2/(Q ωp L) (49)
According to Equation (48), the admittance Y11 at ωp of the sense circuit 501 of
Applying Equations (42) and (44) to Equation (9), the fractional change ΔY′ for the parallel resonant configuration of the circuit 500 of
ΔY′=ΔY/Gp≈ΔZr/(R+RLp) (50)
showing that the admittance change ΔY is substantially proportional to the reflected impedance ΔZr. Therefore, the angle arg{ΔY} of the measured admittance change ΔY is indicative of the angle arg{ΔZr}. As previously described with reference the circuit 500 of
Defining the Q-factor of the parallel resonant configuration of the sense circuit 501 of
Q
p=ωp(Lp+L)/(R+RLp)≈nL(1+nL)/(Gp ωpL) (51)
which may be also expressed in terms of the Q-factors Q and QLp as:
Q
p
=Q(1+nL)/((Q/QLp)+nL) (52)
using the definition of Equation (6) of the normalized reflected impedance at ωp, and applying Equations (51) and (47) to Equation (50), the fractional admittance change ΔY′ may also be written as:
ΔY′≈Qp ΔZ′r nL/(1+nL) (53)
For the case QLp=Q, the fractional change becomes:
ΔY′≈Q ΔZ′r nL/(1+nL) (54)
ΔY′≈Q ΔZ′r (55)
The fractional change |ΔY′| of the parallel resonant configuration of the circuit 500 of
In a further aspect, the drive current level I0, the resulting voltage V at the measurement port 508, and the drive power level P are considered. In some implementations based on the circuit 500 of
I0≈|IL| (56)
resulting in a voltage across the measurement port 508:
V≈|Z11,0|I0≈R|IL| (57)
and in a drive power level:
P≈V I0≈R|IL|2 (58)
Using Equations (42), (47), and (51) for the parallel resonant configuration of the circuit 500 of
I0≈|IL|(1+nL)/Qp≈|IL|((Q/QLp)+nL)/Q (59)
The voltage across the measurement port 508 becomes approximately:
V≈I0/|Y11,0|≈I0/Gp≈|IL|/(Q G nL) (60)
and the drive power:
P≈I0 V≈(|IL|2/G)((Q/QLp)+nL)/(Q2 nL) (61)
In a further aspect, the SNR in the voltage V at the measurement port 508 may be considered. As with the fractional change, the SNR may determine the sensitivity of the multi-purpose detection circuit 100. It may be distinguished between the intrinsic SNR (the sense signal-to-circuit intrinsic noise ratio) and the extrinsic SNR (the sense signal-to-circuit extrinsic noise ratio). With reference to
It may be further distinguished between a narrowband SNR resulting at the nominal sense frequency in the bandwidth of the voltage measurement circuit 510 and a broadband SNR defined in a larger bandwidth e.g., also covering the WPT operating frequency. The former mainly relates to the sensitivity of a multi-purpose detection circuit 100, while the latter may determine the dynamic range and filtering requirements of the voltage measurement circuit 510.
In another aspect, it may be meaningful to define the narrowband SNR at the measurement port 508 of the circuit 500 of
In yet a further aspect, it may be meaningful to define the broadband extrinsic SNR at the measurement port 508 of the circuit 500 of
SNRW=|V|/VW (62)
where |V| denotes the magnitude of the sense signal voltage and VW the disturbance voltage at the fundamental WPT operating frequency, which may be a prominent component in Vn when WPT is active. More specifically, the voltage |V| may refer to the r.m.s. voltage of the sense signal and VW to the r.m.s. disturbance voltage as measured at the measurement port 508 at the fundamental WPT operating frequency.
Using Equation (19), the differential narrowband extrinsic SNR for the series resonant configuration of the circuit 500 of
ΔSNRex,s≈|ΔZr||IL|/Vsn≈|ΔZ′r||VL|/Vsn=|ΔZ′r|ωs L|IL|/Vsn (63)
with |IL| denoting the magnitude of the sense signal current in the sense coil 502, which approximately equals the source current level I0, and Vsn the noise voltage as indicated in
Since the sense circuit 501 transforms the voltage drop across ΔZr to ΔV in the same way as it transforms Vsn to Vn, Equation (21) may also apply to the parallel resonant configuration of the circuit 500, meaning that:
ΔSNRex,p=(|Il|/Vsn)ωs L|ΔZ′r| (64)
Equation (63) and (64) show that the differential narrowband extrinsic SNR for the circuit 500 of
In some implementations (e.g., where the sense signal is numerically generated and converted to an analog signal using a digital-to-analog converter (DAC) e.g., in the signal generator circuit 406 with reference to
Vn≈R I0,n (65)
while the voltage change |ΔV| in presence of an object (e.g., object 110) is:
|ΔV|≈|Il||ΔZr|≈|I0||ΔZr| (66)
Applying Equations (35), (65), and (66) to Equation (10), the differential narrowband intrinsic SNR with respect to the noise current I0,n for the series resonant configuration of the circuit 500 of
ΔSNRint,s≈(|I0|/I0,n)|ΔZr|/R (67)
Using Equation (33), Equation (67) may also be written in terms of the Q-factor Qs and the normalized reflected impedance ΔZ′r as:
ΔSNRint,s≈(|I0|/I0,n)Qs|ΔZ′r| (68)
Using:
ΔY′<<1 (69)
which follows from Equations (23) and (25) and Equation (38), the magnitude |ΔV| of the voltage change in the parallel resonant configuration of the circuit 500 of
|ΔV|=|(I0/Y11)−(I0/Y11,0)|=|I0||(Y11,0+ΔY)−1−Y11,0)−1|≈|I0||ΔY|/|Y11,0|2 (70)
With the noise current I0,n as the predominant contribution, the noise voltage Vn at the parallel resonant frequency becomes:
V
n
=I
0,n
/|Y
11,0| (71)
Applying Equations (70), (71), and (50) to Equation (10), the differential narrowband intrinsic SNR with respect to the noise current I0,n for the parallel resonant configuration of the circuit 500 of
ΔSNRint,p≈(|I0|/I0,n)|ΔY′|≈(|I0|/I0,n)|ΔZr|/(RLp+R) (72)
Using Equation (53), Equation (72) may also be written in terms of the Q-factor Qp and the normalized reflected impedance ΔZ′r as:
ΔSNRint,p≈(|I0|/I0,n)|ΔZ′r|Qp nL/(1+nL) (73)
Similar considerations may be made for the thermal noise though likely less significant in practical implementations as further shown below with reference to TABLE 2. As previously mentioned, a thermal noise voltage is generated by the series equivalent loss resistances RLp and R. The noise voltage component Vn at the series resonant frequency may be considered as the thermal noise voltage generated by the series resonant resistance Rs as defined by Equation (30) and becomes approximately:
V
n=(4 k T Bm Rs)1/2≈(4 k T Bm R)1/2 (74)
where k denotes the Boltzmann constant, T the absolute temperature of the sense coil 502, and Bm the equivalent noise bandwidth of the voltage measurement circuit 510. Applying Equation (66) and (74) to Equation (10) provides for the differential narrowband intrinsic SNR with respect to thermal noise for the series resonant configuration of the circuit 500 of
ΔSNRint,s≈|IL||ΔZr|/Vn≈|IL|ωs L|ΔZ′r|/(4 k T Bm R)1/2 (75)
Accordingly, the thermal noise voltage Vn as resulting at parallel resonance may be considered as the thermal noise generated by the parallel resonant conductance Gp as defined by Equation (42). Assuming equal temperature T for the sense coil 502 and the parallel inductor 506, the noise voltage Vn becomes approximately:
Vn≈(4 k T Bm/Gp)1/2 (76)
Using Equation (70), (50), (42), (51), and the relation:
|I0|=|V|Gp≈|IL|ωp Lp Gp (77)
the voltage change ΔV may be expressed as:
|ΔV|≈|I0||ΔY|/Gp2≈|IL|ωp Lp|ΔY′|≈|IL|ωp Lp|ΔZr|/(R+RLp) (78)
Applying Equations (76) and (78) to Equation (10) also using Equation (42), provides for the differential narrowband intrinsic SNR with respect to the thermal noise for the parallel resonant configuration of the circuit 500 of
ΔSNRint,p≈|IL|ωp L|ΔZ′r|/(4 k T Bm (R+RLp))1/2 (79)
In a further aspect, the broadband extrinsic SNR as defined by Equation (62) with respect to the induced voltage component VsW at the fundamental WPT operating angular frequency ωW is considered. Assuming the magnetic field coupling as the predominant contribution, the disturbance signal voltage Vsn may relate to the WPT coil current IWPT as follows:
Vsn≈VsW≈ωW LsW IWPT (80)
where LsW denotes the mutual inductance between the sense coil 502 and the WPT coil (e.g., WPT coil 202 with reference to
1/(ωW Cs)<<ωW L (81)
ωs>>ωW (82)
and using Equation (29) and (47), the disturbance voltage component VW in the voltage V for the series resonant configuration of the circuit 500 of
V
n
=V
W
≈V
W ωW Cs ωW Lp≈VsW (ωW/ωs)2/nL (83)
The factor (ωW/ωs)2/nL may be considered as the attenuation of the low frequency induced voltage VsW by the high pass filter effect of the sense circuit 501. Using:
|V|=|I0|Rs≈|IL|R (84)
and applying Equations (33), (83), (80), (84), and (47) to Equation (62), the broadband extrinsic SNR for the series resonant configuration of the circuit 500 of
SNRW,s≈(|IL|/VsW)Ψs L(ωs/ωW)2 nL/Qs (85)
Using Equation (41), (81), and (82), the disturbance voltage component VW in the voltage V for the parallel resonant configuration of the circuit 500 of
V
n
=V
W
≈V
sW ωW Cs ωW Lp≈VsW(ωW/ωp)2/(1+nL) (86)
The factor (ωW/ωp)2/(1+nL) may be considered as the attenuation of the low frequency induced voltage VsW by the high pass filter effect of the sense circuit 501. Further, expressing the sense signal voltage |V| at the angular frequency ωp in terms of the sense coil current |IL| using Equation (40):
|V|≈|IL|((ωp Cs)−1−ωp L)≈|IL|ωp Lp (87)
and applying Equations (86) and (87) to Equation (62), the broadband extrinsic SNR with respect to the WPT fundamental disturbance voltage component VsW for the parallel resonant configuration of the circuit 500 of
SNRW,p≈(|IL|/VsW)ωp L(ωp/ωW)2(1+nL)/nL (88)
In yet another aspect, the temperature sensitivity as defined by Equations (11) and (12) for the real and imaginary part of Z11, respectively, is considered. Using Equation (35), the real part temperature sensitivity of the circuit 500 of
S
ϑ,R
=Re{ΔZ′
ϑ
}/Re{ΔZ′}≈Re{ΔZ′
ε}/(Qs Re{ΔZ′r}) (89)
Equation (89) shows that the real part temperature sensitivity reduces as the Q-factor Qs of the sense circuit 501 increases. However, the imaginary part temperature sensitivity may not improve and may only reduce by lowering a temperature coefficient associated with the inductive and capacitive elements of the sense circuit 501.
In some implementations, components and materials with a low temperature coefficient (e.g., NP0-type capacitors) are used. In other implementations, temperature sensitivity is reduced e.g., using a combination of components or materials with a positive temperature coefficient and components or materials with a negative temperature coefficient in a manner such that the overall thermal drift is canceled out.
Equations (8) to (89) may also apply to the circuit 520 of
To analyze the circuit 520 with respect to the series resonant configuration of the circuit 520 of
Lσ<<L (90)
ω Lm>>RLm (91)
n
T
2
ω L
m
=α|Z
11,0| (92)
α>>1 (93)
The ratio nT:1 refers to the transformation ratio of the ideal transformer as used in the transformer's 526 equivalent circuit model with reference to
Qw≈ω Lm/Rw (94)
and using Equations (30) and (34), the impedance magnitude |Z11,0| at the measurement port 528 for the series resonant configuration of the circuit 520 of
|Z11,0|≈Rs≈nT2(R+Rw)≈nT2((ωs L/Q)+(ωs Lm/Qw))≈nT2 ωs Lm/α (95)
yielding for the inductance ratio nL for satisfying Equation (92):
n
L
=L/L
m≈(Q/α)−(Q/Qw)>0 (96)
Equation (95) may also be written as:
|Z11,0|≈Rs≈nT2(1+Q/(nL Qw))R≈nT2(Qw/(Qw−α))R (97)
For Qw>>α and nT>1, the series resonant resistance Rs may be nT2 R.
Defining the Q-factor of the series resonant configuration of the sense circuit 521 as:
Qs≈nT2 ωs L/Rs≈ωs L/(R+Rw) (98)
and substituting Rs by Equation (97) yields for the Q-factor of the series resonant configuration of the sense circuit 521 of
Qs≈Q(1−α/Qw) (99)
and for the fractional change using Equation (35):
ΔZ′≈Qs ΔZ′r≈Q(1−α/Qw)ΔZ′r (100)
The factor 1−α/Qw and thus Qs degrades as Qw decreases or α increases. This factor may also apply to the SNRs that are related to Qs as given above e.g., by Equations (68) and (85). In an example implementation configured with α=10 and Qw=30, this factor may be 2/3. It may be appreciated that the Q-factor Qw relates to the component volume rather than to the transformation ratio nT:1.
In some implementations, the transformer impact factor a represents a trade-off between an error in the measured impedance change ΔZ (e.g., with respect to the angle arg{ΔZr} as previously discussed with reference to
To analyze the circuit 520 with respect to the parallel resonant configuration of the circuit 520 of
ω Lm>>RLm (101)
Defining the inductance ratio:
nL≈L/Lm (102)
and the Q-factor of the transformer 526 with respect to RLm (e.g., core losses) as:
QLm≈ω Lm/RLm (103)
the admittance |Y11,0| at the measurement port 528 for the parallel resonant configuration of the circuit 520 of
|Y11,0|≈Gp≈(nL/nT2)((Q/Qw)+(Q/QLm)+nL)/(Q ωp L) (104)
with Gp denoting the parallel resonant conductance of the sense circuit 521. For Q≈Qw≈QLm and nL>1, the parallel resonant conductance becomes:
Gp≈(nL/nT2)(nL+2)/(Q ωp L) (105)
Applying Equation (104) to Equation (51), the Q-factor for the parallel resonant configuration of the sense circuit 521 may be expressed as:
Qp≈ωp(L+Lm)/(R+RW+RLm)≈Q(1+nL)/(nL+(Q/Qw)+(Q/QLm)) (106)
and the fractional change using Equation (53):
ΔY′≈Qp ΔZ′r nL/(1+nL)≈Q ΔZ′r nL/(nL+(Q/Qw)+(Q/QLm)) (107)
As nL increases, the factor Qp nL/(1+nL) approaches the Q-factor Q of the sense coil 502. In an example implementation configured with Lm=L (nL=1) and Qw=QLm=Q, this factor may be Q/3. For nL>>1, the fractional change ΔY′ may equal ΔZ′ of the series resonant configuration of the circuit 520 of
Based on Equation (107), an example implementation of the circuit 520 configured for parallel resonance with Lm=L (nLm=1) and Qw=QLm=Q and a transformer 526 with a transformation ratio nT2≈1/3 provides a fractional change ΔY′≈0.33 Q ΔZ′r. Another example implementation of the circuit 520 using a transformer 526 with nT=1 and Qw=QLm=Q and an inductance ratio nL≈2.16 to provide the same admittance |Y11|, yields a fractional change ΔY′≈0.52 Q ΔZ′r. A further example implementation of the circuit 500 (without transformer 526) with QLp=Q with an inductance ratio nL=1 and yields a fractional change ΔY′≈0.72 Q ΔZ′r.
From above examples, it may be concluded that using the transformerless circuit 500 may be preferable in a parallel resonant configuration. If a transformer (e.g., transformer 526) is indispensable e.g., for purposes of balancing as previously discussed with reference to
The equivalent circuit model 540-1 as illustrated in
With the assumption of an identical sense coil 502 in the circuits 500 and 540, the following relations may apply:
ΔY′r=ΔZ′r (108)
G≈R/(ω L)2 (109)
ΔYr≈ΔZr/(ω L)2 (110)
Isn≈Vsn/(ω L) (111)
with ΔY′r, ΔZ′r, ΔZr, R, and Vsn referring to the normalized reflected admittance, the normalized reflected impedance, the reflected impedance of the object 110 in the sense coil 502, the equivalent series resistance of the sense coil 502, and the disturbance voltage Vsn with reference to the circuit 500 of
To analyze the series and parallel resonant configuration of the circuit 540 of
1/ω L>>G (112)
|ΔYr|<<G (113)
are made for a frequency range about the resonant frequency.
In an implementation configured for parallel resonance and with a susceptance:
ω Cs>>|Y11| (114)
in a frequency range about the resonant frequency, the admittance Y11 at the measurement port 548 of the circuit 540 of
Y11≈G+(jω L)−1+jω Cp+ΔYr (115)
In absence of a foreign object, a local minimum of |Y11,0(ω)| (parallel resonance) occurs substantially at an angular frequency ω satisfying:
(jω L)−1+jω Cp≈0 (116)
yielding the parallel resonant angular frequency:
ωp≈(L Cp)−1/2 (117)
At this frequency, the admittance Y11,0 becomes approximately real:
Y
11,0
≈Re{Y
11,0
}=G
p
≈G (118)
with Gp denoting the parallel resonant conductance, while the admittance Y11 in presence of an object (e.g., object 110) is approximately:
Y11≈Gp+ΔY≈G+ΔYr (119)
with ΔYr referring to the reflected admittance as previously defined with reference to
Applying Equations (118) and (119) to Equation (9), the fractional change ΔY′ for the parallel resonant configuration of the circuit 540 of
ΔY′≈ΔY/Gp≈ΔYr/G (120)
Defining the normalized reflected admittance:
ΔY′r=ΔYr ωp L (121)
the Q-factor of the sense coil 502:
Q=1/(ωp L G) (122)
and the Q-factor of the parallel resonant configuration of the sense circuit 541 of
Qp≈1/(ωp L Gp)≈Q (123)
the fractional change may also be written in terms of ΔY′r and Qp:
ΔY′≈Qp ΔY′r (124)
To analyze the series resonant configuration of the circuit 540 of
|ω Cp−(ω L)−1|>>G (125)
is made for a frequency range about the resonant frequency. The impedance Z11 at the measurement port 548 in presence of an object (e.g., object 110) may be expressed as:
Z
11=(jω Cs)−1+(G+jω Cp+(jω L)−1+ΔYr)−1 (126)
Using Equations (112), (113), (125), and (38), Equation (126) may be approximated as:
Z11≈(jω Cs)−1+(jω Cp+(jω L)−1)−1+(G+ΔYr)(ω Cp+(ω L)−1)−2 (127)
In absence of a foreign object, a local minimum of |Z11,0(ω)| (series resonance) occurs substantially at an angular frequency ω satisfying:
(jω L)−1+jω(Cp+Cs)≈0 (128)
yielding for the series resonant angular frequency:
ωs≈(L(Cp+Cs))−1/2 (129)
At this frequency, the impedance Z11,0 becomes substantially real:
Z11,0≈Re{Z11,0}≈Rs≈G/(ωs Cs)2 (130)
with Rs denoting the series resonant resistance, while the impedance Z11 in presence of an object (e.g., object 110) is approximately:
Z11≈Rs+ΔZ≈(G+ΔYr)/(ωs Cs)2 (131)
with:
ΔZ≈ΔYr/(ωs Cs)2 (132)
Further, defining the Q-factor of the sense coil 502:
Q=1/(ωs L G) (133)
and the capacitance ratio:
n
C
=C
p
/C
s (134)
the impedance Z11,0 of Equation (130) at ωs may be expressed as:
Z
11,0
=R
s≈1/(Q ωs L ωs2 Cs2)≈(1+nC)2 ωs L/Q (135)
For nC>>1, the series resonant resistance Rs becomes approximately:
Rs≈nC2 ωs L/Q (136)
and approximately 9 R for the case nC=2. According to Equation (135), the impedance Z11 at ωs of the sense circuit 541 can be modified (e.g., increased) by adjusting the capacitance ratio nC=Cp/Cs accordingly, while maintaining series resonance substantially at the nominal sense frequency. Therefore, in some implementations, the series resonant configuration of the circuit 540 of
Applying Equations (130) and (131) to Equation (8), the fractional change ΔZ′ for the series resonant configuration of the circuit 540 of
ΔZ′=ΔZ/Rs≈ΔYr/G (137)
showing that the impedance change ΔZ is substantially proportional to the reflected admittance ΔYr. Therefore, the angle arg{ΔZ} of the measured impedance change ΔZ may be indicative of the angle arg{ΔYr}. As previously described with reference to the circuit 540 of
Defining the Q-factor of the series resonant configuration of the sense circuit 541 of
Qs≈ωs(Cs+Cp)/G≈Q (138)
which approximately equals the Q-factor Q of the sense coil 502 as defined by Equation (133), using Equation (7), and applying Equation (133) to (137), the fractional impedance change ΔZ′ may also be written as:
ΔZ′≈Qs ΔY′r≈Q ΔY′r (139)
In a further aspect, the drive voltage level V0, the resulting current I at the measurement port 548, and the drive power level P are considered. In some implementations based on the circuit 540 of
V0≈ωp L|IL| (140)
Using Equations (118) and (122), the current I at the measurement port 548 may be expressed approximately as:
I≈|Y
11,0
|V
0
≈|I
L|ωp L G=|IL|/Q (141)
and the drive power level:
P≈V
0
I=|I
L|2 ωp L/Q (142)
For the series resonant configuration of the circuit 540 of
|VL|≈ωs L|IL| (143)
and the current I using Equations (129), (134), and (143):
I≈|V
L|ωs Cs=ωs2 L Cs|IL|=|IL|Cs/(Cs+Cp)=|IL|/(1+nC) (144)
respectively. The voltage V0 and the drive power P required to drive the sense coil 502 with a current level |IL| can be found to be:
V
0
≈I R
s
≈I(1+nC)R=|IL|ωs L/Qs (145)
P≈I2 Rs≈|IL|2 ωs L/(Qs(1+nC)) (146)
In a further aspect, it may be meaningful to define the narrowband SNR at the measurement port 548 of the circuit 540 of
In another aspect, it may be meaningful to define the broadband extrinsic SNR at the measurement port 548 of the circuit 540 of
SNRW=|I|/IW (147)
where |I| denotes the magnitude of the sense signal current and IW the disturbance current at the fundamental WPT operating frequency, which may be a prominent component in In when WPT is active. More specifically, the current |I| may refer to the r.m.s. current and IW to the r.m.s. disturbance current as measured at the measurement port 548 at the fundamental WPT operating frequency.
Using Equation (14) and (7), the differential narrowband extrinsic SNR of the parallel resonant configuration of the circuit 540 of
ΔSNRex,p≈|ΔYr|ωp L|IL|/Isn=|ΔY′r||IL|/Isn (148)
with Isn the noise current as illustrated in
Since the sense circuit 541 transforms the shunt current through ΔYr to ΔI in the same way as it transforms Isn to In, Equation (148) also applies to the series resonant configuration, meaning that:
ΔSNRex,s≈ΔSNRex,p (149)
In operations of the circuit 540 where the noise voltage V0,n is predominant as previously discussed, the noise current In in the parallel resonant configuration of the circuit 540 is approximately:
In≈Gp V0,n (150)
and the current change in presence of an object (e.g., object 110) is:
|ΔI|≈|V0||ΔYr| (151)
Applying Equations (123), (124), (150), and (151) to Equation (14), the differential narrowband intrinsic SNR with respect to the noise voltage V0,n for the parallel resonant configuration of the circuit 540 of
SNRint,p≈(|V0|/V0,n)|ΔYr|/Gp≈(|V0|/V0,n)|ΔY′| (152)
Using Equation (124), Equation (152) may also be written in terms of the Q-factor Qp and the normalized reflected admittance ΔY′r as:
ΔSNRint,p≈(|V0|/V0,n)Qp|ΔY′r′ (153)
Using:
ΔZ′<<1 (154)
which follows from assumptions of Equations (112) and (113) and Equation (38), the magnitude |ΔI| of the current change in the series resonant configuration of the circuit 540 of
|ΔI|=|(V0/Z11)−(V0/Z11,0)|=|V0||(Z11,0+ΔZ)−1−Z11,0)−1|≈|V0||ΔZ|/|Z11,0|2 (155)
With the noise voltage V0,n as the predominant noise contribution, the noise current In at the series resonant frequency becomes:
I
n
=V
0,n
/|Z
11,0| (156)
Applying Equations (155), (156), and (137) to Equation (14), the differential narrowband intrinsic SNR with respect to the noise voltage V0,n for the series resonant configuration of the circuit 540 of
ΔSNRint,s≈(|V0|/V0,n)|ΔZ′|≈(|V0|/V0,n)|ΔYr|/G (157)
Using Equation (139), Equation (157) may also be written in terms of the Q-factor Qs and the normalized reflected admittance ΔY′r as:
ΔSNRint,s≈(|V0|/V0,n)Qs|ΔY′r| (158)
Similar considerations may be made for thermal noise, though likely less significant in practical implementations as shown below with reference to TABLE 2. As previously mentioned, a thermal noise current is generated by the equivalent parallel conductance G of the sense coil 502. The noise current In may be considered as the thermally generated by the parallel resonant conductance Gp as defined by Equation (118) and becomes approximately:
I
n=(4 k T Bm Gp)1/2≈(4 k T Bm G)1/2 (159)
Applying Equation (151), (159), and (7) to Equation (14), the differential narrowband intrinsic SNR with respect to thermal noise of the parallel resonant configuration of the circuit 540 may be expressed as:
ΔSNRint,p≈|IL|ωp L|ΔYr|/In≈|IL||ΔY′r|/(4 k T Bm G)1/2 (160)
where k denotes the Boltzmann constant, T the absolute temperature, and Bm the equivalent noise bandwidth of the current measurement circuit 550.
In the series resonant configuration of the circuit 540, the noise current component In as thermally generated by the series resonant resistance Rs as defined by Equation (118) becomes:
I
n=(4 k T Bm/Rs)1/2 (161)
Using Equations (137), (129), and the relation:
|V0|=|I|Rs≈|IL|ωs2 L Cs Rs (162)
the current change |ΔI| may be expressed as:
|ΔI|≈|V0||ΔZ|/Rs2≈|IL|ωs2 L Cs|ΔZ′|≈|IL||ΔYr|/((1+nC)G) (163)
Applying Equation (161) and (163) to Equation (14), the differential narrowband intrinsic SNR with respect to thermal noise of the series resonant configuration of the circuit 540 may be expressed in terms of the sense coil 502 current |IL| and the normalized reflected admittance |ΔY′r| as:
ΔSNRint,s≈|IL||ΔY′r|/(4 k T Bm G)1/2 (164)
In yet a further aspect, the broadband extrinsic SNR as defined by Equation (147) with respect to the induced current component IsW at the fundamental WPT operating angular frequency ωW is considered. Assuming:
Isn=IsW (165)
1/(ωW Cp)>>ωW L (166)
and using Equation (117), the disturbance current component IW in the current I for the parallel resonant configuration of the circuit 540 of
I
n
=I
W
≈I
sW ωW L ωW Cs≈IsW(ωW/ωp)2/nC (167)
The factor (ωW/ωp)2/nC may be considered as the attenuation of the low frequency induced current IsW by the high pass filter effect of the sense circuit 541. Using:
|I|≈|VL|G (168)
and applying Equations (167), (168), and (134) to Equation (147), the broadband extrinsic SNR of the parallel resonant configuration of the circuit 540 of
SNRW,p≈(|VL|G/IsW)(ωp/ωW)2 nC (169)
with VL denoting the voltage across the sense electrode 702.
Using the relation:
|VL|≈|IL|ωp L (170)
and Equation (123), Equation (169) may also be written as:
SNRW,p≈(|IL|/IsW)(ωp/ωW)2 nC/Qp (171)
Using Equations (82), (129), (134), and (166), the disturbance current IW in the current I for the series resonant configuration of the circuit 540 of
IW≈IsW ωW L ωW Cs≈IsW(ωW/ωs)2(1+nC) (172)
The factor (ωW/ωp)2 (1+nC) may be considered as the attenuation of the low frequency induced current IsW by the high pass filter effect of the sense circuit 541. Further, expressing the sense signal current |I| at the angular frequency ωs in terms of the sense coil 502 voltage |VL| using Equations (138) and (168):
|I|≈|VL|(ωs Cs−(ωs L)−1)≈|VL|ωs Cs (173)
and applying Equations (172) (173), and (134) to Equation (14), the broadband extrinsic SNR with respect to the WPT fundamental disturbance current component IsW for the series resonant configuration of the circuit 540 of
SNRW,s≈(|VL|ωs Cs/IsW)(ωs/ωW)2(1+nC) (174)
Using the relation:
|VL|≈|IL|ωs L (175)
Equations (129), and (165), Equation (174) may also be written as:
SNRW,s≈(|IL|/IsW)(ωs/ωW)2 (176)
Based on Equations (171) and (176) and ωs=ωp, the following relation between the broadband extrinsic SNRs of the parallel and series resonant configurations of the circuit 540 of
SNRW,s≈SNRW,p Qp/nC (177)
TABLE 1 provides example parameter values as used for a numerical analysis of the series and parallel resonant configuration of the circuit 500 of
Numerical results as obtained from a circuit analysis using the numerical assumptions of TABLE 1 are listed in TABLE 2. While the SNR values are obtained using the corresponding approximate equations as defined above with reference to
ε≈arg{ΔZ exp(−j arg{Z11,0})}−arg{ΔZ′r} (178)
where ΔZ exp(−j arg{Z11,0}) denotes the impedance change with the angle correction applied. For the parallel resonant configuration of the circuit 500 of
ε≈arg{ΔY exp(−j arg{Y11,0})}−arg{ΔZ′r} (179)
For the parallel resonant configuration of the circuit 540 of
ε≈arg{ΔY exp(−j arg{Y11,0})}−arg{ΔY′r} (180)
and for the series resonant configuration of the circuit 540 of
ε≈arg{ΔZ exp(−j arg{Z11,0})}−arg{ΔY′r} (181)
Further, TABLE 2 includes the drive current level I0, the drive power level P required to drive the sense coil 502 of the sense circuit 501 with the sense current |IL| as specified in TABLE 1. Accordingly, it includes the drive voltage level V0, the drive power level P required to drive the sense coil 502 of the sense circuit 541 with the sense current |IL| as specified in TABLE 1.
Based on the numerical results of TABLE 2, the following conclusions may be drawn. The high impedance magnitude |Z11,0| as generally presented by the parallel resonant configuration of the circuit 500 of
The complex plane 600 and the shaded areas (e.g., angle ranges 602 to 610) may also apply to the reflected admittance ΔYr by simply relabeling the real and imaginary axis by Re{ΔYr} and j Im{ΔYr}, respectively (not shown in
Further,
The angle range 602 (e.g., close to −90°) in quadrant 4 may be characteristic for an object (e.g., object 110a) exhibiting a relatively high electric conductivity (e.g., σ>50 MS/m) and that is substantially non ferromagnetic (relative permeability μr≈1) at the sense frequency. For a sense frequency in the MHz range, the impedance ΔZr reflected by a copper coated coin may be found in the angle range 602.
The angle range 604 (e.g., around −80°) in quadrant 4 may be characteristic for an object (e.g., 110) exhibiting a substantially lower equivalent conductivity (e.g., σ>5 MS/m) and substantially no ferromagnetic effect (relative permeability μr≈1) at the sense frequency. A piece of thin foil or metallized paper (e.g., aluminum coated paper) as illustrated in
The angle range 606 (around 0°) in quadrant 4 and 1 may be characteristic for an object (e.g., object 110c) exhibiting a relatively high conductivity (e.g., σ>10 MS/m) and a substantial ferromagnetic effect (e.g., μr>50) at the sense frequency. An object made of ferromagnetic steel may reflect an impedance ΔZr in the angle range 606 for a sense frequency in the MHz range. Ferromagnetism (μr>1) in the metallic object 110c generally reflects an impedance ΔZr with an imaginary part Im{ΔZr}>0. On the other hand, the electrical conductivity of the metallic object 110d generally reflects an impedance ΔZr with Im{ΔZr}<0 and Re{ΔZr}>0. Superimposing the two opposing effects may result in a net reflected impedance ΔZr e.g., in the angle range 606.
The angle range 608 (e.g., around 45°) in quadrant 1 may be characteristic for an object (e.g., object 110d) exhibiting a relatively high conductivity (e.g., σ>10 MS/m) and a substantial ferromagnetic effect (e.g., μr>50) at the sense frequency and with a length substantially larger than a thickness. An object made of ferromagnetic steel and with such geometry may reflect an impedance ΔZr in the angle range 606 for a sense frequency in the MHz range. Ferromagnetism (μr>1) in the metallic object 110d generally reflects an impedance ΔZr with a positive imaginary part that prevails the conductivity effect acting in the opposite direction as described above with reference to the object 110c. Superimposing the two effects results is a net reflected impedance ΔZr with a positive imaginary part (Im{ΔZr}>0) substantially equal to the real part Re{ΔZr} corresponding to the angle range 608. A reflected impedance ΔZr in this angle range may also be observed from a paper clip made of ferromagnetic steel (not shown in
Finally, the angle range 610 (e.g., close to 90°) in the quadrant 1 may be characteristic for a substantially non-conductive object (e.g., object 112, object 114) that exhibits a dielectric effect (εr>1) at the sense frequency. A dielectric object (e.g., object 112) may reflect an impedance ΔZr in the angle range 610. A living object (e.g., object 114) may also reflect an impedance ΔZr in the angle range 610. As previously discussed in connection with
In an aspect of the multi-purpose detection circuit 100, objects 110 (e.g., object 110a, 110b, 110c, 110d) reflecting an impedance ΔZr in the respective angle ranges 602, 604, 606, and 608 or anywhere between these ranges may be subject of induction heating if exposed to the strong WPT magnetic field. This may be particularly true for thin foils (e.g., object 110b) and objects that are both substantially electrically conductive and ferromagnetic (e.g., objects 110c and 110d). Ferromagnetism in a metallic object (e.g., object 110c) may result in a pronounced skin effect displacing the induced eddy currents into a thin layer (skin) at the surface of the object. This may substantially reduce the effective electrical conductivity of the object causing substantially higher power dissipation if compared to a non-ferromagnetic metallic object. Further, lengthy ferromagnetic, metallic objects (e.g., objects 110d) that may reflect an impedance ΔZr in the angle range 608 tend to experience magnetic saturation resulting in excessive hysteresis losses and consequent heating. Therefore, this object category may be characterized by the highest loss power density (e.g., in Watt per unit surface area) and thus highest heating temperatures. Therefore, it may be desirable to selectively increase a sensitivity of a multi-purpose detection circuit 100 to objects (e.g., objects 110) of this category as disclosed in U.S. Pat. No. 10,495,773 titled Improving Foreign Object Detection for Ferromagnetic Wire-Like Objects, the entire contents of which are hereby incorporated by reference.
In another aspect of the multi-purpose detection circuit 100, the inductive sense circuit (e.g., inductive sense circuit 501 of
In a further aspect of the multi-purpose detection circuit 100 relying on a time-differential detection scheme as previously described with reference to
It may be appreciated that discriminating objects (e.g., object 112 from object 110a) reflecting an impedance ΔZrclose to the imaginary axis (angle ranges 610 and 602, respectively) based on the angle of complex outputs of a time-differential detector may require accurate measurement of the angle arg{ΔZr}. With the object 112, an error in the measured angle of a few degrees may cause the output of the time-differential detector to infringe into the angle range 602, when the object 112 is removed from the proximity of the sense coil 502 as explained above. Such event may pretend a metal object (e.g., object 110a) brought into the proximity of sense coil 502 and a cause for a false detection.
In some implementations of the multi-purpose detection circuit 100, a reliable discrimination between the object 112 and the object 110a based on the angle of complex outputs of a time-differential detector can hardly be achieved. This may cause an issue e.g., in presence of water accumulation on the surface of the housing of a wireless power transfer structure (e.g., housing 328 of the wireless power transfer structure 200 with reference to
Because the magnitude of the complex output ΔZ/Δt of the time-differential detector increases with the speed of movement (first time derivative) of the water object, the probability of a false positive detection of a metal object (e.g., metal object 110a) increases with the speed of the water flow.
In another aspect, experiments with drain water flows on the surface of a wireless power transfer structure have shown that the impedance ΔZras reflected by the water object depends on its shape but also on its ion concentration (e.g., salinity). More specifically, these experiments with water of any non-zero ion concentration (excluding pure distilled water) have shown the angle arg{ΔZr} changing with the shape of the water object. An elongated water bubble typically reflects an impedance ΔZrwith an angle arg{ΔZr} substantially smaller than 90° (with a substantial real part), while the angle produced by a more rotund bubble and an ion-concentration equal to or less than sea water can be found close to 90°. This water phenomena are described in more detail in connection with
In a further aspect to reduce the probability of false positive detections due to presence of drain water flow on the surface of a wireless power transfer structure, some implementations of the multi-purpose detection circuit 100 employ a water flow detector configured for detecting formation and flow of water puddles. Outputs of the water flow detector are used to discriminate a water object (e.g., object 112, water bubble) from a metal object (e.g., object 110a) and thus to prevent a false positive detection.
In some implementations, the water flow detector is based on a pattern recognition approach using a plurality of time series of complex outputs of a time-differential detector. Each time series is associated to a sense element of the plurality of sense elements 107a, 107b, . . . , 107n in the array 107. More specifically, water flow detection exploits the peculiar characteristics in the temporal and spatial patterns as produced by the drain water flow affecting a cluster of sense elements of the plurality of sense elements 107a, 107b, . . . , 107n in the array 107. In some implementations, a spatial pattern is created by mapping quasi-concurrent outputs of the time-differential detector onto a two-dimensional matrix consisting of a plurality of elements arranged in lines and columns. The position of each matrix element corresponds to the position of the associated sense element in the two-dimensional array 107. A temporal and spatial pattern refers to a three-dimensional matrix consisting of a plurality of elements arranged in lines, first columns, and second columns where the elements of a second column are consecutively received outputs of the time-differential detector associated to a sense coil (e.g., sense element 107a with reference to
In another implementation, the water flow detector is based on a pattern recognition approach using at least in part a plurality of time series of complex detector outputs indicative of the impedance Z11 with reference to
The descriptions of the circuits 700, 710, 720, 730, 740, and 750 of
In some implementations, the sense signal is a high frequency signal with a spectrum substantially in the MHz range (e.g., in a range from 2.5 MHz to 3.5 MHz). In other implementations, the sense signal is constraint to the range from 3.155 MHz to 3.400 MHz for frequency regulatory reasons as previously mentioned in connection with
The ground symbol shown in the schematic diagrams of
The circuit 700 of
The sense circuit 701 comprises a single-electrode capacitive sense element 702 (single-ended sense electrode 702) having a signal terminal 703, a capacitance C and an equivalent series resistance R, a series inductor 704 having an inductance Ls and an equivalent series resistance RLs electrically connected in series to the sense electrode 702 at the signal terminal 703, and a parallel inductor 706 having an inductance Lp and an equivalent series resistance RLp electrically connected to the series inductor 704 and in parallel to the measurement port 708.
It may be appreciated that electrical losses in the series inductor 704 and in the parallel inductor 706 are the most prominent losses in the capacitive sense circuit 701. These losses may prevail the electrical losses intrinsic to the sense electrode 702 and extraneous losses in its surrounding materials (e.g., the Litz wire of the WPT coil 202, the ferrite, and the plastic housing of the wireless power transfer structure 200 where the sense electrode 702 may be integrated). These materials may interact with the predominantly electric field as generated by the sense electrode 702 causing some losses that may be included in the equivalent series resistance R as indicated in
The sense electrode's 702 capacitance C may include various capacitances as indicated in
The sense electrode 702 may also include a self-inductance (not indicated in
The sense circuit 701 may be configured to provide a local minimum in the impedance magnitude |Z11,0(ω)| (series resonance) substantially at the nominal sense frequency (e.g., at 3 MHz), where Z11,0 refers to the impedance as presented by the sense circuit 701 at the measurement port 708 in absence of a foreign object with reference to
In an example series resonant configuration of the sense circuit 701, the reactance of the series inductor 704 substantially compensates for the reactance of the sense electrode 702 at the nominal sense frequency providing an impedance Z11,0 that is substantially real (resistive). In this configuration, the inductance Lp of the parallel inductor 706 may be similar or larger than the inductance Ls of the series inductor 704. In other terms, the impedance magnitude of the parallel inductor 706 may be substantially (e.g., 10 times) higher than the impedance magnitude |Z11,0| as presented at the nominal sense frequency. In this configuration, the parallel inductor 706 may exert a negligible impact on the impedance |Z11,0| at the nominal sense frequency.
In an example parallel resonant configuration of the sense circuit 701, the reactance of the series inductor 704 undercompensates for the reactance of the sense electrode 702 at the nominal sense frequency. The residual capacitive susceptance of the series connection of the inductor 704 and the sense electrode 702 is substantially compensated for by the susceptance of the parallel inductor 706 providing an admittance Y11,0 that is substantially real (resistive). In this configuration, the inductance Lp of the parallel inductor 706 may be smaller, similar, or larger than the inductance Ls of the series inductor 704. Stated in other terms, the admittance magnitude of the parallel inductor 706 may be substantially (e.g., 20 times) higher than the admittance magnitude |Y11,0| as presented at the nominal sense frequency. In this configuration, the parallel inductor 706 exerts a significant impact on the admittance Y11,0 at the nominal sense frequency.
In some implementations, the parallel inductor 706 together with the series inductor 704 are used for purposes of resonance tuning and impedance transformation, e.g., to transform the impedance Z11 to match the sense circuit 701 with an operating impedance range as previously mentioned with reference to
Impedance transformation may be particularly effective, if the sense circuit 701 is configured for parallel resonance. More specifically, increasing the inductance ratio Ls/Lp, while maintaining parallel resonance at the nominal sense frequency, may substantially increase the admittance |Y11,0| of the parallel resonant configuration at the nominal sense frequency.
Increasing the inductance ratio Ls/Lp, while maintaining resonance at the nominal sense frequency, may also somewhat decrease the impedance |Z11,0| as presented at the nominal sense frequency in the series resonant configuration of the sense circuit 701. However, impedance transformation may be limited and far less effective than that of the parallel resonant configuration.
In another aspect of resonance tuning, at least one of the series inductor 704 and the parallel inductor 706 include a variable inductor as previously discussed with reference to
In yet another aspect, the sense electrode's 702 capacitance C in combination with the parallel inductor 706 form a 2nd order high pass filter to attenuate a low frequency disturbance component in the voltage V for purposes as previously discussed with reference to
With reference to
In some implementations, the current source 512 may be characterized by a quasi-ideal current source and the voltage measurement circuit 510 by a quasi-ideal voltage measurement circuit as previously defined with reference to
Though not shown herein, other impedance measurement techniques (e.g., the voltage source current measurement technique) may also be contemplated as previously discussed with reference to the circuit 500 of
Further, in some implementations, measurement of the voltage V and thus of the impedance Z11 may be affected by noise and other disturbance signals reducing a detection sensitivity of the multi-purpose detection circuit 100. The noise may include circuit intrinsic noise as generated in active and passive components of the circuit 700 of
Moreover, in implementations employing a selective voltage measurement circuit 510 as discussed above, the sense signal waveform as generated by the current source 512 and the corresponding filter of the voltage measurement circuit 510 are adapted e.g., to improve the SNR and consequently to improve the detection sensitivity as previously discussed with reference to
With reference to
In an implementation of the circuit 700 based on measuring the admittance Y11, presence of the object 110, 112, 114, or vehicle 330 may cause a change ΔY with respect to the admittance Y11,0 as measured in absence of a foreign object. Analogously, presence of an object (e.g., object 110) may be determined if ΔY satisfies certain criteria (e.g., the magnitude of ΔY exceeds a detection threshold).
As previously discussed with reference to the circuit 500 of
With reference to Equation (8) and (9), the fractional change ΔZ′ (or ΔY′) caused by a defined test object (e.g., object 112) placed at a defined position relative to the sense electrode 702 may relate to the detection sensitivity of an object detection circuit (e.g., the multi-purpose detection circuit 100 of
As further analyzed and discussed below with reference to
As previously discussed with reference to the circuit 500 of
In some implementations and configurations of the circuit 700 of
In an aspect of reducing an error in the measurement of the angle arg{ΔZ}, some implementations of a multipurpose detection circuit 100 employ a phase calibration of the analog circuitry (e.g., the analog front end portion of the measurement circuit 104 with reference to
Reactance compensation (resonance tuning) in the sense circuit 701 produces a local extremum (minimum or maximum) in the impedance magnitude function |Z11,0(ω)| and hence in the voltage magnitude |V| across the measurement port 708. Therefore, reactance compensation provides a mean to calibrate the voltage measurement circuit 510 and hence the impedance measurement with respect to the angle arg{ΔZ}.
In a first step of an example calibration procedure applicable to the series resonant configuration of the circuit 700 of
Applying the angle correction of Equation (13), an object (e.g., object 114) reflecting an impedance ΔZr that is imaginary (reactive) may cause a measured voltage change ΔVcal that is substantially imaginary. Nevertheless, a small residual error may remain in the angle arg{ΔVcal} due to the impact of the parallel inductor 706 and the electrical losses in the sense circuit 701. The residual angle error of an example series resonant configuration of the circuit 700 and for an example object 114 is provided in TABLE 4.
In some implementations, the residual error described above is reduced by configuring the parallel inductor 706 with an inductance Lp whose impedance ZLp is substantially larger (e.g., 10 times larger) than the series resonant resistance of the sense circuit 701. In other implementations, the residual error is reduced by measuring the impedance Z11,0 at two or more substantially different frequencies and by determining the elements of an equivalent circuit model of the sense circuit 701 (e.g., the equivalent circuit model illustrated in
In an implementation of the multipurpose detection circuit 100 using a plurality of capacitive sense circuits (e.g., capacitive sense circuits 108a, 108b, . . . , 108n), each including a respective capacitive sense element (e.g., capacitive sense element 109a, 109b, . . . , 109n), a residual error may be caused by a parasitic resonance effect of sense circuits associated to adjacent capacitive sense elements of an arrangement of sense electrodes. More precisely, a residual error in a first sense circuit (e.g., capacitive sense circuit 108a) including a first capacitive sense element (e.g., capacitive sense element 109a) may be caused by a parasitic resonance effect of at least one second capacitive sense circuit (e.g., capacitive sense circuit 108b) including a second capacitive sense element (e.g., capacitive sense element 109b) that is located adjacent to the first capacitive sense element.
Therefore, in some implementations of the multipurpose detection circuit 100, the measurement accuracy of the angle arg{ΔZ} and thus of the angle arg{ΔZr} is increased by an optimized design of the sense electrode 702 and by introducing some spacing between adjacent sense electrodes 702 in an arrangement of sense electrodes.
In an implementation configured for parallel resonance as defined above, the circuit 700 may be configured to measure the admittance Y11 and corresponding changes ΔY of Y11 as caused by the object 110, 112, 114, or vehicle 330. In this case, the admittance change ΔY may be indicative of the reflected impedance ΔZr as previously introduced. As discussed above with reference to the series resonant configuration, the angle arg{ΔY} may be subjected to an error and therefore may require calibration to reduce an error in the measurement of the angle arg{ΔY} and thus of the angle arg{ΔZr}.
In an implementation configured for parallel resonance, the circuit 700 may be calibrated analogously to the series resonant configuration however using the local minimum of the admittance function |Y11,0(ω)| where susceptance compensation occurs.
In a first step of an example calibration procedure applicable to the parallel resonant configuration of the circuit 700 of
Applying the angle correction of Equation (13), an object (e.g., object 114) reflecting an impedance ΔZr that is imaginary (reactive) produces a measured voltage change ΔVcal that is substantially imaginary. A residual error may remain in the angle arg{ΔVcal} due to the transformation of ΔZr to ΔY in the lossy sense circuit (e.g., sense circuit 701). The residual angle error of an example parallel resonant configuration of the circuit 700 and for an example reflected impedance ΔZr is provided in TABLE 4.
In an example implementation, the residual error due to the transformation of ΔZr to ΔY is reduced by measuring the admittance Y11,0 at two or more substantially different frequencies, supposing absence of a foreign object, and by determining the elements of an equivalent circuit model (e.g., the equivalent circuit model of
The series and the parallel resonant configuration of the circuit 700 of
The circuit 710 of
As the sense circuit 701 of
Though not shown in
As with the sense circuit 701 of
In an example series resonant configuration of the sense circuit 711, the reactance of the series inductor 714 substantially compensates for the reactance of the sense electrode 702 in parallel to the capacitor 715 at the nominal sense frequency providing an impedance Z11,0 that is substantially real (resistive). In this configuration, the inductance Lp of the parallel inductor 706 may be similar or larger than the inductance Ls of the series inductor 714. In other terms, the impedance magnitude of the parallel inductor 716 may be substantially (e.g., 10 times) higher than the impedance magnitude |Z11,0| as presented at the nominal sense frequency. In this configuration, the parallel inductor 716 may exert a negligible impact on the impedance |Z11,0| at the nominal sense frequency.
In an example parallel resonant configuration of the sense circuit 711, the reactance of the series inductor 714 undercompensates for the reactance of the sense electrode 702 in parallel to the capacitor 715 at the nominal sense frequency. The residual capacitive susceptance of the series connection of the inductor 704 and the parallel connection of the sense electrode 702 and capacitor 715 is substantially compensated for by the susceptance of the parallel inductor 716 providing an admittance Y11,0 that is substantially real (resistive). In this configuration, the inductance Lp of the parallel inductor 706 may be smaller, similar, or larger than the inductance Ls of the series inductor 714. Stated in other terms, the admittance magnitude of the parallel inductor 716 may be substantially (e.g., 20 times) higher than the admittance magnitude |Y11,0| as presented at the nominal sense frequency. In this configuration, the parallel inductor 716 exerts a significant impact on the admittance Y11,0 at the nominal sense frequency.
In some implementations, the parallel inductor 716 together with the series inductor 714 and the parallel capacitor 715 are used for purposes of resonance tuning and impedance transformation, e.g., to transform the impedance Z11 to match the sense circuit 711 with an operating impedance range as previously mentioned with reference to
If configured for series resonance, the impedance magnitude |Z11,0| may be decreased mainly by decreasing the capacitance ratio C/Cp. If configured for parallel resonance, the admittance magnitude |Y11,0| may be increased mainly by increasing the inductance ratio Ls/Lp.
In another aspect of resonance tuning, the parallel capacitor 715 may include a variable capacitor whose capacitance Cs can be electronically controlled (e.g., a DC controlled capacitor) forming a variable capacitor 715. In some implementations of the circuit 700, a variable capacitor 715 is used to compensate for a temperature drift, an aging, or a detuning of the sense circuit 701 caused by an external impact and to maintain its resonance substantially at the nominal sense frequency. In a further aspect, the variable capacitor 714 in combination with a variable inductor 714 are used to vary the impedance |Z11,0| of the sense circuit 701.
In a further aspect, the sense electrode's 702 capacitance C in combination with the parallel inductor 716 form a 2nd order high pass filter to attenuate a low frequency disturbance component in the voltage V for purposes as previously discussed with reference to
With reference to
The circuit 720 of
As the sense circuit 701 of
Though not indicated in
The sense circuit 721 may be configured to provide a local minimum in the impedance magnitude function |Z11,0(ω)| (series resonance) substantially at the nominal sense frequency. Alternatively, it may be configured to provide a local minimum in the admittance magnitude function |Y11,0(ω)| (parallel resonance) substantially at the nominal sense frequency using the transformer's 726 secondary referred main inductance Lm in a manner similar to using the inductance Lp as described above with reference to
In an example series resonant configuration of the sense circuit 721, the reactance of the series inductor 724 together with the transformer's 726 secondary referred leakage inductance Lσ substantially compensates for the reactance of the sense electrode 702 at the nominal sense frequency providing an impedance Z11,0 at the measurement port 728 that is substantially real (resistive). In this configuration, the transformer's 726 secondary referred main inductance Lm may be similar or larger than the inductance Ls of the series inductor 724. Stated in other terms, the primary referred open-circuit impedance of the transformer 726 may be substantially (e.g., 10 times) higher than the impedance magnitude |Z11,0| as presented at the nominal sense frequency. Apart from the impedance transformation by the factor nT2, the transformer 726 may exert a negligible impact on the impedance |Z11,0| at the nominal sense frequency.
In an example parallel resonant configuration of the sense circuit 721, the reactance of the series inductor 724 together with the transformer's 726 secondary referred leakage inductance Lσ undercompensates for the reactance of the sense electrode 702 at the nominal sense frequency. The residual capacitive susceptance of the series connection of the inductor 724, the transformer's 726 leakage inductance Lσ, and the sense electrode 702 is substantially compensated for by the susceptance of the transformer's 726 secondary referred inductance Lm providing an admittance Y11,0 that is substantially real (resistive). In this configuration, the inductance Lm may be smaller, similar, or larger than the inductance Ls of the series inductor 724. Stated in other terms, the primary referred open-circuit admittance of the transformer 726 may be substantially (e.g., 20 times) higher than the admittance magnitude |Y11,0| as presented at the nominal sense frequency. In this configuration and apart from the admittance transformation, the transformer 726 exerts a significant impact on the admittance Y11,0 at the nominal sense frequency.
The transformer 726 may serve for various purposes. In some implementations, the transformer 726 is a nT:1 transformer with nT≠1 used at least for impedance transformation e.g., to match the impedance magnitude |Z11| of the sense circuit 721 with an operating impedance range as previously mentioned with reference to
Apart from the transformation ratio nT:1, the inductance ratio Ls/Lm may be an additional parameter to match the admittance magnitude |Y11,0| of the parallel resonant configuration with an operating admittance range of the multi-purpose detection circuit 100 with reference to
In a further aspect, the sense electrode's 702 capacitance C in combination with the transformer's 726 main inductance Lm form a 2nd order high pass filter to attenuate a low frequency disturbance component in the voltage V for purposes as previously discussed in connection with
With reference to
The circuit 730 of
As opposed to the sense circuits 701, 711, and 721, the sense circuit 731 is operated in a differential mode and uses a substantially symmetric double-electrode capacitive sense element (e.g., double-ended sense electrode 732) composed of electrodes 732a and 732b (also referred to herein as a double-ended sense electrode) providing a differential-mode capacitance C. The sense circuit 731 may be split into a first branch and a second branch with an equal topology. The sense circuit 731 may be substantially symmetric (balanced) with respect to its capacitances and inductances. Further, the sense circuit 731 comprises a differential-mode series inductor 734 having an inductance Ls/2 in each branch and that is electrically connected to the double-ended sense electrode 732. Moreover, the sense circuit 731 comprises a transformer 736 with a transformation ratio nT:1 and secondary referred main inductance Lm with reference to
The double-ended sense electrode 732 provides a differential-mode capacitance C that may include various capacitances as indicated in
Though not indicated in
The sense circuit 731 may be configured to provide a local minimum in the impedance magnitude function |Z11,0(ω)| (series resonance) substantially at the nominal sense frequency. Alternatively, it may be configured to provide a local minimum in the admittance magnitude function |Y11,0(ω)| (parallel resonance) substantially at the nominal sense frequency using the transformer's 736 secondary referred main inductance Lm as described above with reference to
In an example series resonant configuration of the sense circuit 731, the reactance of the differential-mode series inductor 734 together with the transformer's 736 secondary-referred leakage inductance Lσ substantially compensates for the reactance of the double-ended sense electrode 732 at the nominal sense frequency providing an impedance Z11,0 at the measurement port 738 that is substantially real (resistive). In this configuration, the transformer's 736 secondary referred main inductance Lm may be similar or larger than the inductance Ls of the differential-mode series inductor 734. Stated in other terms, the primary referred open-circuit impedance of the transformer 736 may be substantially (e.g., 10 times) higher than the impedance magnitude |Z11,0| as presented at the nominal sense frequency. Apart from the impedance transformation by the factor nT2, the transformer 736 may exert a negligible impact on the impedance |Z11,0| at the nominal sense frequency.
In an example parallel resonant configuration of the sense circuit 731, the reactance of the differential-mode series inductor 734 together with the transformer's 736 secondary referred leakage inductance Lσ undercompensates for the reactance of the double-ended sense electrode 732 at the nominal sense frequency. The residual capacitive susceptance of the series connection of the differential-mode series inductor 734, the transformer's 736 leakage inductance Lσ, and the double-ended sense electrode 732 is substantially compensated for by the susceptance of the transformer's 736 secondary referred inductance Lm providing an admittance Y11,0 that is substantially real (resistive). In this configuration, the inductance Lm may be smaller, similar, or larger than the inductance Ls of the differential-mode series inductor 734. Stated in other terms, the primary referred open-circuit admittance of the transformer 736 may be substantially (e.g., 20 times) higher than the admittance magnitude |Y11,0| as presented at the nominal sense frequency. In this configuration and apart from the admittance transformation, the transformer 736 exerts a significant impact on the admittance Y11,0 at the nominal sense frequency.
In a further aspect, the double-ended sense electrode's 732 capacitance C in combination with the transformer's 736 main inductance Lm form a 2nd order high pass filter to attenuate a low frequency disturbance component in the voltage V for purposes as previously discussed with reference to
In some implementations, the differential-mode series inductor 734 is configured to also provide a common mode inductance e.g., to attenuate a disturbance signal component in the voltage V emanating from a common mode current capacitively and inductively coupled into the double-ended sense electrode 732 (e.g., via capacitances Cew,a and Cew,b) during wireless power transfer.
The transformer 736 may serve for various purposes. In some implementations, the transformer 736 is used to match the impedance Z11 of the sense circuit 731 with an operating impedance range as previously discussed with reference to
Apart from the transformation ratio nT:1, the inductance ratio Ls/Lm may be an additional parameter to match the admittance magnitude |Y11,0| of the parallel resonant configuration with an operating admittance range of the multi-purpose detection circuit 100 with reference to
With reference to
With reference to
In another aspect, the use of a double-ended sense electrode (e.g., double-ended sense electrode 732) may reduce a disturbance voltage component in the voltage V e.g., emanating from the voltage capacitively coupled into the sense electrode by the electric field as generated during wireless power transfer. Due to its symmetry, the double-ended sense electrode 732 integrated into a wireless power transfer structure (e.g., wireless power transfer structure 200 of
In a further aspect, the electromagnetic emissions produced by the double-ended sense electrode 732 when driving the sense circuit 731 with a current I0 may be substantially lower than that of the equivalent single-ended sense electrode 702 when driving the sense circuit 701 with the same current I0. Therefore, if the drive current to is emission constraint (e.g., for frequency regulatory reasons), the sense circuit 731 may be driven with a substantially higher current I0 than the sense circuit 701 improving the SNR to compete with the sense circuit 701 using the single-ended sense electrode 702.
The circuit 740 of
The circuit 740 may be considered as electrically dual to the circuit 700 of
In an example implementation (not shown herein), the non-ground-based current source is accomplished by using a ground-based current source with an output transformer providing galvanic isolation.
As previously discussed with reference to the circuit 700 of
In an example parallel resonant configuration of the sense circuit 741, the susceptance of the parallel inductor 744 substantially compensates for the susceptance of the sense electrode 702 at the nominal sense frequency providing an admittance Y11,0 that is substantially real (resistive). In this configuration, the capacitance Cs of the series capacitor 746 may be similar or larger than the capacitance C of the sense electrode 702. Stated otherwise, the admittance magnitude of the series capacitor 746 may be substantially (e.g., 10 times) higher than the admittance magnitude |Y11,0| as presented at the nominal sense frequency. In this configuration, the series capacitor 746 may exert a negligible impact on the admittance |Y11,0| at the nominal sense frequency.
In an example series resonant configuration of the sense circuit 741, the susceptance of the parallel inductor 744 overcompensates for the susceptance of the sense electrode 702 at the nominal sense frequency. The residual inductive reactance of the parallel connection of the parallel inductor 744 and the sense electrode 702 is substantially compensated for by the reactance of the series capacitor 746 providing an impedance Z11,0 that is substantially real (resistive). In this configuration, the capacitance Cs of the series capacitor 746 may be smaller, similar, or larger than the capacitance C of the sense electrode 702. Stated otherwise, the impedance magnitude of the series capacitor 746 may be substantially (e.g., 20 times) higher than the impedance magnitude |Z11,0| as presented at the nominal sense frequency. In this configuration, the series capacitor 746 exerts a significant impact on the impedance Z11,0 at the nominal sense frequency.
In some implementations, the series capacitor 746 together with the parallel inductor 744 are used for purposes of resonance tuning and impedance transformation e.g., to transform the impedance Z11 to match the sense circuit 741 with an operating impedance range as previously mentioned with reference to
Impedance transformation may be particularly effective, if the sense circuit 741 is configured for series resonance. More specifically, increasing the capacitance ratio C/Cs, while maintaining series resonance at the nominal sense frequency, may substantially increase the impedance magnitude |Z11,0| at the nominal sense frequency. Therefore, in an aspect, the sense circuit 741 in the series resonant configuration may be considered as an alternative to the sense circuit 711 of
Increasing the capacitance ratio C/Cs, while maintaining resonance at the nominal sense frequency, may also somewhat decrease the admittance magnitude |Y11,0| as presented at the nominal sense frequency in the parallel resonant configuration of the sense circuit 741. However, impedance transformation may be limited and far less effective than that of the series resonant configuration.
In a further aspect, the sense electrode's 702 capacitance C in combination with the parallel inductor 744 and the series capacitor 746 form a higher order high pass filter for purposes as previously discussed in connection with
With reference to
The fractional change ΔY′ (or ΔZ′) as defined by Equations (8) and (9) and with respect to a defined test object (e.g., object 112) placed at a defined position relative to the sense electrode 702 may relate to the detection sensitivity of an object detection circuit (e.g., the multi-purpose detection circuit 100 of
As non-limiting examples, the fractional change may be increased by optimizing the design of the sense electrode 702 with respect to its geometry and its integration into the wireless power transfer structure (e.g., wireless power transfer structure 200 with reference to
As previously discussed with reference to the circuit 700 of
In some implementations and configurations of the circuit 740 of
Susceptance compensation in the sense circuit 741 exhibiting a local extremum (minimum or maximum) in the admittance magnitude function |Y11,0(ω)| and hence in the resulting current magnitude |I| at the measurement port 748 provides a mean to calibrate the current measurement circuit 550 and hence the admittance measurement with respect to the angle arg{ΔY}.
In a first step of an example calibration procedure applicable to the parallel resonant configuration of the circuit 740 of
Applying the angle correction of Equation (15), an object (e.g., object 114) reflecting an admittance ΔYr that is imaginary (reactive) may result in a measured current change ΔIcal that is substantially imaginary. Nevertheless, a residual error may remain in the angle arg{ΔIcal} due to the impact of the series capacitor 746 and the electrical losses in the sense circuit 741. The residual angle error of an example parallel resonant configuration of the circuit 740 and for an example object 110 is provided in TABLE 4.
In some implementations, the residual error is reduced by configuring the series capacitor 746 with a capacitance Cs whose admittance YCs is substantially larger (e.g., 10 times larger) than the parallel resonant conductance of the sense circuit 741. In other implementations, the residual error is reduced by measuring the admittance Y11,0 at two or more substantially different frequencies and by determining the elements of an equivalent circuit model of the sense circuit 741 (e.g., the equivalent circuit model illustrated in
In an implementation configured for series resonance as defined above, the circuit 540 may be configured to measure the impedance Z11 and corresponding changes ΔZ of Z11 as caused by the object 110, 112, 114, or vehicle 330. In this case, the impedance change ΔZ may be indicative of the reflected admittance ΔYr as previously introduced. As discussed above with reference to the parallel resonant configuration, the angle arg{ΔZ} may be subjected to an error and therefore may require calibration to reduce an error in the measurement of the angle arg{ΔZ} and thus of the angle arg{ΔYr}.
In an implementation configured for series resonance, the circuit 740 may be calibrated analogously to the parallel resonant configuration however using the local minimum of the impedance function |Z11,0(ω)| where reactance compensation occurs.
In a first step of an example calibration procedure applicable to the series resonant configuration of the circuit 740 of
Applying the angle correction of Equation (15), an object (e.g., object 114) reflecting an admittance ΔYr that is imaginary (reactive) may result in a measured current change ΔIcal that is substantially imaginary. Nevertheless, a residual error may remain in the angle arg{ΔIcal} due to the transformation of ΔYr to ΔZ in the lossy sense circuit (e.g., sense circuit 741). The residual angle error of an example series resonant configuration of the circuit 540 and for an example object 110 is provided in TABLE 4.
In an example implementation, the residual error due to the transformation of ΔYr to ΔZ is reduced by measuring the impedance Z11,0 at two or more substantially different frequencies, supposing absence of a foreign object, and by determining the elements of an equivalent circuit model (e.g., the equivalent circuit model of
In some implementations and configurations, the change ΔY in the admittance Y11, if correctly measured at the measurement port 748, directly relates to the reflected admittance ΔYr as previously defined. Therefore, in an aspect of object discrimination, the circuit 740 may be configured to determine the angle arg{ΔY} and thus the angle arg{ΔYr} with sufficient accuracy. However, in some implementations, measuring the admittance Y11 including the change ΔY may be subject to errors for various reasons. In particular, there may exist an unknown phase error between the generated sense voltage V0 as generated by the voltage source 552 and the current I as measured by the current measurement circuit 550 causing an error in the angle arg{Y11} and thus in the admittance change ΔY related to ΔYr.
Analogously to reactance compensation in the sense circuit 701 of
In an implementation configured for series resonance as defined above, the circuit 740 may be configured to measure the impedance Z11 and corresponding changes ΔZ of Z11 as caused by the object 110, 112, 114, or vehicle 330. However, as opposed to the parallel resonant configuration, the angle arg{ΔZ} may disagree with the angle arg{ΔZr} of the reflected impedance as previously defined with reference to
In an example implementation configured for series resonance, calibration is performed by measuring the impedance Z11,0 at substantially different frequencies, supposing absence of a foreign object, and by determining the elements of an equivalent circuit model (e.g., the equivalent circuit model of
In an implementation variant of the circuit 740 of
In an implementation variant of the circuit 740 of
The circuit 750 of
The circuit 750 is a modification of the circuit 740 of
Though not indicated in
As previously discussed with reference to the circuit 700 of
In some implementations, the transformer's 757 main inductance Lm, the series capacitor 756, and the parallel inductor 754 are used for purposes of resonance tuning and impedance transformation, e.g., to transform the impedance Z11 to match the sense circuit 711 with an operating impedance range as previously mentioned with reference to
In some implementations, the transformer 757 is a 1:1 transformer and serves for balancing. In other implementations, it is a nT:1 transformer (nT≠1) and is also used for impedance transformation.
In a further aspect, the double-ended sense electrode's 732 capacitance C in combination with the parallel inductor 754, the series capacitor 756, and the transformer's 757 main inductance Lm form a higher order high pass filter to attenuate a low frequency disturbance component in the current I for purposes as previously discussed in connection with
With reference to
In an implementation variant of the circuit 750 of
The circuit 760 of
The sense circuit 761 comprises a double-electrode capacitive sense element 762 comprising a first single-ended sense electrode 762a having a single terminal 763a and a second single-ended-sense electrode 762b having a single terminal 763b. The sense circuit 761 further comprises a first series inductor 764 having an inductance Ls,1 electrically connected in series to the first sense electrode 762a at the terminal 763a and a second series inductor 765 having an inductance Ls,2 electrically connected in series to the second sense electrode 762b at the terminal 763b. The sense circuit 761 further comprises a first parallel inductor 766 having an inductance Lp,1 electrically connected to the first series inductor 764 and in parallel to the measurement port 768 and a second parallel inductor 767 having an inductance Lp,2 electrically connected to the second series inductor 765 and in parallel to the measurement port 769. The circuit 760 further illustrates the sense signal current source 512 connected to the measurement port 768 and the voltage measurement circuit 510 connected to the measurement port 769.
Though not indicated in
Analogously to the self-inductances L1, L2, and the mutual inductance LM of a two-port inductive sense element (e.g., inductive sense element 562 of
Neglecting any effect of R1, R2, and RM, the following relations may apply between the capacitances Cag, Cab, Cbg and the capacitances C1, C2, CM:
C
1
=C
ag
+C
ab (182)
C
2
=C
bg
+C
ab (183)
CM=Cab (184)
Analogously to the inductive coupling factor, a capacitive coupling factor may be defined as:
k
C
=C
M(C1 C2)−1/2 (185)
Substituting C1, C2, CM in Equation (185) by Equations (182), (183), and (184) provides:
k
C
=C
ab(Cag+Cab)−1/2(Cbg+Cab)−1/2 (186)
Analogously to the “T”-equivalent circuit model 562-1 of a two-port inductive sense element (e.g., inductive sense element 562) illustrated in
Iind,1=jω CM V2 (187)
Iind,2=jω CM V1 (188)
representing the currents induced into the primary and secondary electrodes, respectively, as indicated in
In some implementations, the reactance of Ls,1 substantially compensates for the reactance of C1 providing a local impedance minimum |Z11| (series resonance) substantially at the nominal sense frequency, while the reactance of Ls,2 substantially compensates for the reactance of C2 L2 providing a local impedance minimum |Z22| (series resonance) substantially at the nominal sense frequency.
In another implementation, the sense circuit 761 is configured to provide a local minimum of the admittance magnitude functions |Y11(ω)| and |Y22(ω)| (parallel resonance) substantially at the nominal sense frequency.
In a further implementation, the sense circuit 761 is configured to provide a local minimum of the admittance magnitude function |Y11(ω)| (parallel resonance) and a local minimum of the impedance magnitude function |Z22(ω)| (series resonance) substantially at the nominal sense frequency.
In yet another implementation, the sense circuit 761 is configured to provide a local minimum of the impedance magnitude function |Z11(ω)| (series resonance) and a local minimum of the admittance magnitude function |Y22(ω)| (parallel resonance) substantially at the nominal sense frequency.
In implementations configured for primary-side and secondary-side series resonance, the reactance of the parallel inductors 766 and 767 is substantially higher than the impedance magnitudes |Z11| and |Z22|, respectively, of the sense circuit 761 at the nominal sense frequency.
In a further example implementation, at least one of the series inductors 764 and 765 is omitted and the sense circuit 761 is operated as a non-resonant or partially resonant circuit.
In a further aspect, the capacitance C1 of the first sense electrode 762a in combination with the first parallel inductor 766 form a 2nd order high pass filter to attenuate a low frequency disturbance component in the voltage V1. Likewise, the capacitance C2 of the second sense electrode 762b form a 2nd order high pass filter to attenuate a low frequency disturbance component in the voltage V2 for purposes as previously discussed in connection with
With reference to
Though not shown herein, other transimpedance measurement techniques such as the voltage source current measurement technique or any other combination may apply (e.g., a voltage source voltage measurement technique). In some implementations (also not shown herein), at least one of the impedances Z11 and Z22 of the sense circuit 781 is additionally measured to the transimpedance Z21 (e.g., using one or more of the techniques as previously discussed with reference to
Moreover, at least one of an impedance transformation and balancing may apply to at least one of the primary-side and secondary-side of the sense circuit 761 (not shown herein). More specifically, with reference to the sense circuit 721 of
With reference to
Using a quasi-ideal current source 512, a change ΔZ in the transimpedance Z21 (e.g., due to presence of the object 114) manifests in a change ΔV in the voltage V2 while the current I0,1 remains substantially unaffected. Therefore, measuring the complex voltage V2 may be equivalent to measuring the complex transimpedance Z21. In other words, the complex voltage V2 may be indicative of the complex transimpedance Z21 and there may be no requirement for additionally measuring the current I0,1 thus reducing complexity of the measurement circuit (e.g., measurement circuit 104 of
As with the sense circuit 561 of
As non-limiting examples, the fractional change ΔZ′ (or ΔY′) may be increased by optimizing the design and the arrangement of the sense electrodes 762a and 762b, their integration into the wireless power transfer structure (e.g., wireless power transfer structure 200 with reference to
The circuit 770 of
As opposed to the sense circuit 761, the sense circuit 771 is operated in a differential mode. It may be split into a first branch and a second branch with an equal topology. The sense circuit 771 may be substantially symmetric (balanced) with respect to its capacitances and inductances. The sense circuit 771 includes a quad-electrode capacitive sense element 772 comprising a first double-ended sense electrode composed of sense electrodes 772a and 772b having respective terminals 773a and 773b and a second double-ended sense electrode composed of sense electrodes 772c and 772d having respective terminals 773c and 773d. The first double-ended sense electrode 772a/b is electrically connected to a differential-mode inductor 774 providing an inductance Ls,1/2 in each branch. The second double-ended sense electrode 772c/d is electrically connected to a differential-mode inductor 775 providing an inductance Ls,2/2 in each branch. The sense circuit 771 also includes a transformer 776 with a transformation ratio n1:1 and a secondary referred main inductance Lm,1. Its primary winding is electrically connected in parallel to the measurement port 778, while its secondary winding is electrically connected to the differential-mode inductor 774. Further, the sense circuit 771 includes a transformer 777 with a transformation ratio 1:n2 and a primary referred main inductance Lm,2. Its primary winding is electrically connected to the differential-mode inductor 775, while its secondary winding is electrically connected in parallel to the measurement port 779. The circuit 770 further illustrates the sense signal current source 512 electrically connected to the measurement port 778 and the voltage measurement circuit 510 electrically connected to the measurement port 779.
Though not indicated in
Further,
As with the capacitive sense element 762 of
The following relations between the capacitances Cab, Cac, Cad, Cbc, Cbd, Ccd and C1, C2, CM may be found:
C
1
=C
ab+((Cac+Cad)(Cbc+Cbd)/ΣC) (189)
C
2
=C
cd+((Cac+Cbc)(Cad+Cbd)/ΣC) (190)
C
M=(Cac Cbd−Cad Cbc)/ΣC (191)
with ΣC denoting the sum of the coupling capacitances:
ΣC=(Cac+Cad+Cbc+Cbd) (192)
For an entirely symmetric capacitive sense element 772 with capacitances:
Cab=Ccd=Ca (193)
Cac=Cdb=Cb (194)
Cad=Cbc=Cc (195)
the mutual capacitance of Equation (191) becomes:
C
M=(Cb−Cc)/2 (196)
and the self-capacitances of Equations (189) and (190):
C
1
=C
2
=C
a+((Cb+Cc)/2) (197)
and the capacitive coupling factor:
k
C=(Cb−Cc)/(2Ca+Cb+Cc) (198)
In some implementations, the reactance of Ls,1 substantially compensates for the reactance of C1 providing a local impedance minimum |Z11| (series resonance) substantially at the nominal sense frequency, while the reactance of Ls,2 substantially compensates for the reactance of C2 providing a local impedance minimum |Z22| (series resonance) substantially at the nominal sense frequency.
In another implementation, the sense circuit 771 is configured to provide a local minimum of the admittance magnitude functions |Y11(ω)| and |Y22(ω)| (parallel resonance) substantially at the nominal sense frequency.
In a further implementation, the sense circuit 771 is configured to provide a local minimum of the admittance magnitude function |Y11(ω)| (parallel resonance) and a local minimum of the impedance magnitude function |Z22(ω)| (series resonance) substantially at the nominal sense frequency.
In yet another implementation, the sense circuit 771 is configured to provide a local minimum of the impedance magnitude function |Z11(ω)| (series resonance) and a local minimum of the admittance magnitude function |Y22(ω)| (parallel resonance) substantially at the nominal sense frequency.
In implementations configured for primary-side and secondary-side series resonance, the reactance of the transformer's 776 primary referred main inductance and the transformer's 776 secondary referred main inductance is substantially higher than the impedance magnitudes |Z11| and |Z22|, respectively, of the sense circuit 771 at the nominal sense frequency.
In a further example implementation, at least one of the differential-mode inductors 774 and 775 is omitted and the circuit 771 is operated as a non-resonant or partially resonant circuit.
In a further aspect, the capacitance C1 of the first double-ended sense electrode 772a/b in combination with the first transformer's 776 main inductance Lm,1 form a 2nd order high pass filter to attenuate a low frequency disturbance component in the voltage V1. Likewise, the capacitance C2 of the second double-ended sense electrode 772c/d in combination with the second transformer's 777 main inductance Lm,2 form a 2nd order high pass filter to attenuate a low frequency disturbance component in the voltage V2 for purposes as previously discussed in connection with
With reference to
Though not shown herein, other transimpedance and impedance measurement techniques as previously discussed with reference to
With reference to
As with the sense circuit 561 of
As non-limiting examples, the fractional change ΔZ′ (or ΔY′) may be increased by optimizing the design and the arrangement of the double-ended sense electrodes 772a/b and 772c/d, their integration into the wireless power transfer structure (e.g., wireless power transfer structure 200 with reference to
In an example implementation and in analogy to the implementation of the inductive sense element 562 as described with reference to
Cac Cbd≈Cad Cbc (199)
For an entirely symmetric capacitive sense element 772 using Equation (196), the mutual capacitance vanishes if:
Cb≈Cc (200)
An example implementation of a capacitive sense element 772 using an arrangement of four single-ended sense electrodes that may provide a substantially zero mutual capacitance (CM≈0) is illustrated in
The circuit 780 of
The sense circuit 781 includes the double-electrode capacitive sense element 762 with reference to
Though not indicated in
The basic topology of the circuit 780 of
In an example implementation, the sense electrodes 762a and 762b are tightly coupled resulting in a capacitive coupling factor kc as defined by Equation (186) that is near unity (kC≈<1).
The sense circuit 781 may be configured to provide a local minimum in the transimpedance magnitude function |Z21,0(ω)| (series resonance) substantially at a nominal sense frequency. Alternatively, the sense circuit 581 may be configured to provide a local minimum in the transadmittance magnitude function |Y21,0(ω)| substantially at the nominal sense frequency.
In an example parallel resonant configuration of the sense circuit 781 using a capacitive sense element 762 with kC26 <1, the susceptance of the parallel inductor 784 substantially compensates for the susceptance of the mutual capacitance CM providing a local minimum in the transadmittance magnitude function |Y21,0(ω)| (parallel resonance) substantially at the nominal sense frequency. The principle of mutual susceptance compensation may become more evident by contemplating
In this parallel resonant configuration, the capacitances Cs,1 and Cs,2 of the series capacitor 786 and 787, respectively, may be similar or larger than the capacitance C1 and C2 of the sense electrodes 762a and 762b, respectively. Stated in other terms, the admittance magnitude of the series capacitor 786 and 787 may be substantially higher than the admittance magnitude |Y11| and |Y22|, respectively, of the sense circuit 781 at the nominal sense frequency. In this configuration, the series capacitor 786 and 787 may exert a negligible impact on the admittances and transadmittance |Y11|, |Y22|, and |Y21|, respectively, at the nominal sense frequency.
In an example series resonant configuration of the sense circuit 781 using a capacitive sense element 762 with kL≈<1, the susceptance of the parallel inductor 784 overcompensates for the susceptance of the mutual capacitance CM at the nominal sense frequency. The residual inductive reactance of the parallel connection of the parallel inductor 784 and the mutual capacitance CM is substantially compensated for by the reactance of the series capacitors 786 and 787 providing a transimpedance Z21,0 that is substantially real (resistive).
In this series resonant configuration, the capacitances Cs,1 and Cs,2 of the series capacitors 786 and 787, respectively, may be smaller, similar, or larger than the capacitances C1 and C2 of the sense electrodes 762a and 762b, respectively. Stated in other terms, the impedance magnitude of each of the series capacitors 786 and 787 may be substantially (e.g., 20 times) higher than the impedance magnitudes |Z11| and |Z22|, respectively, as presented at the nominal sense frequency. In this configuration, the series capacitors 786 and 787 exert a significant impact on the impedance and transimpedance magnitudes |Z11|, |Z22|, and |Z21|, respectively, at the nominal sense frequency.
In a further aspect, the capacitance C1 of the first sense electrode 762a in combination with the first series capacitor 786 form a high pass filter to attenuate a low frequency disturbance component in the current I1. Likewise, the capacitance C2 of the second sense electrode 762b in combination with the second series capacitor 787 form a high pass filter to attenuate a low frequency disturbance component in the current I2 for purposes as previously discussed in connection with
In yet a further aspect, the capacitance CM of the capacitive sense element 762 in combination with the parallel inductor 784 form a 2nd order high pass filter to attenuate a differential low frequency disturbance voltage between the sense electrodes 762a and 762b.
With reference to
Though not shown herein, other transimpedance and impedance measurement techniques may apply as previously mentioned with reference to
Moreover, as previously mentioned with reference to
With reference to
As with the sense circuit 561 of
It may be appreciated that the sense circuit 781 configured for a capacitive coupling factor kC≈<1 reduces the impact of the equivalent series resistances R1 and R2 on the fractional change, if compared e.g., to the circuit 700 of
For purposes of comparison, an identical sense electrode 702 an equal sense electrode current level |IC| is assumed for both configurations of the circuits 700, 710, 720, and 740, though practical implementations configured for parallel resonance may prefer a sense electrode 702 with a higher capacitance C. Comparing SNRs at the same sense electrode current level |IC| may be meaningful e.g., if the current level |IC| is emission or power constraint. Further, it is assumed that the circuits in both configurations are adjusted to a common resonant frequency substantially corresponding with the nominal sense frequency that is substantially higher than the WPT operating frequency.
The equivalent circuit model 700-1 as illustrated in
It may be appreciated that the equations for the series and parallel resonant configuration of the sense circuit 701 with respect to the impedance Z11, the admittance Y11, the respective resonant angular frequencies ωs and ωp, the impedance change ΔZ, the admittance change ΔY, the fractional changes ΔZ′ and ΔY′, and the various circuit intrinsic and extrinsic SNRs as previously defined with reference to
To analyze the series and parallel resonant configuration of the circuit 700 of
ω Ls>>RLs (201)
ω Lp>>RLp (202)
|ΔZr|<<R (203)
are made for a frequency range about the resonant frequency.
In an implementation configured for series resonance and with a reactance:
ω Lp>>|Z11| (204)
in a frequency range about the resonant frequency, the impedance Z11 at the measurement port 708 of the circuit 700 of
Z11≈RLs+jω Ls+(jω C)−1+ΔZr (205)
In absence of a foreign object, a local minimum of |Z11,0(ω)| (series resonance) occurs substantially at an angular frequency ω satisfying:
(jω Ls)−1+jω C≈0 (206)
yielding the series resonant angular frequency:
ωs≈(C Ls)−1/2 (207)
At this frequency, the impedance Z11,0 becomes substantially real:
Z
11,0
≈Re{Z
11,0
}=R
s
≈R
Ls (208)
with Rs denoting the series resonant resistance, while the impedance Z11 in presence of an object (e.g., object 112) is approximately:
Z11≈Rs+ΔZ≈RLs+ΔZr (209)
with ΔZr referring to the reflected impedance as previously defined with reference to
The fractional change ΔZ′ for the series resonant configuration of the circuit 700 of
ΔZ′=ΔZ/Rs≈ΔZr/RLs (210)
Defining the normalized reflected impedance:
ΔZ′r=ΔZr ωs C (211)
the Q-factor of the series inductor 704:
Q
Ls=ωs Ls/RLs (212)
and the Q-factor of the series resonant configuration of the sense circuit 701 of
Qs≈1/(ωs C Rs)≈QLs (213)
the fractional change may also be written in terms of ΔZ′r and Qs or QLs:
ΔZ′≈Qs ΔZ′r≈QLs ΔZ′r (214)
To analyze the parallel resonant configuration of the equivalent circuit model 700-1 of
ω Ls−(ω C)−1|>>RLs (215)
is made for a frequency range about the resonant frequency. The admittance Y11 at the measurement port 708 in presence of an object (e.g., object 112) may be expressed:
Y
11=(RLp+jω Lp)−1+(RLs+jω Ls+(jω C)−1+ΔZr)−1 (216)
and using Equation (38) approximated as:
Y11≈(jω Lp)−1−RLp(jω Lp)−2+(jω Ls+(jω C)−1)−1−(RLs+ΔZr)(ω Ls+(ω C)−1)−2 (217)
In absence of a foreign object, a local minimum of |Y11,0(ω)| (parallel resonance) occurs substantially at an angular frequency ω satisfying:
(jω C)−1+jω(Ls+Lp) (218)
yielding for the parallel resonant angular frequency:
ωp≈(C(Ls+Lp))−1/2 (219)
At this frequency, the admittance Y11,0 becomes substantially real:
Y
11,0
≈Re{Y
11,0
}=G
p≈(RLs+RLp)/(ωp Lp)2 (220)
with Gp denoting the parallel resonant conductance, while the admittance Y11 in presence of an object (e.g., object 112) is approximately:
Y11≈Gp+ΔY≈(RLs+RLp+ΔZr)/(ωp Lp)2 (221)
with:
ΔY≈ΔZr/(ωp Lp)2 (222)
Defining the Q-factor of the series inductor 704 in terms of Ls and RLs:
QLs≈ωp Ls/RLs (223)
the Q-factor of the parallel inductor 706 in terms of Lp and RLp:
QLp≈ωp Lp/RLp (224)
and the inductance ratio:
n
L
=L
s
/L
p (225)
the admittance Y11,0 at ωp may be expressed as:
Y
11,0
=G
p≈(1+nL)((1/QLp)+(nL/QLs))ωp C (226)
For QLp=QLs and nL>1, the parallel resonant conductance Gp as presented at the measurement port 708 becomes approximately (1+nL)2 ωp C/QLs. Using Equations (219) and (225), the admittance Y11 at ωp of Equations (221) may also be expressed as:
Y11≈Gp+ΔY≈Gp+(1+nL)2(ωp C)2 (227)
According to Equation (226), the admittance Y11 at ωp of Equation (225) of the sense circuit 701 of
Based on Equation (9), the fractional change ΔY′ for the parallel resonant configuration of the circuit 700 of
ΔY′=ΔY/Gp≈ΔZr/(RLs+RLp) (228)
showing that the admittance change ΔY is substantially proportional to the reflected impedance ΔZr. Therefore, the angle arg{ΔY} of the measured admittance change ΔY is indicative of the angle arg{ΔZr}. In some implementations, the accuracy of the measured angle is improved by applying the angle correction based on the calibration procedure as previously described with reference to the circuit 500 of
Qp≈ωp(Lp+Ls)/(RLs+RLp) (229)
which may be expresses in terms of the Q-factors:
Qp≈QLs(1+nL)/((QLs/QLp)+nL) (230)
and the normalized reflected impedance:
ΔZ′r=ΔZr ωp C (231)
the fractional change ΔY′ may also be written as:
ΔY′≈Qp ΔZ′r (232)
With reference to the circuit 710 of
Z11≈RLs+jω Ls+(jω Cp+((jω C)−1+ΔZr)−1)−1 (233)
Using the approximation of Equation (38), the impedance Z11 may be approximated as:
Z11≈RLs+jω Ls+(jω(C+Cp))−1+ΔZr C2/(C+Cp)2 (234)
Series resonance in absence of a foreign object occurs approximately at an angular frequency satisfying:
(jω(C+Cp))−1+jω Ls≈0 (235)
yielding the series resonant angular frequency:
ωs≈(Ls(C+Cp))−1/2 (236)
At this frequency, the impedance Z11,0 becomes substantially real:
Z
11,0
≈Re{Z
11,0
}=R
s
≈R
Ls (237)
with Rs denoting the series resonant resistance, while the impedance Z11 in presence of an object (e.g., object 110) is approximately:
Z11≈Rs+ΔZ≈RLs+ΔZr C2/(C+Cp)2 (238)
Defining the capacitance ratio:
n
C
=C/C
p (239)
the impedance Z11,0 of Equation (238) may also be expressed in terms of the Q-factor QLs of the series inductor 714, the capacitance C of the sense electrode 702, and the capacitance ratio nC as:
Z11,0≈Rs≈nC/((1+nC)ωs C QLs) (240)
According to Equation (240), the impedance Z11 of the sense circuit 711 of
Based on Equation (238), the impedance change ΔZ resulting at the measurement port 718 of the sense circuit 711 of
ΔZ≈(nC/(1+nC))2 ΔZr (241)
Defining the Q-factor of the series resonant configuration of the sense circuit 711 of
Q
s
≈ω
s
L
s
/R
s
=Q
Ls (242)
the fractional change ΔZ′ may be expressed as:
ΔZ′≈ΔZ/Rs≈nC/(1+nC)Qs ΔZ′r (243)
Based on Equation (243), adding a parallel capacitor 715 (as shown in
In some implementations based on the circuit 700 of
I0≈|IC| (244)
resulting in a voltage across the measurement port 708:
V≈|Z11,0|I0≈RLs IC (245)
and in a drive power level:
P≈V I0≈RLs|IC|2 (246)
For the parallel resonant configuration of the circuit 700 of
I0≈|IC|(ωp Ls−(1/ωp C))Gp≈|IC|ωp Lp Gp≈|IC|(1+nL)/Qp (247)
The voltage across the measurement port 708 becomes approximately:
V≈I0/|Y11,0|≈|IC|ωp Lp (248)
and the power:
P≈V I0≈(RLs+RLp)|IC|2 (249)
In a further aspect, the differential narrowband extrinsic SNR of the series resonant configuration of the circuit 700 of
ΔSNRex,s≈|ΔZr||IC|/Vsn≈(|IC|/Vsn)|ΔZ′r|/(ωs C) (250)
with |IC| denoting the magnitude of the sense signal current in the sense electrode 702, which approximately equals the current magnitude |I0|, and Vsn the noise voltage capacitively coupled into the sense electrode 702 as illustrated in
Since the sense circuit 701 transforms the voltage drop across ΔZr to ΔV in the same way as it transforms Vsn to Vn, Equation (250) also applies to the parallel resonant configuration as well as to the series and parallel resonant configuration of the circuit 700 of
ΔSNRex,p=ΔSNRex,s (251)
Equations (250) and (251) indicate that the differential narrowband extrinsic SNR of the series and parallel resonant configuration of the circuit 700 of
Adding the capacitor 715 with reference to the circuit 710 of
In implementations and operations where the noise current I0,n as indicated in
Vn≈RLs I0,n (252)
while the voltage change |ΔV| in presence of an object (e.g., object 112) is:
|ΔV|=|IC||ΔZr|≈|I0||ΔZr| (253)
The differential narrowband intrinsic SNR with respect to the drive current noise I0,n for the series resonant configuration of the circuit 700 of
ΔSNRint,s≈(|I0|/I0,n)|ΔZr|/RLs=(|I0|/I0,n)|ΔZ′| (254)
Equation (254) may also be written in terms of the Q-factor Qs and ΔZ′r as:
ΔSNRint,s≈(|I0|/I0,n)Qs|ΔZ′r| (255)
With the noise current I0,n as the predominant contribution, the noise voltage Vn at parallel resonance becomes approximately:
Vn≈I0,n/|Y11,0| (256)
Using Equation (256) and applying Equation (70), the differential narrowband intrinsic SNR with respect to the noise current I0,n for the parallel resonant configuration of the circuit 700 of
ΔSNRint,p≈(|I0|/I0,n)|ΔZr|/(RLp+RLs)=(|I0|/I0,n)|ΔY′| (257)
Equation (257) may also be written in terms of the Q-factor Qp and ΔZ′r as:
ΔSNRint,p≈(|I0|/I0,n)Qp|ΔZ′r| (258)
which is a linear function of the Q-factor Qp.
Similar considerations may be made for the thermal noise though not repeated herein for the circuit 700 of
Since ΔSNRint,s and ΔSNRint,p are both proportional to the magnitude of the fractional change |ΔZ′| and since adding the capacitor 715 reduces the fractional change |ΔZ′| by the factor nC/(1+nC), the differential narrowband intrinsic SNR of the circuit 710 of
In another aspect, the broadband extrinsic SNR as defined by Equation (62) with respect to the induced voltage component VsW at the fundamental WPT operating angular frequency ωW is considered. Assuming electric field coupling as the predominant contribution, the disturbance signal voltage Vsn may relate to the WPT coil voltage VWPT as follows:
Vsn≈VsW≈(CsW/C)VWPT (259)
where CsW denotes the mutual capacitance between the sense electrode 702 and the WPT coil (e.g., WPT coil 202 with reference to
1/(ωW C)>>ωW Ls (260)
ωs>>ωW (261)
the disturbance voltage component VW in the voltage V for the series resonant configuration of the circuit 700 of
VW≈VsW ωW C ωW Lp≈VsW(ωW/ωs)2/nL (262)
The factor (ωW/ωs)2/nL may be considered as the attenuation of the low frequency induced voltage VsW by the high pass filter effect of the sense circuit 701. Using:
|V|=|I0|Rs≈|IC|RLs (263)
the broadband extrinsic SNR with respect to the fundamental WPT fundamental disturbance component VsW for the series resonant configuration of the circuit 700 of
SNRW,s≈(|IC|/VsW)(1/ωs C)(ωs/ωW)2 nL/Qs (264)
For a given ratio |IC|/VsW and susceptance ωs C, the broadband extrinsic SNR for the series resonant configuration of the circuit 700 of
The disturbance voltage component VW in the voltage V for the parallel resonant configuration of the circuit 700 of
V
n
=V
W
≈V
sW ωW C ωW Lp≈VsW(ωW/ωp)2/(1+nL) (265)
The factor (ωW/ωp)2/(1+nL) may be considered as the attenuation of the low frequency induced voltage VsW by the high pass filter effect of the sense circuit 701. Further, expressing the sense signal voltage |V| at the angular frequency ωp in terms of the sense electrode current |IC|:
|V|≈|IC|((ωp C)−1−ωp Ls)≈|IC|ωp Lp (266)
the broadband extrinsic SNR with respect to the WPT fundamental disturbance voltage component VsW for the parallel resonant configuration of the circuit 700 of
SNRW,p≈(|IC|/VsW)(1/ωp C)(ωp/ωW)2 (267)
For a given ratio |IC|/VsW and susceptance ωp C, the broadband extrinsic SNR for the parallel resonant configuration of the circuit 700 of
For ωs=ωp, the following relation may be found between the broadband extrinsic SNRs of the series and parallel resonant configurations of the circuit 700 of
SNRW,p≈SNRW,s Qs/nL (268)
where Qs refers to the Q-factor of the series resonant configuration of the circuit 700. From Equation (268), it can be seen that the broadband extrinsic SNR for the parallel resonant configuration of the circuit 700 of
It may be appreciated that adding the capacitor 715 with reference to
(ωW Cp)−1>>ωW(Ls+Lp) (269)
and:
ωs=ωp>>ωW (270)
holds. The same is true for the sense signal voltage |V| that may be expressed for the series resonant configuration of the circuit 700 of
|V|≈I0 Rs≈|IC|/(ωs C QLs) (271)
if the sense current level |IC| is maintained by adjusting I0 to:
I0≈|IC|(1+nC)/nC (272)
Therefore, it may be concluded that the broadband extrinsic SNR given by Equation (264) also applies to the series resonant configuration of the circuit 710 of
Equations (201) to (268) may also apply to the circuit 720 of
The equivalent circuit model 740-1 as illustrated in
With the assumption of an identical sense electrode 702 in the circuits 700 and 740, the following relations may apply:
ΔY′r=ΔZ′r (273)
ΔYr≈ΔZr(ω C)2 (274)
Isn≈Vsn ω C (275)
ΔY′r, ΔZ′r, ΔZr, and Vsn referring to the normalized reflected admittance, the normalized reflected impedance, the reflected impedance of the object 110 in the sense electrode 702, and the disturbance voltage Vsn with reference to the circuit 700 of
To analyze the series and parallel resonant configuration of the circuit 740 of
1/ω Lp>>GLp (276)
|ΔYr|<<GLp (277)
are made for a frequency range about the resonant frequency.
In an implementation configured for parallel resonance and with a susceptance:
ω Cs>>|Y11| (278)
in a frequency range about the resonant frequency, the admittance Y11 at the measurement port 748 of the circuit 740 of
Y11≈GLp+(jω Lp)−1+jω C+ΔYr (279)
In absence of a foreign object, a local minimum of |Y11,0(ω)| (parallel resonance) occurs substantially at an angular frequency ω satisfying:
(jω Lp)−1+jω C≈0 (280)
yielding the parallel resonant angular frequency:
ωp≈(Lp C)−1/2 (281)
At this frequency, the admittance Y11,0 becomes approximately real:
Y
11,0
≈Re{Y
11,0
}=G
p
≈G
Lp (282)
with Gp denoting the parallel resonant conductance, while the admittance Y11 in presence of an object (e.g., object 112) is approximately:
Y11≈Gp+ΔYr≈GLp+ΔYr (283)
with ΔYr referring to the reflected admittance as previously defined with reference to
The fractional change ΔY′ for the parallel resonant configuration of the circuit 740 of
ΔY′≈ΔYr/Gp≈ΔYr/GLp (284)
Defining the normalized reflected admittance as:
ΔY′r=ΔYr/(ωp C) (285)
the Q-factor of the parallel inductor 744:
Q
Lp=1/(ωp Lp GLp) (286)
and the Q-factor of the parallel resonant configuration of the sense circuit 741 of
Qp≈ωp C/Gp≈QLp (287)
the fractional change may also be written in terms of ΔY′r and Qp:
ΔY′≈Qp ΔY′r (288)
According to Equation (288) with (214), the parallel resonant configuration of the circuit 740 of
To analyze the series resonant configuration of the circuit 740 of
|ω Cp−(ω Lp)−1|>>GLp (289)
is made for a frequency range about the resonant frequency. The impedance Z11 at the measurement port 748 in presence of an object (e.g., object 112) may be expressed as:
Z
11=(jω Cs)−1+(GLp+(jω Lp)−1+jω C+ΔYr)−1 (290)
Using the approximation of Equation (38), the impedance Z11 may be approximated as:
Z11≈(jω Cs)−1+(jω C+(jω Lp)−1)−1+(GLp+ΔYr)/(ω C−(ω Lp)−1)2 (291)
In absence of a foreign object, a local minimum of |Z11,0(ω)| (series resonance) occurs substantially at an angular frequency ω satisfying:
(jω Lp)−1+jω(C+Cs)≈0 (292)
yielding for the series resonant angular frequency:
ωs≈(L(C+Cs))−1/2 (293)
At this frequency, the impedance Z11,0 becomes substantially real:
Z
11,0
≈Re{Z
11,0
}=R
s
=G
Lp/(ωs Cs)2 (294)
with Rs denoting the series resonant resistance, while the impedance Z11 in presence of an object (e.g., object 112) is approximately:
Z11≈Rs+ΔZ≈(GLp+ΔYr)/(ωs Cs)2 (295)
with:
ΔZ≈ΔYr/(ωs Cs)2 (296)
Further, the Q-factor of the parallel inductor 744 may be defined as:
Q
Lp=1/(ωs Lp GLp) (297)
and the capacitance ratio:
n
C
=C/C
s (298)
Equation (294) at ωs may be expressed as:
Z
11,0
=R
s
≈n
C(1+nC)/(QLp ωs C) (299)
For nC>>1, the series resonant resistance Rs becomes approximately:
Rs≈nC2/(QLp ωs C) (300)
According to Equation (299), the impedance Z11 at ωs of the sense circuit 741 of
The fractional change ΔZ′ for the series resonant configuration of the circuit 740 of
ΔZ′=ΔZ/Rs≈ΔYr/GLp (301)
According to Equation (301), the impedance change ΔZ is proportional to the reflected admittance ΔYr. Therefore, the angle arg{ΔZ} of the measured impedance change ΔZ is indicative of the angle arg{ΔYr}. In some implementations, the accuracy of the measured angle is improved by applying the angle correction based on the calibration procedure as previously described with reference to the circuit 700 of
The Q-factor of the series resonant configuration of the sense circuit 741 of
Q
s≈ωs(C+Cs)/GLp≈1/(ωs Lp GLp)=QLp (302)
which equals the Q-factor of the parallel inductor 744. Using definitions above, Equation (302) may also be expressed in terms of the series resonant resistance Rs, the electrode 702 capacitance C, and the capacitance ratio nC as:
Qs≈nC(1+nC)/(Rs ωs C) (303)
Further, the normalized reflected admittance may be defined as:
ΔY′r=ΔYr/(ωs C) (304)
The fractional impedance change ΔZ′ may also be written in terms of Qs and ΔY′r as:
ΔZ′≈Qs ΔY′r nC/(1+nC) (305)
In some implementations based on the circuit 740 of
V0≈|IC|/(ωp C) (306)
The current I at the measurement port 748 becomes approximately:
I≈|Y11,0|V0≈GLp V0 (307)
and the drive power level:
P≈V
0
I≈|I
C|2 GLp/(ωp C)2=|IC|2 QLp/(ωp C) (308)
For the series resonant configuration of the circuit 740 of
|VC|≈|IC|/(ωs C) (309)
yielding for the drive power level:
P≈|VC|2 GLp≈|IC|2 GLp/(ωs C)2≈|IC|2 QLp/(ωs C) (310)
which equals the drive power level of the parallel resonant configuration of the circuit 740 of
In a further aspect, it may be meaningful to define the narrowband SNR at the measurement port 748 of the circuit 740 of
In another aspect, it may be meaningful to define the broadband extrinsic SNR at the measurement port 748 of the circuit 740 of
Using Equation (14), the differential narrowband extrinsic SNR of the parallel resonant configuration of the circuit 740 of
ΔSNRex,p≈(|IC|/Isn)(|ΔYr|/ωp C)=(|IC|/Isn)|ΔY′r| (311)
with Isn the noise current as illustrated in
Since the sense circuit 741 transforms the shunt current through ΔYr to the current change ΔI in the same way as it transforms Isn to In, Equation (311) also applies to the series resonant configuration, meaning that:
ΔSNRex,s≈ΔSNRex,p (312)
In implementations with the noise voltage V0,n causing the predominant noise contribution in In as previously discussed, the noise current In for the parallel resonant configuration of the circuit 740 is approximately:
In≈Gp V0,n (313)
while the current change in presence of an object (e.g., object 112) is:
|ΔI|=|VL||ΔYr|≈|V0||ΔYr| (314)
Applying Equations above to Equation (14), the differential narrowband intrinsic SNR with respect to the noise voltage V0,n for the parallel resonant configuration of the circuit 740 of
ΔSNRint,p≈(|V0|/V0,n)|ΔYr|/Gp (315)
Equation (315) may also be written as:
ΔSNRint,p≈(|V0|/V0,n)Qp|ΔY′r (316)
Using:
ΔZ′<<1 (317)
the current change magnitude |ΔI| for the series resonant configuration of the circuit 740 of
|ΔI|≈|V0||ΔZ|/|Z11,0|2 (318)
With the noise voltage V0,n as the predominant noise contribution, the noise current In at series resonance becomes:
I
n
=V
0,n
/|Z
11,0| (319)
The differential narrowband intrinsic SNR with respect to the noise voltage V0,n for the series resonant configuration of the circuit 540 of
ΔSNRint,s≈(|V0|/V0,n)|ΔZ′|≈(|V0|/V0,n)|ΔYr|/GLp (320)
Equation (320) may also be written in terms of the Q-factor Qs and the normalized reflected admittance ΔY′r as:
ΔSNRint,s≈(|V0|/V0,n)Qs|ΔY′r|nC/(1+nC) (321)
Similar considerations may be made for the thermal noise though not repeated herein for the circuit 740 of
In a further aspect, the broadband extrinsic SNR as defined by Equation (147) with respect to the induced current component:
Isn=IsW (322)
at the fundamental WPT operating angular frequency ωW is considered. Further, assuming:
1/(ωW C)>>ωW Lp (323)
the disturbance current component IW in the current I for the parallel resonant configuration of the circuit 740 of
I
n
=I
W
≈I
sW ωW Lp ωW Cs≈IsW(ωW/ωp)2/nC (324)
The factor (ωW/ωp)2/nC may be considered as the attenuation of the low frequency induced current IsW by the high pass filter effect of the sense circuit 741. Using:
|I|≈|VC|GLp (325)
the broadband extrinsic SNR of the parallel resonant configuration of the circuit 740 of
SNRW,p≈(|VC|GLp/IsW)(ωp/ωW)2 nC (326)
with:
|VC|≈|IC|/(ωp C) (327)
representing the voltage across the sense electrode 702. Equation (326) may also be written in terms of the Q-factor Qp and the inductance ratio nC as:
SNRW,p≈(|IC|/IsW)(ωp/ωW)2 nC/Qp (328)
The disturbance current IW in the current I for the series resonant configuration of the circuit 740 of
IW≈IsW ωW L ωW Cs≈IsW(ωW/ωs)2/(1+nC) (329)
The factor (ωW/ωp)2/(1+nC) may be considered as the attenuation of the low frequency induced current IsW by the high pass filter effect of the sense circuit 741. Further, expressing the sense signal current |I| at the angular frequency co, in terms of the sense electrode 702 voltage |VC| is as follows:
|I|≈|VC|(ωs C−(ωs Lp)−1)≈|VC|ωs Cs (330)
The broadband extrinsic SNR with respect to the WPT fundamental disturbance current component IsW for the series resonant configuration of the circuit 740 of
SNRW,s≈(|VC|ωs Cs/IsW)(ωs/ωW)2(1+nC) (331)
Using the relation:
|VC|≈|IC|/ωs C (332)
Equation (331) may also be written as:
SNRW,s≈(|IC|/IsW)(ωs/ωW)2(1+nC)/nC (333)
Based on Equations (328) and (333) and ωs=wp, the following relation between the broadband extrinsic SNRs of the parallel and series resonant configurations of the circuit 740 of
SNRW,s≈SNRW,p Qp(1+nC)/nC2 (334)
A selection of equations with respect to the resonant frequency, the Q-factor, the impedance/admittance of the sense circuit, the fractional change, and the various SNRs for the series and parallel resonant configurations of the circuit 700 of
TABLE 3 provides example parameter values as used for a numerical analysis of the series and parallel resonant configuration of the circuit 700 of
Numerical results as obtained based on the numerical assumptions of TABLE 3 using the relevant equations as defined above with reference to
Based on numerical results listed in TABLE 4, the following conclusions may be drawn. The high impedance magnitude |Z11,0| as generally presented by the parallel resonant configuration of the circuit 700 of
The complex plane 800 and the shaded areas (e.g., angle ranges 802 and 804) may also apply to the reflected impedance ΔZr by simply relabeling the real and imaginary axis by Re{ΔZr} and jIm{ΔZr}, respectively (not shown in
Further,
In some implementations of the multi-purpose detection circuit 100, water dripping from a wet underbody of a vehicle (e.g., vehicle 330) onto the housing of a wireless power transfer structure (e.g., housing 328 of wireless power transfer structure 200 of
The effect of an object (e.g., object 110, 112, 114, or vehicle 330) proximate to a single-ended capacitive sense element (e.g., sense electrode 702) having a signal terminal 703 may be modeled empirically by a ground-related one-port equivalent circuit model 810 illustrated in
It is assumed that any real heterogenous object (e.g., object 110, 112, or 114) composed of different materials may be substituted by an artificial homogenous object consisting of an equivalent material having a complex relative permittivity defined as:
εr=ε′r−j ε″r (335)
where ε′r refers to the relative real permittivity and ε″r to the relative imaginary permittivity related to the electrical losses of the material. The relative imaginary permittivity may comprise a dielectric loss coefficient εd,r″ attributed to bound charge and dipole relaxation phenomena of the material and another loss coefficient attributed to the material's electrical conductivity σ. The relative imaginary permittivity may be defined as:
ε″r=εd,r″+σ/(ω ε0) (336)
where ω denotes the angular frequency and ε0 refers to the absolute permittivity of a vacuum. In case of an inhomogeneous object, the complex relative permittivity of the equivalent material may also depend on the size, geometry, position, and orientation of the real object relative to the capacitive sense element (e.g., sense electrode 702). The ratio εr″/εr′ is commonly known as the loss tangent of a dielectric material:
tan δ=(εd,r″+σ/(ω ε0))/ε′r (337)
where δ is also referred to as the loss angle. At frequencies below 10 MHz, the dielectric loss coefficient εd,r″ is normally negligible meaning that electrically insulating materials with virtually zero conductivity (σ→0) exhibit a loss angle close to zero (δ→0). For example, the imaginary relative permittivity of distilled water is ε″r≈0, while for water with some ion content, ε″r is governed by its conductivity σ. In the opposite extreme of an electrical conductor (e.g., metal object 110), tan δ may approach infinity (δ→90°).
In the equivalent circuit model 810 of
Y
ob
=G
ob
+jω C
ob
=jω C
ob,0
f(εr) (338)
where Cob,0 refers to the capacitance of a fictitious (“stealth”) object of a material with ε′r=1 and ε″r=0 (σ=0) that is indistinguishable from air and thus from the absence of a foreign object. From other theoretical considerations (not included herein), the function f(εr) may be of the form:
f(εr)=(εr+a−1)/a=(χ+a)/a (339)
The factor a is indicative of how easily a dielectric object (e.g., object 112) can be polarized by an externally applied electric field and herein referred to as polarizability. The polarizability a generally depends on the size, geometry, position, and orientation of the object relative to the sense electrode 702 and the ground. For objects made of an ordinary dielectric material, the polarizability must be in a range 1≤a<∞. The coefficient χ=εr−1 is known as the electric susceptibility of the dielectric material. Equation (339) satisfies f(1−j 0)=1, f(|εr|→∞)→∞ for any a in the range defined above. For a→∞, the function converges to 1 for any εr and ε′r>1.
In analogy to exposing a ferromagnetic object to a magnetic field, exposing a dielectric object (e.g., object 112) to an electric field polarizes its material in turn starting to generate its own (secondary) electric field referred to as a polarization field. This polarization field counteracts the applied field in the object's material until an equilibrium state is reached. This mechanism is also known as depolarization. Depolarization is commonly quantified by the depolarization factor in analogy to the magnetic depolarization in ferromagnetic bodies. The depolarization factor may be defined as the reciprocal 1/a of the polarization factor. For small a (e.g., 1≤a<<|εr|), the internal electric field (in the material) may be substantially reduced (e.g., virtually canceled out) by the polarization field. In contrast, a dielectric object with high a (e.g., a>>|εr|) experiences little depolarization, resulting in an internal field substantially equal to the applied field.
Polarizability of a thin dielectric sheet may be close to one (a→1) with respect to an electric field perpendicular to the sheet. For an oblate dielectric sphere subjected to a uniform electric field parallel to its short (minor) axis, the polarizability can be found in the range 1<a<3. The polarizability a is higher in any other axis and is thus anisotropic. The polarizability of a dielectric sphere is isotropic and a=3. For a dielectric ellipsoid, polarizability is again anisotropic and highest in its major axis (a>3). Therefore, the major axis is sometimes also referred to as the “easy” axis. For an infinitely thin dielectric needle, the polarizability a with respect to its major axis approaches infinity (a→∞). If the needle is rotated by an angle φ relative to the applied field, its polarizability aφ may be lower (e.g., aφ=a cos φ) but still very high.
While the polarization field neutralizes a portion of the internal field, it contributes to the external field. It may be appreciated that an increase of the external field also causes an increase of the capacitance and thus an increase of the susceptance (imaginary admittance) as measured at terminal 703 when the object is introduced. Since electric losses can only be generated in the material by the internal field, even an object with a substantial loss angle (e.g., δ=45°) may generate comparatively low losses if there is substantial depolarization. In other words, depolarization may prevent generating losses resulting in a reflected admittance (susceptance) that is substantially imaginary (e.g., with arg{ΔYr} in the angle range 802 of
Returning to Equation (339), the function f(εr) may be considered as an effective complex relative permittivity of the object denoted as:
εr,eff=f(εr) (340)
For certain dielectric bodies (e.g., a sphere) producing a uniform internal field when a uniform field is applied, the effective relative permittivity εr,eff corresponds to the ratio of the applied field strength to the internal field strength. For the thin sheet (a→1), this ratio approaches the material's relative permittivity (εr,eff→εr). For the sphere, this ratio is known to be εr,eff=(εr+2)/3. However, with increasing a, |εr,eff| converges to unity (|εr,eff|→1). The thin dielectric needle (a→∞) experiences virtually no depolarization, hence the inner field strength approaches the applied field strength. Based on the equivalent circuit model 810, the admittance as presented at the signal terminal 703 of the sense electrode 702 in presence of an object (e.g., object 110, 112, 114) may be found as:
Y=jω C
∞+ω2 CM2/Yob (341)
Defining the capacitive coupling factor kC according to Equation (185) and with respect to the capacitance Cob,0 as:
k
C
2
=C
M
2/(C∞ Cob,0) (342)
and applying Equations (338) and (340)to Equation (341) yields:
Y=jω C
∞(1−kC2/εr,eff) (343)
Further, using Equation (343) and defining C as the sense electrode's 702 capacitance in absence of a foreign object (εr,eff=1), the capacitance C∞ may be expressed as:
C
∞
=C/(1−kC2) (344)
and the admittance in absence of a foreign object as:
Y0=jω C (345)
Using Equations (343), (344), and (345), the reflected admittance of the object (e.g., object 110, 112, 114, or vehicle 330) may be expressed in terms of the capacitance C, the capacitive coupling factor kC, and the effective relative permittivity εr,eff as:
ΔYr=Y−Y0=jω C kC2/(1−kC2)(1−1/εr,eff) (346)
From further theoretical considerations (not included herein), it may be found that the mutual capacitance CM and thus kC generally are also a function of the polarizability a of the object and may be of the form:
k
C
2=1/(1+1/(α(a−1))) (347)
where α is a geometry factor related to the size, position, and orientation of the object (e.g., object 110, 112, 114) relative to the sense electrode 702 and ground. It may be found in the range 0<α<1. For the thin dielectric sheet (a→1), the coupling factor approaches zero (kC→0) while for the thin dielectric needle (a→∞), it approaches unity (kC→1).
According to the discussions above, the shape (form factor) of an object (e.g., object 112) is responsible for its polarizability in a substantially uniform electric field. Exposing an object (e.g., object 112) to a substantially uniform field requires the object to be small compared to its distance to the sense electrode 702. Referring to
ΔYr=Y−Y0=jω C(εr−1) (348)
Equation (348) corresponds to the admittance increase when inserting dielectric material with εr into the parallel plate capacitor. Using the definition of the complex relative permittivity εr in Equation (335) and the loss angle δ of Equation (337), Equation (348) demonstrates that the angle arg{ΔYr} directly reflects the complementary loss angle 90°−δ of a dielectric fill material with |εr|>>1 similarly to a dielectric object (e.g., object 112) with a formfactor favorable for high polarization if subjected to a substantially uniform electric field as previously discussed. It may be argued that the field as produced by the parallel plate capacitor is also uniform. However, this is only true for the space between the capacitor's plates (in the dielectric fill material) but false if the space outside is considered too. In fact, the parallel plate capacitor shows an example of a highly non-uniform field.
Summarizing, it may be concluded that the angle arg{ΔYr} is indicative of the complementary loss angle 90°−δ of a dielectric object (e.g., object 112) if the object's formfactor is favorable for polarization (a>>|εr|) or if the object is in a strongly coupled regime. An object with low polarizability in a weakly coupled regime may reflect an angle arg{ΔYr} close to 90° regardless of its loss angle δ.
In a further aspect, it may be useful to normalize the reflected admittance ΔYr as defined by Equation (346) on the magnitude of the limit reflected admittance:
ΔYr,∞=jω C kC2/(1−kC2) (349)
as obtained for an infinite relative permittivity (|εr|→∞) as follows:
ΔYr/|ΔYr,∞|=j(1−1/εr,eff) (350)
Note that this normalization does not alter the angle of ΔYr. Inspecting Equation (352), it may be appreciated that an object (e.g., object 112) with low polarizability (e.g., 1<a<<|εr|) thus high |εr,eff|>>1 reflects a normalized admittance ΔYr/|ΔYr,∞|≈j (e.g., in the angle range 802 of
In a series of lab experiments, various living and non-living objects (e.g., objects 110, 112, and 114) were tested with respect to the reflected admittance ΔYr at a sense frequency of 3 MHz when brought into proximity of a capacitive sense element (e.g., sense electrode 702, which may correspond to the capacitive sense element 109a of
More specifically, living objects (e.g., a human body extremity of an adult, of an infant, a cat, etc.) were tested with respect to their reflected admittance when brought into proximity of the capacitive sense element inside the housing. All tested living objects reflected an admittance ΔYr close to the imaginary axis (e.g., in the angle range 802 of
TABLE 5 lists the real and imaginary part of the complex permittivity εr and the related conductivity σ of various human tissue types at 3 MHz as they may be found in scientific publications. These tissue types together may constitute a substantial portion of a human body extremity. The numbers in TABLE 5 suggest a relative permittivity generally in a range ε′r>30 and ε″r>36 for a living object (e.g., object 114).
Electromagnetic field exposure regulations may require a living object to be detected already at a distance from the capacitive sense element (e.g., sense electrode 702 of
A reflected admittance ΔYr virtually at the imaginary axis (e.g., in the angle range 802 of
Further, tests were performed with nonliving dielectric objects (e.g., object 112) e.g., a piece of plastic with ε′r<3, a polyethylene terephthalate (PET) plastic bottle filled with distilled water (σ≈0), filled with tap water (σ≈0.5 mS/m), filled with salt water (σ≈40 mS/m), potting soil, wet foliage, snow, and ice. All test objects reflected an admittance ΔYr virtually at the imaginary axis (e.g., in the angle range 802 of
~78
~78
~78
~3000
<0.15
Tests were also performed with water dripping on the plastic housing of a wireless power transfer structure (e.g., housing 328 of the wireless power transfer structure 200 with reference to
Based on the theory above and more specifically by Equation (352), a low angle arg{ΔYr} (e.g., in the range 804 of
Specific lab experiments were carried out to further investigate this phenomenon using a test set up with water contained in a plastic hose disposed proximate to a capacitive sense element (e.g., capacitive sense element 109a). The plastic hose was connected to a water reservoir allowing the water level in the hose to be accurately adjusted. More specifically, in a first experiment, the reflected admittance ΔYr was measured for a change of water level by 60 mm in a first plastic hose with a diameter of 2 mm. In a second experiment, it was measured for a change of water level by 15 mm in a second plastic hose having a diameter of 4 mm. In both experiments, changing the water level by 60 mm and 15 mm, respectively, may be considered equivalent to adding a cylindrically shaped sample of water (e.g., object 112) with a volume of 188.5 mm3 to the water already existing in the hose, however with a different shape (form factor). Being equal in volume, the shape of the two water samples may be quantified by their volume-to-surface area ratio in Millimeter units. The volume-to-surface area ratio of the 2×60 mm water sample used in the first experiments amounts to 0.49 mm vs. 0.88 mm in the second experiment. For comparison, a spherical water droplet of 4 mm diameter provides a volume-to-surface area of 0.66 mm. Further, to investigate the impact of the water's conductivity σ, the first and second experiments were performed with commercially off-the-shelf distilled water, tap water with about 0.03% of calcium and magnesium ions, and with water of different salinity using a NaCl solution. Starting with distilled water, the NaCl concentration was successively increased (doubled) in a series of measurements.
To illustrate the variation of the angle arg{ΔYr} over the range of tested ion concentrations in the 2×60 mm water sample,
Likewise,
In a further aspect, the effective electric susceptibility defined as:
χe,eff=εr,eff′−1 (351)
and the effective conductivity σeff of the 2×60 mm and 4×15 mm water samples are analyzed based on the measured reflected admittances and Equations (337), (335), and (336) assuming a negligible loss coefficient εd,r″ as previously discussed.
Finally, the diagram 880 of
The outcomes of the lab experiments as described above indicate the potential to discriminate between water (e.g., object 112) dripping from the vehicle's (e.g., vehicle 330) underbody and living objects (e.g., living object 114) based on the angle arg{ΔYr}. Therefore, in some implementations of the multi-purpose detection circuit 100 (e.g., based on the circuit 700 of
Though described above for measuring a change in an admittance or impedance, discriminating water (e.g., object 112) dripping from the vehicle's (e.g., vehicle 330) underbody may also be accomplished based on other electrical characteristics as they may be measured in some implementations of the multi-purpose detection circuit 100 and as mentioned in any of the US patent applications herein incorporated by reference. As observed in the reflected admittance ΔY water (e.g., object 112) dripping from the vehicle's (e.g., vehicle 330) underbody may also cause a change in an electrical characteristic different from a change produced by other objects (e.g., object 110 and 114).
The circuit 900 may be subdivided into a driver circuit 402, a plurality of inductive sense circuits 106, a plurality of capacitive sense circuits 108, and a measurement amplifier circuit 404 with reference to the generic block diagram of
In the example implementation shown in
Each of the plurality of inductive sense circuits 106 provides a first measurement port 936 (indicated in
The driver circuit 402 includes an input multiplexer circuit 910 to selectively (e.g., sequentially) drive each of the plurality of sense circuits 106 and 108 with the current I1. Likewise, the measurement amplifier circuit 404 includes an output multiplexer circuit 940 configured to selectively (e.g., sequentially) measure the voltage V2 in each of the plurality of sense circuits 106 and 108. More specifically, but not indicated in
The circuit 900 may be configured and operated in a mode to selectively (e.g., sequentially) measure an intra-sense circuit transimpedance Z21 e.g., between the measurement ports 936 and 937 of each of the plurality of the sense circuits 106. This intra-sense circuit transimpedance Z21 and may be defined for the sense circuit 106a as:
Z2a1a≈V2a/I1a (352)
For certain configurations of the sense circuits 106 and 108 (example given below), the two-port transimpedance Z21 substantially equals the one-port impedance Z11 as it may be measured at the first measurement port (e.g., measurement port 936) with the second measurement port (e.g., measurement port 937) open-circuited.
However, the circuit 900 may also be configured and operated in a mode to selectively (e.g., sequentially) measure an inter-sense circuit transimpedance Z21 e.g., between each of a plurality of pairs of sense circuits associated with neighboring sense elements (e.g., inductive sense element 107a and 107b) providing sufficient cross-coupling. The inter-sense circuit transimpedance Z21 as measured between the measurement port 936 of sense circuit 106a and the measurement port 937 of sense circuit 106b may be defined as:
Z2a1b≈V2b/I1a (353)
In some implementations or operations, inter-sense circuit transimpedance Z21 measurements are performed between pairs of inductive sense circuits (e.g., inductive sense circuits 106a and 106b) and between pairs of capacitive sense circuits (e.g., capacitive sense circuits 108a and 108b). For simplicity, the intra-sense circuit transimpedance Z21 herein is often referred to as the impedance Z11 and the inter-sense circuit transimpedance Z21 as the transimpedance Z21. However, in a strict sense, both Z11 and Z21 may represent a transimpedance.
In an aspect, an object (e.g., object 110) proximate to at least one of the neighboring sense elements (e.g., 107a and 107b) may change both the impedance Z11 and the transimpedance Z21. Therefore, additionally measuring Z21 may improve the detection reliability of the multipurpose detection circuit 100. Example implementations and operations of object detection circuits configured to measure both the impedance Z11 and the transimpedance Z12 are described in U.S. patent application Ser. No. 16/358,534, titled Foreign Object Detection Circuit Using Mutual Impedance Sensing, the entire contents of which are hereby incorporated by reference.
The inductive sense circuit 106a includes an inductive sense element 107a including a sense coil (e.g., sense coil 502 of
In an example implementation or configuration of the circuit 900 of
In some implementations, at least the capacitor 504 is of a type with a low temperature coefficient providing high thermal stability (e.g., a NP0-type capacitor) reducing thermal drift of an electrical characteristic (e.g., an impedance Z11) as measured at each of the plurality of inductive sense circuits 106a, 106b, . . . , 106n. In other implementations, the capacitor 504 is a temperature compensation capacitor configured to compensate at least a portion of a temperature drift of the inductive sense element 107a. In a further aspect, the inductor 506 may use a ferrite core or may be an air coil e.g., for purposes of a higher linearity.
In yet further implementations of the circuit 900 using a printed circuit board (PCB), the plurality of inductors 506 is arranged to reduce a magnetic field coupling between neighboring inductors 506, e.g., by an alternating orientation. In yet other implementations, the inductor 506 is electromagnetically shielded to reduce at least one of a magnetic field coupling between neighboring inductors 506 and a disturbance voltage induced into the inductor 506 e.g., by the magnetic field as generated by the wireless power transfer structure (e.g., wireless power transfer structure 200 with reference to
Moreover, as previously mentioned in connection with the circuit 500 of
The second capacitor 928 may be needed in some implementations to block any DC flow at the output of the driver circuit 402. In an aspect, the capacitor 928 may also help to attenuate any residual low frequency voltage component (e.g., at the WPT operating frequency) at the output of the driver circuit 402. Moreover, in some implementations, it may also be used to compensate or partially compensate for the effect of the reactance of parallel inductor 506 in the measured impedance (e.g., Z11) and hence to reduce an error in the measured angle arg{ΔZ} as previously discussed with respect to
The impedance |Z11| of the inductive sense circuit 106a at series resonance is assumed in the suitable measuring range of the measurement circuit (e.g., measurement circuit 104 with reference to
The capacitive sense circuit 108n includes a capacitive sense element 109n including a sense electrode (e.g., sense electrode 702 of
In some implementations, the transformer 726 comprises a primary winding and a galvanically isolated secondary winding, both windings wound on a common ferrite core as indicated in
In the example implementation or configuration of the circuit 900 of
In some implementations, at least the inductor 704 may be of a type with a low temperature coefficient providing higher thermal stability reducing thermal drift of the impedance Z11 as measured at each of the plurality of the capacitive sense circuits 108. In other implementations, the inductor 704 is a temperature compensation inductor configured to compensate at least a portion of a temperature drift of the capacitive sense element 109n. In further implementations (not shown herein), a supplementary temperature compensation capacitor is electrically connected in parallel to the capacitive sense element 109n (e.g., as illustrated by capacitor 715 in
In another aspect and in some implementations of the circuit 900 built on a printed circuit board, the plurality of inductors 724 is arranged to reduce a magnetic field coupling between neighboring inductors 724 and a neighboring inductor 506 of an inductive sense circuit (e.g., inductive sense circuit 106a), e.g., by an alternating orientation. In yet other implementations, at least one of the inductor 724 and the transformer 726 is electromagnetically shielded to reduce at least one of a magnetic field coupling between neighboring inductors 724, 506, and a disturbance voltage induced e.g., by the magnetic field as generated by the wireless power transfer structure (e.g., wireless power transfer structure 200 with reference to
In further implementations, at least a portion of the required inductance Ls is realized by a leakage inductance of the transformer 726 as previously mentioned with reference to the circuit 720 of
As also mentioned in connection with the circuit 720 of
In a further aspect, the transformer 726 may be employed to transform the impedance Z11 of the sense circuit 108n as presented at the series resonance into the suitable measuring range of the measurement circuit (e.g., measurement circuit 104 of
The driver circuit 402 includes a driver amplifier circuit 902, an input multiplexer circuit 910 illustrated in
The driver amplifier circuit 902 together with series resistors (e.g., series resistor 914) mimic a current source characteristic at each of the plurality of outputs of the driver circuit 402. It may be appreciated that the series resistor (e.g., series resistor 914) with a resistance (e.g., Rser1) substantially larger (e.g., 10 times larger) than the impedance magnitude |Z11| at series resonance e.g., of the sense circuit 106 may transform the voltage source output of the driver amplifier circuit 902 into a current source output meeting the requirements of a current source 512 as previously defined in connection with
In an alternative configuration, the current source characteristic is realized using a resistor (e.g., series resistor 914) with a lower resistance (e.g., Rser1) instead using the DC block capacitor (e.g., capacitor 928) with a higher reactance, together providing an impedance substantially larger (e.g., 10 times larger) than the impedance as presented at the primary port of the transformer 726 at series resonance. In another implementation variant (not shown herein), the resistor (e.g., series resistor 914) is omitted entirely and the high impedance is realized by the DC block capacitor (e.g., capacitor 928). In a further implementation variant (not shown herein), the high series impedance as required to mimic a current source characteristic is realized at least in part by using at least one of an inductor (not shown in
In yet another implementation variant, the current source characteristic is realized using a driver amplifier circuit 902 configured as a regulated current source. An example current source circuit using an operational amplifier is illustrated in
In another implementation variant (not shown herein), the driver amplifier circuit 902 additionally includes an output transformer e.g., for purposes of transforming an output voltage. As opposed to lowering the resistance (e.g., Rser1), the use of an output transformer may allow the drive current level I1 and thus the sense element current levels IL and IC to be increased without compromising the current source characteristic of the driver circuit 402
The input multiplexer circuit 910 includes a plurality of switches 911a, 911b, . . . , 911n and is configured to selectively connect each of the plurality of inductive sense circuits 106 and each of the plurality of capacitive sense circuits 108 via the respective series resistor 914 and 915 to the driver circuit 402 to selectively (e.g., sequentially) drive each of the plurality of inductive sense circuits 106 and each of the plurality of capacitive sense circuits 108 with the current I1 at the sense frequency. Therefore, each of the plurality of switches 911a, 911b, . . . , 911n is electrically connected to the driver amplifier circuit's 902 output that is also referred to as the common input node. The input multiplexer circuit 910 is further configured to receive an input MUX control signal from a control circuit (e.g., from the control and evaluation circuit 102 of
Each of the plurality of switches 911a, 911b, . . . , 911n may be one of a semiconductor analog switch (e.g., a single field effect transistor (FET) switch, a complementary FET switch composed of a p-channel and a n-channel type FET), a micro-mechanical systems (MEMS) switch or any other type of switch providing a sufficiently low series resistance, when the switch is closed (closed-state) and a sufficiently high current signal attenuation, when the switch is open (open-state). An example implementation of an analog switch (e.g., switch 911a) based on a single FET is illustrated in
The switches (e.g., switch 911a) may be characterized by a closed-state series resistance, an equivalent open-state series capacitance, and an equivalent parallel capacitance at each side of the switch. It may be appreciated that the closed-state resistance of the switch (e.g., switch 911a) is non-critical as it merges with the resistance (e.g., Rser1) of the series resistor (e.g., series resistor 914). It may also be appreciated that the total capacitive load produced by the plurality of parallel capacitances at the common input node may be non-critical since it is in parallel to the voltage source output of the driver amplifier circuit 902.
The switch (e.g., switch 911a) of an example input multiplexer circuit 910 may use complementary FET switches with a closed-state resistance of 5Ω, an equivalent open-state series capacitance of 3 pF (corresponding to a series reactance of 17.7 kΩ at a sense frequency of 3 MHz), and an equivalent parallel capacitance of 12 pF on each side of the switch.
In a further aspect, the closed-state resistance of a semiconductor analog switch (e.g., switch 911a) may be subject of a temperature drift that may impact the temperature stability of the driver circuit 402. Therefore, in some implementations, the impact of the input multiplexer circuit 910 switch (e.g., switch 911a) is reduced by using a resistor (e.g., series resistor 914) whose resistance (e.g., Rser1) is substantially larger than the closed-state resistance of the switch. Therefore, in some implementations, the series resistances Rser1 and Rser3 may also represent a trade-off between a temperature stability and a SNR as discussed above.
In an implementation variant of the circuit 900 (not shown herein), the order of the series resistor (e.g., series resistor 914) and the switch (e.g., switch 911a) is reversed, meaning that the plurality of series resistors 914 and 915 are electrically connected to the output of the driver amplifier circuit 902 (common input node) and the input multiplexer circuit 910 is inserted between the plurality of series resistors 914 and 915 and the plurality of sense circuits 106 and 108. Reversing the order may be advantageous for the design of the switch (e.g., switch 911a) as the voltage V2 across the sense circuit may be substantially lower than the voltage at the output of the driver amplifier circuit 902.
As discussed above, a low frequency disturbance voltage (e.g., at WPT frequency) may be present at the driver circuit 402 output and at the measurement amplifier circuit 404 input e.g., due to the voltage induced into the sense element (e.g., sense element 107a) by the WPT magnetic field. If a switch (e.g., switch 911b) is in open-state, a substantial low frequency voltage may also be present across the switch. This may be particularly true during active WPT operation. If too large, the open-switch voltage may affect any of the switch' open-state electrical characteristic or cause damage to the switch. In some implementations, the open-switch voltage is limited by configuring the inductive and capacitive sense circuits (e.g., sense circuit 106a and sense circuit 108) accordingly, trading-off the open-switch voltage vs. other impacts.
The measurement amplifier circuit 404 is configured to operate as the analog front-end part of a voltage measurement circuit (e.g., voltage measurement circuit 510 as described in connection with
The measurement amplifier circuit 404 includes a transimpedance amplifier circuit 952, an output multiplexer circuit 940 illustrated in
The example transimpedance amplifier circuit 952 as illustrated in
The transimpedance amplifier circuit 952 together with the series resistor (e.g., resistor 944) mimic a voltage measurement circuit characteristic at each of the plurality of inputs of the measurement amplifier circuit 404. It may be appreciated that the series resistor (e.g., resistor 944) with a resistance (e.g., Rser2) substantially larger (e.g., 10 times larger) than the impedance magnitude |Z11| at series resonance e.g., of the sense circuit 106a transforms the virtually zero impedance input of the transimpedance amplifier circuit 952 into a high impedance input satisfying the requirements of a voltage measurement circuit 510 as previously specified in connection with
In an aspect, increasing the resistance (e.g., Rser2) of the series resistor (e.g., resistor 944) may improve a voltage measurement circuit characteristic but reduce the input current level Iin. The input current level Iin may impact a SNR as previously defined with reference to
In an example implementation variant (not shown herein), an impact of the output multiplexer circuit 940 switch (e.g., switch 941a) (e.g., a temperature drift) is reduced by omitting the output multiplexer circuit 940, instead using a plurality (bank) of measurement amplifiers (not shown herein), whose inputs are electrically connected to the respective measurement port (e.g., measurement port 937) of the respective sense circuit (e.g., sense circuit 106a) and whose outputs are electrically connected to a common output (Vout). Each measurement amplifier circuit 404 is configured to provide a high input impedance and includes an operational amplifier (e.g., amplifier 954) providing a mute control input to apply a logic signal indicative for the output MUX control signal as indicted in
In an example configuration of the circuit 900, the voltage measurement circuit characteristic is realized using a resistor (e.g., resistor 944) with a lower resistance (e.g., Rser2) instead using the DC block capacitor (e.g., capacitor 929 of sense circuit 106a) with a higher reactance, together providing a series impedance substantially larger (e.g., 10 times larger) than the impedance of the series circuit e.g., of the sense element 107a and the capacitor 504 at series resonance. In an implementation variant, the series resistor (e.g., resistor 944) is omitted and the voltage measurement circuit characteristic is realized using the DC block capacitor (e.g., capacitor 929) configured to provide a high enough series impedance. In another implementation variant (not shown herein), the high series impedance as required to mimic a current source characteristic is realized at least in part by using at least one of an inductor (not shown in
The feedback capacitor 958 provides the transimpedance amplifier circuit 952 with a first order low pass filter characteristic to attenuate disturbance signal components at frequencies higher than the sense frequency (e.g., high order WPT harmonics). In some implementations, the capacitance Cf may be a trade-off between a reduction in gain at the sense frequency and an attenuation of the high frequency signal components. The feedback capacitor 958 may reduce a risk for signal clipping or non-linear distortion in the amplifier 954 or in the further processing (e.g., the signal processing circuit 408 with reference to
In an implementation variant (not shown herein), the transimpedance amplifier circuit 952 is further enhanced by a supplementary feedback inductor electrically connected in parallel to the feedback capacitor 958 providing the transimpedance amplifier circuit 952 with a bandpass characteristic tuned to the sense frequency. This inductor may help to further suppress low frequency disturbance signal components (e.g., the WPT fundamental and low frequency harmonics thereof). In such an implementation, the feedback capacitor 958 may be a temperature compensation capacitor to compensate for at least a portion of the feedback inductor's temperature drift.
In another implementation variant (not shown in
The output multiplexer circuit 940 including the plurality of switches 941a, 941b, . . . , 941n is configured to selectively connect each of the plurality of inductive sense circuits 106 and each of the plurality of capacitive sense circuits 108 via the respective series resistors 944 and 945 to the transimpedance amplifier circuit 952 to selectively (e.g., sequentially) buffer and amplify the voltage V2 at each of the plurality of sense circuits 106 and 108 in response to a current I0 at the sense frequency. Therefore, each of the plurality of switches 941a, 941b, . . . , 941n is electrically connected to the common output node that is electrically connected to the negative input of the transimpedance amplifier circuit 952. The output multiplexer circuit 940 is further configured to receive an output MUX control signal from a control circuit (e.g., from the control and evaluation circuit 102 of
As for the input multiplexer circuit 910, each of the plurality of switches 941a, 941b, . . . , 1041n may be one of a semiconductor analog switch (e.g., a single FET switch, a complementary FET switch composed of a p-channel and a n-channel type FET), a micro-mechanical systems (MEMS) switch or any other type of switch providing a sufficiently high current signal attenuation in the open-state. An example implementation of an analog switch (e.g., switch 941a) based on a single FET is illustrated in
As previously discussed in connection with the input multiplexer circuit 910, each of the plurality of switches 941a, 941b, . . . , 941n may exhibit a closed-state series resistance, an equivalent open-state series capacitance, and an equivalent parallel capacitance on each side of the switch. It may be appreciated that the closed-state series resistance (e.g., of switch 941a) may be non-critical for the functioning of the measurement amplifier circuit 404 since it merges into an overall series resistance Rser2 or Rser4. It may also be appreciated that the total capacitive load produced by the plurality of parallel capacitances at the common output node of the output multiplexer circuit 940 may be non-critical since it is in parallel to the virtually zero-impedance input of the transimpedance amplifier circuit 952.
An example output multiplexer circuit 940 for a sense frequency of 3 MHz may use complementary FET switches with the same characteristics as those used for the example input multiplexer circuit 910 as specified above.
As previously discussed with reference to the input multiplexer circuit 910, a low frequency voltage (e.g., at WPT frequency) may also be present across the switch (e.g., switch 941a) when WPT is active and the switch in open-state. If too large, this open-switch voltage may affect any of the switch's open-state electrical characteristics or cause damage to the switch. In some implementations, the open-switch voltage is limited as previously discussed for the input multiplexer circuit 910.
In an implementation variant (not shown herein), the order of the series resistor (e.g., resistor 944) and the switch (e.g., switch 941a) is reversed, meaning that the series resistor (e.g., resistor 944) is electrically connected to the input of the transimpedance amplifier circuit 952 (common output node) and the output multiplexer circuit 940 is inserted between the plurality of sense circuits 106 and 108 and the plurality of resistors 944 and 945.
A further implementation variant of the circuit 900 (not shown herein) omits the output multiplexer circuit 940 e.g., for reasons of complexity and cost reduction. In theory, the input multiplexer circuit 910 may be enough to selectively (e.g., sequentially) drive a sense circuit (e.g., sense circuit 106a) with the current I1 and to selectively measure the voltage V2 at its measurement port 937 in response to the current I1. Because the measurement amplifier circuit 404 is configured as a voltage summation amplifier, its output voltage Vout is indicative of the sum of the voltages at each of the plurality of measurement ports 937. Since the voltages V2 at the inactive sense circuits (not driven by the current are ideally zero, the output voltage Vout is indicative of the voltage V2 of the active sense circuit (e.g., sense circuit 106a) driven by the current I1. However, in practice, disturbance voltages e.g., inductively or capacitively coupled into the sense element (e.g., inductive sense element 107b) may also sum up resulting in a lower SNR as compared to a circuit 900 using the output multiplexer circuit 940. Moreover, a circuit 900 omitting the output multiplexer circuit 940 may not support supplementary inter-sense circuit transimpedance Z12 measurements as described above. The inter-sense circuit transimpedance Z12 between the measurement port 936 of sense circuit 106a and the measurement port 937 of sense circuit 106b may be measured by setting the switches 911a and 941b to the close-state and the other switches of the plurality of switches 911a, 911b, . . . , 911n to the open-state.
A numerical specification and some resulting performance figures of an example circuit 900 with respect to the inductive sense circuits 106 configured for a nominal sense frequency fs=3 MHz are given in TABLE 7. A WPT operating frequency fWPT=85 kHz is assumed. TABLE 7 also includes a system Q-factor defined as the ratio
Qs:ys≈|ΔVout′|/|ΔZ′r (354)
where |ΔVout′| denotes the magnitude fractional change of the measurement amplifier circuit's 404 output voltage Vout caused by an object (e.g., object 110) and |ΔZ′r| the magnitude normalized reflected impedance of the object as defined by Equations (6) an (211) for an inductive sense element (e.g., inductive sense element 107a) and a capacitive sense element (e.g., capacitive sense element 109n), respectively. Further, TABLE 7 includes a quality of the measurement circuit 104 defined as the ratio:
Qmc≈|ΔVout′|/|ΔI1′| (355)
where |ΔVout′| denotes the magnitude fractional change of the measurement amplifier circuit's 404 output voltage Vout caused by an object (e.g., object 110) and |ΔI1′| the magnitude fractional change of the driver circuit's 402 output current I1 caused by that object. Moreover, TABLE 7 includes a degradation of the fractional change in the measurement amplifier circuit's 404 output voltage Vout defined as:
γVout≈1−(|ΔVout′|/|ΔZ′|) (356)
where |ΔZ′| denotes the magnitude fractional change of the intra-sense circuit transimpedance Z2a1a as defined by Equation (352) for the sense circuit 106a by example. This degradation may be the result of imperfections in the driver circuit 402 and the measurement amplifier circuit 404 and in other circuit elements as needed in a practical implementation.
A numerical specification of an example circuit 900 with respect to the capacitive sense circuits 108 configured for a nominal sense frequency fs=3 MHz are given in TABLE 8. A WPT operating frequency fWPT=85 kHz is assumed. TABLE 8 also includes the system Q-factor Qsys as defined above by Equation (354), the quality of the measurement circuit 104 as defined by Equation (355), and the degradation of the fractional change in the measurement amplifier circuit's 404 output voltage Vout as defined by Equation (356).
The example measurement amplifier circuit 404 of
Active filters providing a bandpass characteristic may be implemented in a number of ways which may be well known to those skilled in the art. However, not many of these implementations may satisfy the requirements of a voltage measurement circuit (e.g., voltage measurement circuit 510).
In an aspect, the sense element 107a having inductance L in combination with the parallel capacitor 507 having capacitance Cp1 form a 2nd order low pass filter to attenuate a disturbance component in the voltage V2 at a frequency substantially higher than the nominal sense frequency. In some implementations or operations, this disturbance component emanates from a voltage inductively or capacitively coupled into the sense element 107a by the magnetic or electric field, respectively, as generated during wireless power transfer. In other implementations or operations, it emanates from a voltage inductively or capacitively coupled into the sense element 107a by the magnetic or electric field, respectively, as generated by another system (not shown herein) e.g., a transmitter of a communication system operating at a frequency substantially higher than the nominal sense frequency and that is cointegrated into the primary wireless power transfer structure 200 with reference to
In an aspect, the series inductor 724 having inductance Ls in combination with the parallel capacitor 727 having capacitance Cp2 form a 2nd order low pass filter to attenuate a disturbance component in the voltage V2 at a frequency substantially higher than the nominal sense frequency. In some implementations or operations, this disturbance component emanates from a voltage coupled into the sense electrode (e.g., capacitive sense element 109n) e.g., by the electric field as generated during wireless power transfer. In other implementations or operations, it emanates from a voltage coupled into the sense electrode (e.g., capacitive sense element 109n) e.g., by the electric field as generated by another system (not shown herein) but an example is provided above.
In some implementations, each of the parallel capacitors 507 and 727 as part of a 2nd order low pass filter reduces dynamic range requirements of the measurement amplifier circuit 404 and may also protect the driver circuit 402 including the input multiplexer circuit 910 and the measurement amplifier circuit 404 including the output multiplexer circuit 940 from being overloaded. Stated more generally, it may reduce non-linear distortion effects (e.g., signal clipping) in the circuit 900 of
As with the circuit 900 of
In the example implementation shown in
Each of the plurality of inductive sense circuits 106 provides a first measurement port 936 (indicated in
The driver circuit 402 includes an input multiplexer circuit 910 to selectively (e.g., sequentially) apply the voltage V1 to each of the plurality of sense circuits 106 and 108. Likewise, the measurement amplifier circuit 404 includes an output multiplexer circuit 940 configured to selectively (e.g., sequentially) measure the current I2 in each of the plurality of sense circuits 106 and 108. More specifically, but not indicated in
The circuit 1000 may be configured and operated in a mode to selectively (e.g., sequentially) measure the intra-sense circuit transimpedance Z21 e.g., between the measurement ports 936 and 937 of each of the plurality of the sense circuits 106 defined as:
Z2a1a≈V1a/I2a (357)
As mentioned above and further discussed below, the two-port transimpedance Z21 substantially equals the one-port impedance Z11 as it may be measured at the first measurement port (e.g., measurement port 936) with the second measurement port (e.g., measurement port 937) short-circuited.
However, as opposed to the circuit 900 of
The inductive sense circuit 106a includes an inductive sense element 107a including a sense coil (e.g., sense coil 502 of
In an example implementation of the circuit 1000 of
In another aspect, the capacitors 544 and 546 may be configured to provide an impedance magnitude |Z11| at the sense frequency in a suitable range for the measurement circuit (e.g., measurement circuit 104 of
In some implementations, the capacitors 544 and 546 are of a type with a low temperature coefficient providing high thermal stability (e.g., a NP0-type capacitor) reducing thermal drift of an electrical characteristic (e.g., an impedance) as measured at each of the plurality of inductive sense circuits 106a, 106b, . . . , 106n.
Moreover, as previously discussed in connection with
In another aspect, the inductive sense circuits 106 may not need any supplementary capacitors (e.g., capacitors 928 and 929) for purposes of DC blocking as previously discussed with reference to the circuit 900 of
The capacitive sense circuit 108n includes a capacitive sense element 109n including a sense electrode (e.g., sense electrode 702 of
In an implementation variant using a capacitive sense element 109n including a double-ended sense electrode (not shown herein), the sense circuits 108 may not need a transformer (e.g., transformer 726) for purposes of ground-decoupling. In another implementation variant using a double-ended sense electrode, each of the plurality of capacitive sense circuits 108 is based on the sense circuit 781 as illustrated in
In an example implementation of the circuit 1000 of
In another aspect, the transformer 726 may be configured to provide an impedance |Z11| at the sense frequency in a suitable range for the measurement circuit (e.g., measurement circuit 104 of
The secondary-referred main inductance Lm of the transformer 726 together with the capacitance C of the capacitive sense element 109n form a 2nd order high pass filter to attenuate a low frequency disturbance component in the current I2 as previously discussed with reference to
The driver circuit 402 includes a driver amplifier circuit 902, an input multiplexer circuit 910 illustrated in
Further, the driver circuit 402 is configured to operate as a voltage source (e.g., voltage source 552 as described in connection with
The input multiplexer circuit 910 includes a plurality of switches 911a, 911b, . . . , 911n and is configured to selectively connect each of the plurality of inductive sense circuits 106 and each of the plurality of capacitive sense circuits 108 to the driver circuit 402 to selectively (e.g., sequentially) apply the voltage V1 at the sense frequency to each of the plurality of inductive sense circuits 106 and to each of the plurality of capacitive sense circuits 108. Therefore, each of the plurality of switches 911a, 911b, . . . , 911n is electrically connected to the driver amplifier circuit's 902 output that is also referred to as the common input node. The input multiplexer circuit 910 is further configured to receive an input MUX control signal from a control circuit (e.g., from the control and evaluation circuit 102 of
Each of the plurality of switches 911a, 911b, . . . , 911n may be one of a type as previously mentioned with reference to the circuit 900 of
The switch (e.g., switch 911a) of an example input multiplexer circuit 910 may use complementary FET switches with a closed-state resistance of 5Ω, an equivalent open-state series capacitance of 3 pF (corresponding to a series reactance of 17.7 kΩ at a sense frequency of 3 MHz), and an equivalent parallel capacitance of 12 pF on each side of the switch.
While in the circuit 900 series resistors 914 and 915 are used to transform the amplifier's 904 voltage source output into a current source output, parallel resistors 1014 and 1015) are employed in the circuit 1000 to limit the open-switch voltage (e.g., at WPT frequency) across the input multiplexer circuit 910 switch (e.g., switch 911a). It may be appreciated that decreasing the resistance (e.g., Rpar1) of the parallel resistor (e.g., parallel resistor 1014) may reduce the open-switch voltage. However, it will also increase an output current of the driver amplifier circuit 902 as more current will be diverted to ground. This may increase a voltage drop across the switch (e.g., switch 911a) and hence increasing an impact of the switch' as previously discussed with reference to
Therefore, in an implementation variant (not shown herein), the input multiplexer circuit 910 is incorporated into the driver amplifier circuit 902 also employing an additional (third) multiplexer circuit in a feedback path. This implementation variant may provide a regulated (stable) voltage source characteristic at each of the plurality of outputs of the driver circuit 402 substantially eliminating the effect of the switch' temperature dependent closed-state resistance. An example circuit of the driver circuit 402, which is voltage regulated, is disclosed in U.S. patent application Ser. No. 16/226,156, titled Foreign Object Detection Circuit Using Current Measurement.
The measurement amplifier circuit 404 is configured to operate as the analog front-end part of a current measurement circuit (e.g., current measurement circuit 550 as described in connection with
As with the parallel resistor 1014, decreasing the resistance (e.g., Rpar2) of the parallel resistor 1044 will reduce the open-switch voltage (e.g., at WPT frequency) across the output multiplexer circuit 940 switch (e.g., switch 941a). As opposed to the parallel resistor 1014, decreasing the resistance of the parallel resistor 1044 may be less critical since the voltage across the parallel resistor 1044 is low when the switch (e.g., switch 941a) is in closed-state. In an example implementation, the resistances Rpar2 and Rpar4 substantially match the respective impedance |Z11| at series resonance and are substantially higher than the switch' closed-state resistance.
The example transimpedance amplifier circuit 952 as illustrated in
In other implementation variants (not shown herein), the filtering of the transimpedance amplifier circuit 952 is further enhanced in similar ways as previously described with reference to
In a further implementation variant (not shown herein), the transimpedance amplifier circuit 952 additionally includes an input transformer e.g., for purposes of transforming an input current Iin. The transformer may be used e.g., to reduce the current Iin to a level not exceeding an input current constraint of the amplifier 954 and hence allowing the drive current I1 and eventually the sense currents IL and IC in the respective sense elements 107a, 107b, . . . , 107n and 109a, 109b, . . . , 109n to be increased. An example transimpedance amplifier circuit 952 using an input transformer is disclosed in U.S. patent application Ser. No. 16/226,156, titled Foreign Object Detection Circuit Using Current Measurement.
The output multiplexer circuit 940 including the plurality of switches 941a, 941b, . . . , 941n is configured to selectively connect each of the plurality of inductive sense circuits 106 and each of the plurality of capacitive sense circuits 108 to the transimpedance amplifier circuit 952 to selectively (e.g., sequentially) buffer and convert the current I2 at each of the plurality of sense circuits 106 and 108 in response to the voltage V1 at the sense frequency. Therefore, each of the plurality of switches 941a, 941b, . . . , 941n is electrically connected to the common output node that is electrically connected to the negative input of the transimpedance amplifier circuit 952. The output multiplexer circuit 940 is further configured to receive an output MUX control signal from a control circuit (e.g., from the control and evaluation circuit 102 of
As with the input multiplexer circuit 910, each of the plurality of switches 941a, 941b, . . . , 1041n may be one of a type of switch as previously specified with reference to the output multiplexer circuit 940 of the circuit 900 of
An example output multiplexer circuit 940 for a sense frequency of 3 MHz may use complementary FET switches with the same characteristics as those used for the example input multiplexer circuit 910 as specified above.
As mentioned above, the circuit 1000 as illustrated in
A further implementation variant of the circuit 1000 (not shown herein) omits the output multiplexer circuit 940 e.g., for reasons of complexity and cost reduction. In theory, the input multiplexer circuit 910 may be enough to selectively (e.g., sequentially) apply the voltage V1 to a sense circuit (e.g., sense circuit 106a) and to selectively measure the current I2 at its measurement port 937 in response to the voltage V1. Because the measurement amplifier circuit 404 is configured as a current summation amplifier, its output voltage Vout is indicative of the sum of the currents at each of the plurality of measurement ports 937. As the currents I2 at the inactive sense circuits (e.g., sense circuit 106b) (where no voltage V1 is applied) is ideally zero, the output voltage Vout is indicative of the current I2 of the active sense circuit (e.g., sense circuit 106a) where the voltage V1 is applied. However, in practice, disturbance currents e.g., capacitively coupling into the sense element (e.g., inductive sense element 107b) when WPT is active may also sum up resulting in a lower SNR as compared to a circuit 1000 using the output multiplexer circuit 940.
A specification and some resulting performance figures of an example circuit 1000 with respect to the inductive sense circuits 106 configured for a nominal sense frequency fs=3 MHz are given in TABLE 9. A WPT operating frequency fWPT=85 kHz is assumed. TABLE 9 also includes the system Q-factor Qsys as defined above by Equation (354), the quality of the measurement circuit 104 defined as the ration:
Qmc≈|ΔVout′|/|ΔV′1| (358)
where |ΔVout′| denotes the magnitude fractional change of the measurement amplifier circuit's 404 output voltage Vout caused by an object (e.g., object 110) and ΔV1′ the fractional change of the driver circuit's 402 output voltage V1 caused by that object. Further, it includes the degradation of the fractional change in the measurement amplifier circuit's 404 output voltage Vout as defined by Equation (356).
A specification and some resulting performance figures of an example circuit 1000 with respect to the capacitive sense circuits 108 configured for a nominal sense frequency fs=3 MHz are given in TABLE 10. A WPT operating frequency fWPT=85 kHz is assumed. TABLE 10 also includes the system Q-factor Qsys as defined above by Equation (354), the quality of the measurement circuit 104 as defined by Equation (358), and the degradation of the fractional change in the measurement amplifier circuit's 404 output voltage Vout as defined by Equation (356).
Comparing Tables 9 and 10 with Tables 11 and 12, respectively, shows the circuit 1000 more potential if a higher sense current level IL and IC and higher power efficiency was targeted. While an amplifier 904 output voltage constraint may limit the sense currents IL and IC in the circuit 900, an amplifier 954 input current constraint may limit these currents in the circuit 1000.
A further aspect to be considered when comparing the circuits 900 and 1000 are cross-coupling effects between neighboring inductive sense elements (e.g., inductive sense elements 107a and 107b of the array 107). Cross-coupling may degrade the Q-factor of a sense circuit (e.g., sense circuit 106a or 108n) due to energy absorption and may also distort its impedance function Z11(ω) eventually compromising performance and the impedance angle measurement accuracy of the multipurpose detection circuit 100. This may be particularly true in a circuit (e.g., circuit 1000) using a plurality of sense circuits 106 or 108 which, when inactive (e.g., deselected by the input and output multiplexer circuits 910 and 940, respectively), exhibit a parasitic parallel resonance as given by Equation (117) close to the sense frequency. For the example implementation of the circuit 900 specified in TABLE 7, the parasitic parallel resonance may be at 1.3 MHz, while for the example implementation of the circuit 1000 specified in TABLE 9, it may occur close to the sense frequency at 3.29 MHz. Therefore, the impact of cross-coupling in the circuit 1000 may be more significant than in the circuit 900.
In an example implementation of the circuit 1000 using an array of sense coils (e.g., array 107), a cross-coupling effect between inductive sense circuits (e.g., sense circuit 106a and 106b) is reduced by configuring the inductive sense circuits with a lower capacitance ratio nC=Cp/Cs resulting in a lower impedance |Z11| e.g., by trading-off the impact of cross-coupling and thermal drift of the input and output multiplexer circuits 910 and 940, respectively, as previously discussed with reference to
In another example implementation of the circuit 1000 using an array of sense coils (e.g., array 107), a cross-coupling effect between inductive sense circuits (e.g., sense circuit 106a and 106b) is reduced by configuring (tuning) the inductive sense circuits associated with neighboring sense elements (e.g., sense element 107a and 107b) to a different resonant frequency e.g., following a frequency reuse scheme. However, this approach may cause a conflict in a multipurpose detection circuit 100 used to detect a passive beacon transponder (e.g., passive beacon transponder 313 of
In a further example implementation using a planar array of sense coils (e.g., array 107), a gap is introduced between adjacent sense coils (e.g., between elements coil 107a and 107b) to reduce a cross-coupling effect.
In a further aspect of an inductive sense coil array (e.g., array 107), loss effects in the lead lines are considered. In certain implementations, the sense coil (e.g., sense element 107a) may be connected to the capacitor 544 of the associated sense circuit (e.g., sense circuit 106a) via a long lead line. This may apply to an implementation of the circuit 1000 where the array of sense coils (e.g., array 107) is carried on a separate printed circuit board (PCB) excluding any other components of the circuit 1000. A long lead line may cause substantial electrical losses degrading the Q-factor of a sense circuit (e.g., sense circuit 106a) and hence the performance of the multipurpose detection circuit 100.
In an example implementation of the circuit 1000 of
In an example implementation based on the circuit 900 of
The circuit 1100 may be subdivided into an analog front-end part of the measurement circuit 104 and a plurality of inductive and capacitive sense circuits 106 and 108 as previously described with reference to the generic block diagram of
In the example implementation shown in
Each of the plurality of inductive sense circuits 106 provides a measurement port 936 (indicated in
As opposed to the circuits 900 and 1000, the measurement circuit 104 of the circuit 1100 includes a single input multiplexer circuit 910 to selectively (e.g., sequentially) apply the voltage V1 to each of the plurality of sense circuits 106 and 108 and to selectively (e.g., sequentially) measure a current I1 in response to the applied voltage V1. More specifically, but not indicated in
The circuit 1100 may be configured and operated in a mode to selectively (e.g., sequentially) measure the impedance Z11 e.g., at the measurement port 936 of each of the plurality of the sense circuits 106 defined as:
Z11≈V1a/I1a (359)
However, as opposed to the circuit 900 of
The inductive sense circuit 106a includes an inductive sense element 107a including a sense coil (e.g., sense coil 502 of
In an example implementation of the circuit 1100 of
In another aspect, the capacitors 544 and 546 may be configured to provide an impedance magnitude |Z11| at the sense frequency in a suitable range for the measurement circuit (e.g., measurement circuit 104 of
In some implementations, the capacitors 544 and 546 are of a type with a low temperature coefficient providing high thermal stability (e.g., a NP0-type capacitor) reducing thermal drift of an electrical characteristic (e.g., an impedance) as measured at each of the plurality of inductive sense circuits 106a, 106b, . . . , 106n.
Moreover, as previously discussed in connection with
In another aspect, the inductive sense circuits 106 may not need any supplementary capacitor (e.g., capacitor 928) for purposes of DC blocking as previously discussed with reference to the circuit 900 of
The capacitive sense circuit 108n includes a capacitive sense element 109n including a sense electrode (e.g., sense electrode 702 of
In an example implementation of the circuit 1100 of
In another aspect, the transformer 726 may be configured to provide an impedance magnitude |Z11| at the sense frequency in a suitable range for the measurement circuit (e.g., measurement circuit 104 of
The secondary-referred main inductance Lm of the transformer 726 together with the capacitance C of the capacitive sense element 109n form a 2nd order high pass filter to attenuate a low frequency disturbance component in the current I1 as previously discussed with reference to
The measurement circuit 104 includes a driver amplifier circuit 902, a transimpedance amplifier circuit (e.g., amplifier 904), a transformer 1102, an input multiplexer circuit 910 illustrated in
Further, the measurement circuit 104 is configured to operate as a voltage source (e.g., voltage source 552 as described in connection with
The measurement circuit 104 is also configured to operate as the analog front-end part of a current measurement circuit (e.g., current measurement circuit 550 as described in connection with
In other implementations, the filtering of the transimpedance amplifier circuit 952 is further enhanced by a supplementary feedback inductor (not shown in
The transformer 1102 includes a primary winding and a galvanically insulated secondary winding wound on a common core as indicated in
In some alternative implementations of the circuit 900 of
The input multiplexer circuit 910 includes a plurality of switches 911a, 911b, . . . , 911n and is configured to selectively connect each of the plurality of inductive sense circuits 106 and each of the plurality of capacitive sense circuits 108 to the driver amplifier circuit 902 and the transimpedance amplifier circuit 952 to selectively (e.g., sequentially) apply the voltage V1 at the sense frequency to each of the plurality of sense circuits 106 and 108 and to selectively (e.g., sequentially) measure the current I1 in response to the applied voltage V1. The input multiplexer circuit 910 is further configured to receive a MUX control signal from a control circuit (e.g., from the control and evaluation circuit 102 of
Each of the plurality of switches 911a, 911b, . . . , 911n may be one of a type as previously mentioned with reference to the circuit 900 of
As in the circuit 1000 of
A specification and some resulting performance figures of an example circuit 1100 with respect to the inductive sense circuits 106 configured for a nominal sense frequency fs=3 MHz are given in TABLE 11. A WPT operating frequency fWPT=85 kHz is assumed. TABLE 10 also includes the system Q-factor Qsys as defined above by Equation (354), the quality of the measurement circuit 104 as defined by Equation (358), and the degradation of the fractional change in the measurement amplifier circuit's 404 output voltage Vout as defined by Equation (356).
A specification and some resulting performance figures of an example circuit 1100 with respect to the capacitive sense circuits 108 configured for a nominal sense frequency fs=3 MHz are given in TABLE 12. A WPT operating frequency fWPT=85 kHz is assumed. TABLE 10 also includes the system Q-factor Qsys as defined above by Equation (354), the quality of the measurement circuit 104 as defined by Equation (358), and the degradation of the fractional change in the measurement amplifier circuit's 404 output voltage Vout as defined by Equation (356).
The numerical values as shown in TABLEs 11 and 12 for the circuit 1100 are similar to those obtained for the circuit 1000 listed in TABLEs 9 and 10, respectively. The degradation of the fractional change in the circuit 1100 is larger. However, this drawback may be acceptable considering the potential for circuit complexity reduction in the circuit 1100.
More specifically,
Integrated into a wireless power transfer structure (e.g., wireless power transfer structure 200), electrodes of capacitive sense elements 109a, 109b, . . . , 109n may experience substantial eddy current heating due to the strong alternating magnetic fields as generated during wireless power transfer. Therefore, in a further aspect of mitigating eddy current heating, electrodes are designed to increase a surface impedance.
In another implementation, the element 1202 is made of a weakly conductive material providing a sufficiently high surface impedance. The material may represent a trade-off between eddy current heating and an equivalent resistance (e.g., equivalent resistance R as indicated in
In further implementations, other suitable structures or materials are used to increase the surface impedance of the element 1202 trading-off eddy current heating and an equivalent resistance R of the capacitive sense element (e.g., capacitive sense element 109a) at the sense frequency as previously discussed with reference to
In some implementations, the element 1202 is made as a printed circuit board (PCB). In other implementations, the plurality of elements 1202 is a flex print that also includes the inter-element connections as mentioned above with reference to
In further implementations with reference to
The image (snapshot) sequence of
The sequence of images 1500 to 1580 of
Pattern 1502 (all pixels dark gray) refers to the absence of the vehicle 330 or to a vehicle 330 position as shown by image 1500 where the vehicle's 330 impact on the detection outputs of the multi-purpose detection circuit 100 is below the detection threshold.
Pattern 1512 refers to a vehicle 330 position as shown by image 1510 where the front (leading edge) of the vehicle 330 starts to cause a minority of detection outputs to exceed the detection threshold resulting in brighter gray pixels in the 1st and 2nd column of the pattern 1512.
Pattern 1522 refers to a vehicle 330 position as shown by image 1520 where the leading edge of the vehicle 330 causes more detection outputs to exceed the detection threshold or even the saturation level resulting in white pixels in the 1st and 2nd column and brighter gray pixels in the 3rd and 4th column of the pattern 1522.
Pattern 1532 refers to a vehicle 330 position as shown by image 1530 where the leading edge of the vehicle 330 is further advanced and substantially overlapping the surface of the wireless power transfer structure 200, thus causing a majority of detection outputs to exceed the detection threshold and a higher number thereof to exceed the saturation level resulting in white pixels in the first four columns and brighter gray pixels in the 5th and 6th column of the pattern 1532.
Pattern 1542 refers to a vehicle 330 position as shown by image 1540 where the leading edge of the vehicle 330 is almost fully overlapping the wireless power transfer structure 200, thus causing all detection outputs to exceed the detection threshold and a majority thereof to exceed the saturation level resulting in white pixels in the first 6 columns and brighter gray pixels in the 7th and 8th column of the pattern 1542. At this stage, the pattern 1542 also shows a gray area in the first two columns caused by an inhomogeneous structure of the vehicle's 330 underbody (e.g., by a different material or a cavity in the underbody).
Pattern 1552 refers to a vehicle 330 position as shown by image 1550 where the leading edge of the vehicle 330 entirely overlaps the wireless power transfer structure 200, thus causing detection outputs to exceed the detection threshold in all columns. The gray area caused by the inhomogeneous underbody and that has become visible in the pattern 1542 has now moved to the 4th and 6th column of the pattern 1552.
Pattern 1562 refers to a vehicle 330 position as shown by image 1560 where the vehicle-based wireless power transfer structure 310 has reached the edge of the ground-based wireless power transfer structure 200 that starts now to also impact the pattern 1562. Since the wireless power transfer structure 310 includes different materials (e.g., Litz wire made of copper, ferrite, aluminum, and other conductive and non-conductive materials, its impact on the individual inductive sense elements of the array 107 may be highly variable. While ferrite materials tend to produce a positive reactance change, highly conductive materials such as copper and aluminum tend to cause a negative reactance change. Depending on the actual relative position of the wireless power transfer structure 310, the impact of some portions of the wireless power transfer structure 310 on some inductive sense elements (e.g., inductive sense element 107a) may cancel out producing the dark gray area in column 1 of the pattern 1562.
Pattern 1572 refers to a vehicle 330 position as shown by image 1570 where the center of the vehicle-based wireless power transfer structure 310 has just surpassed the edge of the ground-based wireless power transfer structure 200 producing a unique pattern of different gray levels in the first three columns. The gray area caused by the underbody inhomogeneity has now proceeded to the last two columns of the pattern 1572.
Pattern 1582 refers to a vehicle 330 position as shown by image 1580 where the vehicle-based wireless power transfer structure 310 now fully overlaps with the top surface of the ground-based wireless power transfer structure 200. The grayscale pattern as produced by the vehicle-based wireless power transfer structure 310 is now almost entirely visible, while the underbody inhomogeneity has already left the sensitive area of the inductive sensing array 107 (not visible anymore in the pattern of image 1580).
Pattern 1592 refers to a vehicle 330 position as shown by image 1590 where the vehicle-based wireless power transfer structure 310 is now well aligned with the ground-based wireless power transfer structure 200 displaying the grayscale pattern centered in the 8×8 pattern 1592.
The patterns 1502 to 1592 as used in
Patterns (e.g., 2×2 patterns) may also be produced from detection outputs of the multi-purpose detection circuit 100 associated to the plurality of capacitive sense elements 109a, 109b, . . . 109n as illustrated in
Therefore, in some implementations, the patterns 1502 to 1592 may also refer to a pattern produced by mapping detection output values of the multi-purpose detection circuit 100 associated with at least one of the plurality of inductive sense elements 107a, 107b, . . . , 107n and the plurality of capacitive sense elements 109a, 109b, . . . , 109c.
In an aspect of the multi-purpose detection circuit 100, patterns as produced by detection outputs associated with at least one of the plurality of inductive sense circuits 106 and the plurality of capacitive sense circuits as previously described with reference to
Stated more generally, detection outputs associated with at least one inductive sense circuit (e.g., inductive sense circuit 106a) are used to reduce a false positive detection probability of LOD. Conversely, detection outputs associated with at least one capacitive sense circuit (e.g., capacitive sense circuit 108a) are used to reduce a false positive detection probability of FOD.
In some implementations, detection outputs associated with at least one inductive sense circuit (e.g., inductive sense circuit 106a) and at least one capacitive sense circuit (e.g., capacitive sense circuit 108a) are used to dynamically adjust a detection threshold of the multi-purpose detection circuit 100 where the detection threshold refers to at least one of FOD and LOD. Dynamically adjusting a detection threshold is described in U.S. patent application Ser. No. 16/392,464, titled Extended Foreign Object Detection Signal Processing.
In another aspect of the multi-purpose detection circuit 100, patterns as produced by detection outputs associated with at least one of the plurality of inductive sense circuits 106 and the plurality of capacitive sense circuits as previously described with reference to
In a further aspect of the multi-purpose detection circuit 100, patterns as produced by detection outputs associated with at least one of the plurality of inductive sense circuits 106 and the plurality of capacitive sense circuits as previously described with reference to
In some implementations of the multi-purpose detection circuit 100, the relative position is at least in part determined by using an image correlation technique (e.g., a similar technique as employed in the computer mouse using a laser sensor for surface structure detection).
In another implementation of the multi-purpose detection circuit 100, the relative position is determined by tracking a “front-wave” in successively obtained patterns as produced by detection outputs associated with at least one of the plurality of inductive sense circuits 106 and the plurality of capacitive sense circuits and as illustrated by the patterns 1512 to 1542 of
In another aspect of the multi-purpose detection circuit 100, patterns as produced by detection outputs associated with at least one of the plurality of inductive sense circuits 106 and the plurality of capacitive sense circuits as previously described with reference to
In a further aspect of the multi-purpose detection circuit 100, patterns as produced by detection outputs associated with at least one of the plurality of inductive sense circuits 106 and the plurality of capacitive sense circuits as previously described with reference to
In yet another aspect of the multi-purpose detection circuit 100, patterns as produced by detection outputs associated with at least one of the plurality of inductive sense circuits 106 and the plurality of capacitive sense circuits as previously described with reference to
In yet a further aspect of the multi-purpose detection circuit 100, patterns as produced by detection outputs associated with at least one of the plurality of inductive sense circuits 106 and the plurality of capacitive sense circuits as previously described with reference to
Other positioning systems may include systems based on using at least one of an inductive and capacitive passive beacon transponder as previously discussed e.g., with reference to
The various operations of methods described above may be performed by any suitable means capable of performing the corresponding functions. The means may include various hardware and/or software component(s) and/or module(s), including, but not limited to a circuit, an application-specific integrated circuit (ASIC), or processor. Generally, where there are operations illustrated in figures, those operations may have corresponding counterpart means-plus-function components with similar numbering.
As used herein, the term “determining” encompasses a wide variety of actions. For example, “determining” may include calculating, computing, processing, deriving, investigating, looking up (e.g., looking up in a table, a database, or another data structure), ascertaining, and the like. Also, “determining” may include receiving (e.g., receiving information), accessing (e.g., accessing data in a memory), and the like. Also, “determining” may include resolving, selecting, choosing, establishing, and the like.
As used herein, a phrase referring to “at least one of” a list of items refers to any combination of those items, including single members. As an example, “at least one of: a, b, or c” is intended to cover: a, b, c, a-b, a-c, b-c, and a-b-c, as well as any combination with multiples of the same element (e.g., a-a, a-a-a, a-a-b, a-a-c, a-b-b, a-c-c, b-b, b-b-b, b-b-c, c-c, and c-c-c or any other ordering of a, b, and c).
The various illustrative logical blocks, modules and circuits described in connection with the present disclosure may be implemented or performed with a general-purpose processor, a digital signal processor (DSP), an ASIC, a field programmable gate array (FPGA) or other programmable logic device (PLD), discrete gate or transistor logic, discrete hardware components, or any combination thereof designed to perform the functions described herein. A processor may be a microprocessor, but in the alternative, the processor may be any commercially available processor, controller, microcontroller, or state machine. A processor may also be implemented as a combination of computing devices, e.g., a combination of a DSP and a microprocessor, a plurality of microprocessors, one or more microprocessors in conjunction with a DSP core, or any other such configuration.
The methods disclosed herein comprise one or more steps or actions for achieving the described method. The method steps and/or actions may be interchanged with one another without departing from the scope of the claims. In other words, unless a specific order of steps or actions is specified, the order and/or use of specific steps and/or actions may be modified without departing from the scope of the claims.
The functions described may be implemented in hardware, software, firmware, or any combination thereof. If implemented in hardware, an example hardware configuration may comprise a processing system in a wireless node. The processing system may be implemented with a bus architecture. The bus may include any number of interconnecting buses and bridges depending on the specific application of the processing system and the overall design constraints. The bus may link together various circuits including a processor, machine-readable media, and a bus interface. The bus interface may be used to connect a network adapter, among other things, to the processing system via the bus. The bus may also link various other circuits such as timing sources, peripherals, voltage regulators, power management circuits, and the like.
It is to be understood that the claims are not limited to the precise configuration and components illustrated above. Various modifications, changes and variations may be made in the arrangement, operation and details of the methods and apparatus described above without departing from the scope of the claims.
This application claims priority under 35 U.S.C. § 119(e) to U.S. Provisional Application 63/131,291, filed on Dec. 20, 2020 which is incorporated herein by reference in its entirety. This application is additionally a Continuation-in-Part of U.S. Utility application Ser. No. 17/077,124 filed on Oct. 22, 2020, which in turn claims priority under 35 U.S.C. § 119(e) to U.S. Provisional Application 62/926,307, filed on Oct. 25, 2019 which is additionally incorporated herein by reference in its entirety.
Number | Date | Country | |
---|---|---|---|
63244165 | Sep 2021 | US | |
62926307 | Oct 2019 | US |
Number | Date | Country | |
---|---|---|---|
Parent | 17077124 | Oct 2020 | US |
Child | 17931429 | US |