CIRCUITS AND METHODS FOR DRIVING LIGHT SOURCES

Abstract

A driving circuit for driving a light-emitting diode (LED) light source includes a buck-boost converter and a controller. The buck-boost converter receives an input voltage and an input current and powers the LED light source, and comprises a switch controlled by a driving signal. The controller receives a first signal indicating a current through the LED light source, and generates the driving signal based on the first signal to control the switch and to adjust the current through the LED light source. The buck-boost converter further comprises a current sensor which provides a second signal indicating an instant current flowing through the buck-boost converter, wherein the first signal is derived from the second signal, and wherein a reference ground of the controller is different from a ground of the driving circuit.

Description

BACKGROUND

FIG. 1 shows a block diagram of a conventional circuit 100 for driving a light source, e.g., a light emitting diode (LED) string 108. The circuit 100 is powered by a power source 102 which provides an input voltage VIN. The circuit 100 includes a buck converter for providing a regulated voltage VOUT to an LED string 108 under control of a controller 104. The buck converter includes a diode 114, an inductor 112, a capacitor 116, and a switch 106. A resistor 110 is coupled in series with the switch 106. When the switch 106 is turned on, the resistor 110 is coupled to the inductor 112 and the LED string 108, and can provide a feedback signal indicative of a current flowing through the inductor 112. When the switch 106 is turned off, the resistor 110 is disconnected from the inductor 112 and the LED string 108, and thus no current flows through the resistor 110.

The switch 106 is controlled by the controller 104. When the switch 106 is turned on, a current flows through the LED string 108, the inductor 112, the switch 106, and the resistor 110 to ground. The current increases due to the inductance of the inductor 112. When the current reaches a predetermined peak current level, the controller 104 turns off the switch 106. When the switch 106 is turned off, a current flows through the LED string 108, the inductor 112 and the diode 114. The controller 104 can turn on the switch 106 again after a time period. Thus, the controller 104 controls the buck converter based on the predetermined peak current level. However, the average level of the current flowing through the inductor 112 and the LED string 108 can vary with the inductance of the inductor 112, the input voltage VIN, and the regulated voltage VOUT across the LED string 108. Therefore, the average level of the current flowing through the inductor 112 (the average current flowing through the LED string 108) may not be accurately controlled.

SUMMARY

In one embodiment, a driving circuit for driving a light-emitting diode (LED) light source includes a buck-boost converter and a controller. The buck-boost converter receives an input voltage and an input current and powers the LED light source, and comprises a switch controlled by a driving signal. The controller receives a first signal indicating a current through the LED light source, and generates the driving signal based on the first signal to control the switch and to adjust the current through the LED light source. The buck-boost converter further comprises a current sensor which provides a second signal indicating an instant current flowing through the buck-boost converter, wherein the first signal is derived from the second signal, and wherein a reference ground of the controller is different from a ground of the driving circuit.

BRIEF DESCRIPTION OF THE DRAWINGS

Features and advantages of embodiments of the claimed subject matter will become apparent as the following detailed description proceeds, and upon reference to the drawings, wherein like numerals depict like parts, and in which:

FIG. 1 shows a block diagram of a conventional circuit for driving a light source.

FIG. 2 shows a block diagram of a driving circuit, in accordance with one embodiment of the present invention.

FIG. 3 shows an example for a schematic diagram of a driving circuit, in accordance with one embodiment of the present invention.

FIG. 4 shows an example of the controller in FIG. 3, in accordance with one embodiment of the present invention.

FIG. 5 shows signal waveforms of signals associated with a controller in FIG. 4, in accordance with one embodiment of the present invention.

FIG. 6 shows another example of the controller in FIG. 3, in accordance with one embodiment of the present invention.

FIG. 7 shows signal waveforms of signals associated with a controller in FIG. 6, in accordance with one embodiment of the present invention.

FIG. 8 shows another example for a schematic diagram of a driving circuit, in accordance with one embodiment of the present invention.

FIG. 9A shows another block diagram of a driving circuit, in accordance with one embodiment of the present invention.

FIG. 9B shows an example of waveforms of signals generated or received by a driving circuit in FIG. 9A, in accordance with one embodiment of the present invention.

FIG. 10 shows an example for a schematic diagram of a driving circuit, in accordance with one embodiment of the present invention.

FIG. 11 shows an example of a controller in FIG. 9A, in accordance with one embodiment of the present invention.

FIG. 12 illustrates a waveform of signals generated or received by a driving circuit, in accordance with one embodiment of the present invention.

FIG. 13 illustrates a flowchart of operations performed by a circuit for driving a load, in accordance with one embodiment of the present invention.

FIG. 14 shows an example for a schematic diagram of a driving circuit, in accordance with one embodiment of the present invention.

FIG. 15 shows an example of the controller in FIG. 14, in accordance with one embodiment of the present invention.

FIG. 16 shows another example for a schematic diagram of a driving circuit, in accordance with one embodiment of the present invention.

FIG. 17 shows an example for a schematic diagram of a driving circuit, in accordance with one embodiment of the present invention.

FIG. 18 illustrates a waveform of signals generated or received by a driving circuit, in accordance with one embodiment of the present invention.

DETAILED DESCRIPTION

Reference will now be made in detail to the embodiments of the present invention. While the invention will be described in conjunction with these embodiments, it will be understood that they are not intended to limit the invention to these embodiments. On the contrary, the invention is intended to cover alternatives, modifications and equivalents, which may be included within the spirit and scope of the invention as defined by the appended claims.

Furthermore, in the following detailed description of the present invention, numerous specific details are set forth in order to provide a thorough understanding of the present invention. However, it will be recognized by one of ordinary skill in the art that the present invention may be practiced without these specific details. In other instances, well known methods, procedures, components, and circuits have not been described in detail as not to unnecessarily obscure aspects of the present invention.

Embodiments in accordance with the present invention provide circuits and methods for controlling power converters that can be used to power various types of loads, for example, a light source. In one embodiment, the circuit can include a current sensor operable for monitoring a current flowing through an energy storage element, e.g., an inductor, and include a controller operable for controlling a switch coupled to the inductor so as to control an average current of the light source to a target current. The current sensor can monitor the current through the inductor when the switch is on and also when the switch is off.

FIG. 2 shows a block diagram of a driving circuit 200, in accordance with one embodiment of the present invention. The driving circuit 200 includes a rectifier 204 which receives an input voltage from a power source 202 and provides a rectified voltage to a power converter 206. The power converter 206, receiving the rectified voltage, provides output power for a load, e.g., a LED string 208. The power converter 206 can be a buck converter or a boost converter. In one embodiment, the power converter 206 includes an energy storage element 214 and a current sensor 218 for sensing an electrical condition of the energy storage element 214. The current sensor 218 provides a sensing signal ISEN to a controller 210, which indicates an instant current flowing through the energy storage element 214. The driving circuit 200 can further include a filter 212 operable for generating a sensing signal IAVG based on the sensing signal ISEN, which indicates an average current flowing through the energy storage element 214. The controller 210 receives the sensing signal ISEN and the sensing signal IAVG, and controls the average current flowing through the energy storage element 214 to a target current level, in one embodiment.

FIG. 3 shows an example for a schematic diagram of a driving circuit 300, in accordance with one embodiment of the present invention. Elements labeled the same as in FIG. 2 have similar functions. In the example of FIG. 3, the driving circuit 300 includes a rectifier 204, a power converter 206, a filter 212, and a controller 210. By way of example, the rectifier 204 is a bridge rectifier which includes diodes D1˜D4. The rectifier 204 rectifies the voltage from the power source 202. The power converter 206 receives the rectified voltage from the rectifier 204 and provides output power for powering a load, e.g., an LED string 208.

In the example of FIG. 3, the power converter 206 is a buck converter including a capacitor 308, a switch 316, a diode 314, a current sensor 218 (e.g., a resistor), coupled inductors 302 and 304, and a capacitor 324. The diode 314 is coupled between the switch 316 and ground of the driving circuit 300. The capacitor 324 is coupled in parallel with the LED string 208. In one embodiment, the inductors 302 and 304 are both electrically and magnetically coupled together. More specifically, the inductor 302 and the inductor 304 are electrically coupled to a common node 333. In the example of FIG. 3, the common node 333 is between the current sensor 218 and the inductor 302. However, the invention is not so limited; the common node 333 can also locate between the switch 316 and the current sensor 218. The common node 333 provides a reference ground for the controller 210. The reference ground of the controller 210 is different from the ground of the driving circuit 300, in one embodiment. By turning the switch 316 on and off, a current flowing through the inductor 302 can be adjusted, thereby adjusting the power provided to the LED string 208. The inductor 304 senses an electrical condition of the inductor 302, for example, whether the current flowing through the inductor 302 decreases to a first predetermined current level.

The current sensor 218 has one end coupled to a node between the switch 316 and the cathode of the diode 314, and the other end coupled to the inductor 302. The current sensor 218 provides a sensing signal ISEN indicating an instant current flowing through the inductor 302 when the switch 316 is on and also when the switch 316 is off. In other words, the current sensor 218 can sense the instant current flowing through the inductor 302 regardless of whether the switch 316 is on or off. The filter 212 coupled to the current sensor 218 generates a sensing signal IAVG indicating an average current flowing through the inductor 302. In one embodiment, the filter 212 includes a resistor 320 and a capacitor 322.

The controller 210 receives the sensing signal ISEN and the sensing signal IAVG, and controls an average current flowing through the inductor 302 to a target current level by turning the switch 316 on and off. A capacitor 324 absorbs ripple current flowing through the LED string 208 such that the current flowing through the LED string 208 is smoothed and substantially equal to the average current flowing through the inductor 302. As such, the current flowing through the LED string 208 can have a level that is substantially equal to the target current level. As used herein, “substantially equal to the target current level” means that the current flowing through the LED string 208 may be slightly different from the target current level but within a range such that the current ripple caused by the non-ideality of the circuit components can be neglected and the power transferred from the inductor 304 to the controller 210 can be neglected.

In the example of FIG. 3, the controller 210 has terminals ZCD, GND, DRV, VDD, CS, COMP and FB. The terminal ZCD is coupled to the inductor 304 for receiving a detection signal AUX indicating an electrical condition of the inductor 302, for example, whether the current flowing through the inductor 302 decreases to a first predetermined current level, e.g., zero. The detection signal AUX can also indicate whether the LED string 208 is in an open circuit condition. The terminal DRV is coupled to the switch 316 and generates a driving signal, e.g., a pulse-width modulation signal PWM1, to turn the switch 316 on and off. The terminal VDD is coupled to the inductor 304 for receiving power from the inductor 304. The terminal CS is coupled to the current sensor 218 and is operable for receiving the sensing signal ISEN indicating an instant current flowing through the inductor 302. The terminal COMP is coupled to the reference ground of the controller 210 through a capacitor 318. The terminal FB is coupled to the current sensor 218 through the filter 212 and is operable for receiving the sensing signal IAVG which indicates an average current flowing through the inductor 302. In the example of FIG. 3, the terminal GND, that is, the reference ground for the controller 210, is coupled to the common node 333 between the current sensor 218, the inductor 302, and the inductor 304.

The switch 316 can be an N channel metal oxide semiconductor field effect transistor (NMOSFET). The conductance status of the switch 316 is determined based on a difference between the gate voltage of the switch 316 and the voltage at the terminal GND (the voltage at the common node 333). Therefore, the switch 316 is turned on and turned off depending upon the pulse-width modulation signal PWM1 from the terminal DRV. When the switch 316 is on, the reference ground of the controller 210 is higher than the ground of the driving circuit 300, making the invention suitable for power sources having relatively high voltages.

In operation, when the switch 316 is turned on, a current flows through the switch 316, the current sensor 218, the inductor 302, the LED string 208 to the ground of the driving circuit 300. When the switch 316 is turned off, a current continues to flow through the current sensor 218, the inductor 302, the LED string 208 and the diode 314. The inductor 304 magnetically coupled to the inductor 302 detects an electrical condition of the inductor 302, for example, whether the current flowing through the inductor 302 decreases to a first predetermined current level. Therefore, the controller 210 monitors the current flowing through the inductor 302 through the detection signal AUX, the sensing signal ISEN, and the sensing signal IAVG, and control the switch 316 by a pulse-width modulation signal PWM1 so as to control an average current flowing through the inductor 302 to a target current level, in one embodiment. As such, the current flowing through the LED string 208, which is filtered by the capacitor 324, can also be substantially equal to the target current level.

In one embodiment, the controller 210 determines whether the LED string 208 is in an open circuit condition based on the detection signal AUX. If the LED string 208 is open, the voltage across the capacitor 324 increases. When the switch 316 is off, the voltage across the inductor 302 increases and the voltage of the detection signal AUX increases accordingly. As a result, the current flowing through the terminal ZCD into the controller 210 increases. Therefore, the controller 210 monitors the detection signal AUX and if the current flowing into the controller 210 increases above a current threshold when the switch 316 is off, the controller 210 determines that the LED string 208 is in an open circuit condition.

The controller 210 can also determine whether the LED string 208 is in a short circuit condition based on the voltage at the terminal VDD. If the LED string 208 is in a short circuit condition, when the switch 316 is off, the voltage across the inductor 302 decreases because both terminals of the inductor 302 are coupled to ground of the driving circuit 300. The voltage across the inductor 304 and the voltage at the terminal VDD decrease accordingly. If the voltage at the terminal VDD decreases below a voltage threshold when the switch 316 is off, the controller 210 determines that the LED string 208 is in a short circuit condition.

FIG. 4 shows an example of the controller 210 in FIG. 3, in accordance with one embodiment of the present invention. FIG. 5 shows signal waveforms of signals associated with the controller 210 in FIG. 4, in accordance with one embodiment of the present invention. FIG. 4 is described in combination with FIG. 3 and FIG. 5.

In the example of FIG. 4, the controller 210 includes an error amplifier 402, a comparator 404, and a pulse-width modulation signal generator 408. The error amplifier 402 generates an error signal VEA based on a difference between a reference signal SET and the sensing signal IAVG. The reference signal SET can indicate a target current level. The sensing signal IAVG is received at the terminal FB and can indicate an average current flowing through the inductor 302. The error signal VEA can be used to adjust the average current flowing through the inductor 302 to the target current level. The comparator 404 is coupled to the error amplifier 402 and compares the error signal VEA with the sensing signal ISEN. The sensing signal ISEN is received at the terminal CS and indicates an instant current flowing through the inductor 302. The detection signal AUX is received at the terminal ZCD and indicates whether the current flowing through the inductor 302 decreases to a first predetermined current level, e.g., zero. The pulse-width modulation signal generator 408 is coupled to the comparator 404 and the terminal ZCD, and can generate a pulse-width modulation signal PWM1 based on an output of the comparator 404 and the detection signal AUX. The pulse-width modulation signal PWM1 is applied to the switch 316 via the terminal DRV to control a conductance status of the switch 316.

In operation, the pulse-width modulation signal generator 408 can generate the pulse-width modulation signal PWM1 having a first state (e.g., logic 1) to turn on the switch 316. When the switch 316 is turned on, a current flows through the switch 316, the current sensor 218, the inductor 302, the LED string 208 to the ground of the driving circuit 300. The current flowing through the inductor 302 increases such that the voltage of the sensing signal ISEN increases. The detection signal AUX has a negative voltage level when the switch 316 is turned on, in one embodiment. In the controller 210, the comparator 404 compares the error signal VEA with the sensing signal ISEN. When the voltage of the sensing signal ISEN increases above the voltage of the error signal VEA, the output of the comparator 404 is logic 0, otherwise the output of the comparator 404 is logic 1, in one embodiment. In other words, the output of the comparator 404 includes a series of pulses. The pulse-width modulation signal generator 408 generates the pulse-width modulation signal PWM1 having a second state (e.g., logic 0) in response to a negative-going edge of the output of the comparator 404 to turn off the switch 316. The voltage of the detection signal AUX changes to a positive voltage level when the switch 316 is turned off. When the switch 316 is turned off, a current flows through the current sensor 218, the inductor 302, the LED string 208 and the diode 314. The current flowing through the inductor 302 decreases such that the voltage of the sensing signal ISEN decreases. When the current flowing through the inductor 302 decreases to a first predetermined current level (e.g., zero), a negative-going edge occurs to the voltage of the detection signal AUX. Receiving a negative-going edge of the detection signal AUX, the pulse-width modulation signal generator 408 generates the pulse-width modulation signal PWM1 having the first state (e.g., logic 1) to turn on the switch 316.

In one embodiment, a duty cycle of the pulse-width modulation signal PWM1 is determined by the error signal VEA. If the voltage of the sensing signal IAVG is less than the voltage of the reference signal SET, the error amplifier 402 increases the voltage of the error signal VEA so as to increase the duty cycle of the pulse-width modulation signal PWM1. Accordingly, the average current flowing through the inductor 302 increases until the voltage of the sensing signal IAVG reaches the voltage of the reference signal SET. If the voltage of the sensing signal IAVG is greater than the voltage of the reference signal SET, the error amplifier 402 decreases the voltage of the error signal VEA so as to decrease the duty cycle of the pulse-width modulation signal PWM1. Accordingly, the average current flowing through the inductor 302 decreases until the voltage of the sensing signal IAVG drops to the voltage of the reference signal SET. As such, the average current flowing through the inductor 302 can be maintained to be substantially equal to the target current level.

The controller 210 can further include an Under Voltage Lockout (UVLO) circuit 401 coupled to the terminal VDD for selectively turning on one or more components of the controller 210 according to different power conditions. In one embodiment, the UVLO circuit 401 is operable for turning on all the components of the controller 210 when the voltage at the terminal VDD is greater than a first predetermined voltage. The UVLO circuit 401 is operable for turning off all the components of the controller 210 when the voltage at the terminal VDD is less than a second predetermined voltage. In one embodiment, the first predetermined voltage is greater than the second predetermined voltage. The terminal VDD is used to provide power to the controller 210. The terminal GND is coupled to the reference ground for the controller 210.

FIG. 6 shows another example of the controller 210 in FIG. 3, in accordance with one embodiment of the present invention. FIG. 7 shows waveforms of signals associated with the controller 210 in FIG. 6, in accordance with one embodiment of the present invention. FIG. 6 is described in combination with FIG. 3 and FIG. 7.

In the example of FIG. 6, the controller 210 includes an error amplifier 602, a comparator 604, a saw-tooth signal generator 606, a reset signal generator 608, and a pulse-width modulation signal generator 610. The error amplifier 602 generates an error signal VEA based on a reference signal SET and the sensing signal IAVG. The reference signal SET indicates a target current level. The sensing signal IAVG is received at the terminal FB and indicates an average current flowing through the inductor 302. The error signal VEA is used to adjust the average current flowing through the inductor 302 to the target current level. The saw-tooth signal generator 606 generates a saw-tooth signal SAW. The comparator 604 is coupled to the error amplifier 602 and the saw-tooth signal generator 606, and compares the error signal VEA with the saw-tooth signal SAW. The reset signal generator 608 generates a reset signal RESET which is applied to the saw-tooth signal generator 606 and the pulse-width modulation signal generator 610. The switch 316 can be turned on in response to the reset signal RESET. The pulse-width modulation signal generator 610 is coupled to the comparator 604 and the reset signal generator 608, and generates a pulse-width modulation (PWM) signal PWM1 based on an output of the comparator 604 and the reset signal RESET. The pulse-width modulation signal PWM1 is applied to the switch 316 via the terminal DRV to control a conductance status of the switch 316.

In one embodiment, the reset signal RESET is a pulse signal having a constant frequency. In another embodiment, the reset signal RESET is a pulse signal configured in a way such that a time period Toff during which the switch 316 is off is constant. For example, in FIG. 5, the time period during which the pulse-width modulation signal PWM1 is logic 0 can be constant.

In operation, the pulse-width modulation signal generator 610 generates the pulse-width modulation signal PWM1 having a first state (e.g., logic 1) to turn on the switch 316 in response to a pulse of the reset signal RESET. When the switch 316 is turned on, a current flows through the switch 316, the current sensor 218, the inductor 302, the LED string 208 to the ground of the driving circuit 300. The saw-tooth signal SAW generated by the saw-tooth signal generator 606 starts to increase from an initial level INI in response to a pulse of the reset signal RESET. When the voltage of the saw-tooth signal SAW increases to the voltage of the error signal VEA, the pulse-width modulation signal generator 610 generates the pulse-width modulation signal PWM1 having a second state (e.g., logic 0) to turn off the switch 316. The saw-tooth signal SAW is reset to the initial level INI until a next pulse of the reset signal RESET is received by the saw-tooth signal generator 606. The saw-tooth signal SAW starts to increase from the initial level INI again in response to the next pulse.

In one embodiment, a duty cycle of the pulse-width modulation signal PWM1 is determined by the error signal VEA. If the voltage of the sensing signal IAVG is less than the voltage of the reference signal SET, the error amplifier 602 increases the voltage of the error signal VEA so as to increase the duty cycle of the pulse-width modulation signal PWM1. Accordingly, the average current flowing through the inductor 302 increases until the voltage of the sensing signal IAVG reaches the voltage of the reference signal SET. If the voltage of the sensing signal IAVG is greater than the voltage of the reference signal SET, the error amplifier 602 decreases the voltage of the error signal VEA so as to decrease the duty cycle of the pulse-width modulation signal PWM1. Accordingly, the average current flowing through the inductor 302 decreases until the voltage of the sensing signal IAVG drops to the voltage of the reference signal SET. As such, the average current flowing through the inductor 302 can be maintained to be substantially equal to the target current level.

FIG. 8 shows another example for a schematic diagram of a driving circuit 800, in accordance with one embodiment of the present invention. Elements labeled the same as in FIG. 2 and FIG. 3 have similar functions.

The terminal VDD of the controller 210 is coupled to the rectifier 204 through a switch 804 for receiving the rectified voltage from the rectifier 204. A Zener diode 802 is coupled between the switch 804 and the reference ground of the controller 210, and maintains the voltage at the terminal VDD at a substantially constant level. In the example of FIG. 8, the terminal ZCD of the controller 210 is electrically coupled to the inductor 302 for receiving a detection signal AUX indicating an electrical condition of the inductor 302, e.g., whether the current flowing through the inductor 302 decreases to a first predetermined current level, e.g., zero. The common node 333 can provide the reference ground for the controller 210.

Accordingly, embodiments in accordance with the present invention provide circuits and methods for controlling a power converter that can be used to power various types of loads. In one embodiment, the power converter provides a substantially constant current to power a load such as a light emitting diode (LED) string. In another embodiment, the power converter provides a substantially constant current to charge a battery. Advantageously, compared with the conventional driving circuit in FIG. 1, the average current to the load or the battery can be controlled more accurately. Furthermore, the circuits according to present invention can be suitable for power sources having relatively high voltages.

FIG. 9A shows another block diagram of a driving circuit 900, in accordance with one embodiment of the present invention. Elements labeled the same as in FIG. 2 and FIG. 3 have similar functions. In the example of FIG. 9A, the driving circuit 900 includes a filter 920 coupled to a power source 202, a rectifier 204, a power converter 906, a LED string 208, a saw-tooth signal generator 902, and a controller 910. The power source 202 generates an AC input voltage V_AC, e.g., having a sinusoidal waveform, and an AC input current I_AC. The AC input current I_AC flows into the filter 920 and a current I_AC′ flows from the filter 920 to the rectifier 204. The rectifier 204 receives the AC input voltage V_AC via the filter 920 and provides a rectified AC voltage V_IN and a rectified AC current I_IN at the power line 912 coupled between the rectifier 204 and the power converter 906. The power converter 906 converts the rectified AC voltage V_IN to an output voltage V_OUT to power the LED string 208. The controller 910 coupled to the power converter 906 controls the power converter 906 to regulate a current I_OUT through the LED string 208 and correct a power factor of the driving circuit 900.

The controller 910 generates a driving signal 962. In one embodiment, the power converter 906 includes a switch 316 which is controlled by the driving signal 962. As such, a current I_OUT flowing through the LED string 208 is regulated according to the driving signal 962. In one embodiment, the power converter 906 further generates a sensing signal IAVG indicating the current I_OUT through the LED string 208.

In one embodiment, the saw-tooth signal generator 902 coupled to the controller 910 generates a saw-tooth signal 960 according to the driving signal 962. For example, the driving signal 962 can be a pulse-width modulation (PWM) signal. In one embodiment, when the driving signal 962 is logic high, the saw-tooth signal 960 is increased; when the driving signal 962 is logic low, the saw-tooth signal 960 drops to a predetermined voltage level, e.g., zero volt.

Advantageously, the controller 910 generates the driving signal 962 based on signals including the saw-tooth signal 960 and the sensing signal IAVG. The driving signal 962 controls the switch 316 to maintain the current I_OUT through the LED string 208 at a target level, which improves the accuracy of the current control. In addition, the driving signal 962 controls the switch 316 to adjust an average current I_IN—AVG of the rectified AC current I_IN to be substantially in phase with the rectified AC voltage V_IN, which corrects a power factor of the driving circuit 900. The operation of the driving circuit 900 is further described in FIG. 9B.

FIG. 9B shows an example of waveforms of signals associated with the driving circuit 900 in FIG. 9A, in accordance with one embodiment of the present invention. FIG. 9B is described in combination with FIG. 9A. FIG. 9B shows the AC input voltage V_AC, the rectified AC voltage V_IN, the rectified AC current I_IN, the current I_AC′, and the AC input current I_AC.

For illustrative purposes but not limitation, the AC input voltage V_AC has a sinusoidal waveform. The rectifier 204 rectifies the AC input voltage V_AC. In the example of FIG. 9B, the rectified AC voltage V_IN has a rectified sinusoidal waveform, in which positive waves of the AC input voltage V_AC remains and negative waves of the AC input voltage V_AC is converted to corresponding positive waves.

In one embodiment, the driving signal 962 generated by the controller 910 controls the rectified AC current I_IN. In one embodiment, the rectified AC current I_IN increases from a predetermined level, e.g., zero ampere. After the rectified AC current I_IN reaches a level proportional to the rectified AC input voltage V_IN, the rectified AC current I_IN drops to the predetermined level. Thus, as shown in FIG. 9B, the waveform of the average current I_IN—AVG of the rectified AC current I_IN is substantially in phase with the waveform of the rectified AC voltage V_IN.

The rectified AC current I_IN flowing from the rectifier 204 to the power converter 906 is a rectified current of the current I_AC′ flowing into the rectifier 204. As shown in FIG.9B, the current I_AC′ has positive waves similar to those of the rectified AC current I_IN when the AC input voltage V_AC is positive and has negative waves corresponding to those of the rectified AC current I_IN when the AC input voltage V_AC is negative.

In one embodiment, by employing a filter 920 between the power source 202 and the rectifier 204, the AC input current I_AC is equal to or proportional to an average current of the current I_AC′. Therefore, as shown in FIG. 12, the waveform of the AC input current I_AC is substantially in phase with the waveform of the AC input voltage V_AC. Ideally, the AC input voltage V_AC and the AC input current I_AC are in phase. However, in practical application, there might be a slight phase difference due to capacitors in the filter 920 and the power converter 906. Moreover, the shape of the waveform of the AC input current I_AC is similar to the shape of the waveform of the AC input voltage V_AC. Therefore, a power factor of the driving circuit 900 is corrected, which improves the power quality of the driving circuit 900.

FIG. 10 shows an example for a schematic diagram of a driving circuit 1000, in accordance with one embodiment of the present invention. Elements labeled the same as in FIG. 2, FIG. 3 and FIG. 9A have similar functions. FIG. 10 is described in combination with FIG. 4, FIG. 5 and FIG. 9A.

In the example of FIG. 10, the driving circuit 1000 includes a filter 920 coupled to a power source 202, a rectifier 204, a power converter 906, a load such as a LED string 208, a saw-tooth signal generator 902, and a controller 910. In one embodiment, the LED string 208 includes an LED light source such as an LED string. This invention is not so limited; the LED string 208 can include other types of light sources or other types of loads such as a battery pack. The filter 920 can be, but is not limited to, an inductor-capacitor (L-C) filter including a pair of inductors and a pair of capacitors. In one embodiment, the controller 910 includes multiple terminals such as a ZCD terminal, a GND terminal, a DRV terminal, a VDD terminal, an FB terminal, a COMP terminal, and a CS terminal.

In one embodiment, the power converter 906 includes an input capacitor 1008 coupled to the power line 912. The input capacitor 1008 reduces ripples of the rectified AC voltage V_IN to smooth the waveform of the rectified AC voltage V_IN. In one embodiment, the capacitor 1008 has a relatively small capacitance, e.g., less than 0.5 μF, to help eliminate or reduce any distortion of the rectified AC voltage V_IN. Moreover, in one embodiment, a current flowing through the capacitor 1008 can be ignored due to the relatively small capacitance. Thus, the rectified AC current I_IN flowing through the switch 316 is approximately equal to the current from the rectifier 204 when the switch 316 is on.

The power converter 906 operates similarly as the power converter 206 in FIG. 3. In one embodiment, the energy storage element 214 includes inductors 302 and 304 magnetically and electrically coupled with each other. The inductor 302 is coupled to the switch 316 and the LED string 208. Thus, a current I₂₁₄ flows through the inductor 302 according to the conductance status of the switch 316. More specifically, in one embodiment, the controller 910 generates the driving signal 962, e.g., a PWM signal, through the DRV terminal to switch the switch 316 to an ON state or an OFF state. When the switch 316 is turned on, the current I₂₁₄ flows from the power line 912 through the switch 316 and the inductor 302. The current I₂₁₄ increases during the ON state of the switch 316, which can be given according to equation (1):

ΔI₂₁₄=(V_IN−V_OUT)*T_ON/L₃₀₂,   (1)

where T_ON represents a time duration when the switch 316 is turned on, ΔI₂₁₄ represents a change of the current I₂₁₄, and L₃₀₂ represents the inductance of the inductor 302. In one embodiment, the controller 920 controls the driving signal 962 to maintain the time duration T_ON constant. Therefore, the change ΔI₂₁₄ of the current I₂₁₄ during the time T_ON is proportional to the rectified AC voltage V_IN if V_OUT is a substantially constant. In one embodiment, the switch 316 is turned on when the current I₂₁₄ decreases to a predetermined level, e.g., zero ampere. Accordingly, the peak level of the current I₂₁₄ is proportional to the input voltage V_IN.

When the switch 316 is turned off, the current I₂₁₄ flows from the ground through the diode 314 and the inductor 302 to the LED string 208. Accordingly, the current I₂₁₄ decreases according to equation (2):

ΔI₂₁₄=(−V_OUT)*T_OFF/L₃₀₂.   (2)

Thus, the rectified AC current I_IN is substantially equal to the current I₂₁₄ during an ON state of the switch 316 and equal to zero ampere during an OFF state of the switch 316, in one embodiment.

The inductor 304 senses an electrical condition of the inductor 302, e.g., whether the current flowing through the inductor 302 decreases to a predetermined level (e.g., zero ampere). As discussed in relation to FIG. 5, the detection signal AUX has a negative level when the switch 316 is turned on, and has a positive level when the switch 316 is turned off, in one embodiment. When the current I₂₁₄ through the inductor 302 decreases to a first predetermined current level, a negative-going edge occurs to the voltage of the detection signal AUX. The ZCD terminal of the controller 910 coupled to the inductor 304 is used to receive the detection signal AUX.

In one embodiment, the power converter 906 includes an output filter 1024. The output filter 1024 can be a capacitor having a relatively large capacitance, e.g., greater than 400 μF. As such, the current I_OUT through the LED string 208 represents an average level of the current I₂₁₄.

The current sensor 218 generates a sensing signal ISEN indicating the current flowing through the inductor 302. In one embodiment, the signal filter 212 is a resistor-capacitor (RC) filter including a resistor 320 and a capacitor 322. The signal filter 212 removes ripples of the sensing signal ISEN to generate a sensing signal IAVG of the sensing signal ISEN. Thus, in the example of FIG. 10, the sensing signal IAVG indicates the current I_OUT flowing through the LED string 208. The terminal FB of the controller 910 receives the sensing signal IAVG, in one embodiment.

The saw-tooth signal generator 902 coupled to the DRV terminal and the CS terminal is operable for generating a saw-tooth signal 960 at the CS terminal according to the driving signal 962 on the DRV terminal. By way of example, the saw-tooth signal generator 902 includes a resistor 1016 and a diode 1018 coupled in parallel between the terminal DRV and the terminal CS, and further includes a resistor 1012 and a capacitor 1014 coupled in parallel between the CS terminal and ground. In operation, the saw-tooth signal 960 varies according to the driving signal 962. More specifically, in one embodiment, the driving signal 962 is a PWM signal. When the driving signal 962 is logic high, a current I1 flows from the DRV terminal through the resistor 1016 to the capacitor 1014. Thus, the capacitor 1014 is charged and a voltage V₉₆₀ of the saw-tooth signal 960 increases. When the driving signal 962 is logic low, a current I2 flows from the capacitor 1014 through the diode 1018 to the DRV terminal. Thus, the capacitor 1014 is discharged and the voltage V₉₆₀ decreases to zero volts. The saw-tooth signal generator 902 can include other components and is not limited to the example shown in FIG. 10.

In one embodiment, the controller 910 is integrated on an integrated circuit (IC) chip. The resistors 1016 and 1012, the diode 1018, and the capacitor 1014 are peripheral components to the IC chip. Alternatively, the saw-tooth signal generator 902 and the controller 910 are both integrated on a single IC chip. In this condition, the terminal CS can be removed, which further reduces the size and the cost of the driving circuit 1000. The power converter 906 can have other configurations and is not limited to the example in FIG. 10.

FIG. 11 shows an example of the controller 910 in FIG. 9A, in accordance with one embodiment of the present invention. Elements labeled the same as in FIG. 4 and FIG. 9A have similar functions. FIG. 11 is described in combination with FIG. 4, FIG. 5, FIG. 9A and FIG. 10.

In one embodiment, the controller 910 has similar configurations as the controller 210 in FIG. 4, except that the CS terminal receives the saw-tooth signal 960 instead of the sensing signal ISEN. The controller 910 generates the driving signal 962 according to the signals including the saw-tooth signal 960, the sensing signal IAVG, and the detection signal AUX. The controller 910 includes an error amplifier 402, a comparator 404, and a pulse-width modulation (PWM) signal generator 408. The error amplifier 402 amplifies a difference between the sensing signal IAVG and a reference signal SET indicating a target current level to generate the error signal VEA. The comparator 404 compares the saw-tooth signal 960 to the error signal VEA to generate a comparing signal S. The PWM signal generator 408 generates the driving signal 962 according to the comparing signal S and the detection signal AUX.

In one embodiment, the driving signal 962 has a first state, e.g., logic high, to turn on the switch 316 when the detection signal AUX indicates that the current I₂₁₄ through the inductor 302 drops to a predetermined level, e.g., zero ampere. The driving signal 962 has a second state, e.g., logic low, to turn off the switch 316 when the saw-tooth signal 960 reaches the error signal VEA. Advantageously, since the CS terminal receives the saw-tooth signal 960 instead of the sensing signal ISEN, a peak level of the current I₂₁₄ through the inductor 302 is not limited by the error signal VEA. Thus, the current I₂₁₄ through the inductor 302 varies according to the rectified AC voltage V_IN as shown in equation (1). For example, the peak level of the current I₂₁₄ is adjusted to be proportional to the rectified AC voltage V_IN instead of the error signal VEA.

The controller 910 controls the driving signal 962 to maintain the current I_OUT at a target current level represented by the reference signal SET. For example, if the current I_OUT is greater than the target level, e.g., due to the variation of the input voltage V_IN, the error amplifier 402 decreases the error signal VEA to shorten the time duration T_ON of the ON state of the switch 316. Therefore, the average level of the current I₂₁₄ is decreased to decrease the current I_OUT. Likewise, if the current I_OUT is less than the target level, the controller 910 lengthens the time duration T_ON to increase the current I_OUT.

FIG. 12 illustrates a waveform of signals generated or received by a driving circuit, e.g., the driving circuit 900 or 1000, in accordance with one embodiment of the present invention. FIG. 12 is described in relation to FIG. 4, FIG. 9A, FIG. 9B, and FIG. 10. FIG. 12 shows the rectified AC voltage V_IN, the rectified AC current I_IN, the average current I_IN—AVG of the current I_IN, the current I_OUT flowing through the LED string 208, the sensing signal ISEN indicating the current I₂₁₄ flowing through the inductor 302, the error signal VEA, the saw-tooth signal 960, and the driving signal 962.

As shown in the example of FIG. 12, the rectified AC voltage V_IN is a rectified sinusoidal waveform. At time t1, the driving signal 962 is changed to logic high. Thus, the switch 316 is turned on and the sensing signal ISEN indicating the current I₂₁₄ through the inductor 302 increases. Meanwhile, the saw-tooth signal 960 increases according to the driving signal 962.

At time t2, the saw-tooth signal 960 reaches the error signal VEA. Accordingly, the controller 910 adjusts the driving signal 962 to logic low. The saw-tooth signal 960 drops to zero volts. The driving signal 962 turns off the switch 316, thereby decreasing the sensing signal ISEN. In other words, the saw-tooth signal 960 and the error signal VEA determine the time period T_ON when the driving signal 962 is logic high to turn on the switch 316.

At time t3, the current I₂₁₄ decreases to the first predetermined current level, e.g., zero ampere. Thus, the controller 910 adjusts the driving signal 962 to logic high to turn on the switch 316.

In one embodiment, the current I_OUT flowing through the LED string 208 is equal to or proportional to an average level of the current I₂₁₄ over a cycle period of the input voltage V_IN. As described in relation to FIG. 11, the current I_OUT is adjusted to the target current level represented by the reference signal SET. In addition, as shown in FIG. 12, the sensing signal ISEN indicating the current I₂₁₄ between t1 and t4 has same waveforms as those between t5 and t6. Thus, the average level of the current I₂₁₄ between t1 and t4 is equal to the average level of the current I₂₁₄ between t5 and t6. Accordingly, the current I_OUT is maintained at the target level. In one embodiment, the time period T_ON is determined by the saw-tooth signal 960 and the error signal VEA. In one embodiment, the time period T_ON is constant because the time period for the saw-tooth signal 960 to rise from zero volts to the error signal VEA is the same in each cycle of the driving signal 962. Based on equation (1), the change ΔI₂₁₄ of the current I₂₁₄ during the time period T_ON is proportional to the input voltage V_IN. Therefore, the peak level of the sensing signal ISEN is proportional to the rectified AC voltage V_IN as shown in FIG. 12.

The rectified AC current I_IN has a waveform similar to the waveform of the current I₂₁₄ when the switch 316 is turned on, and is substantially equal to zero ampere when the switch 316 is turned off, in one embodiment. The average current I_IN—AVG is substantially in phase with the rectified AC voltage V_IN between time t1 and t6. As described in relation to FIG. 9B, the AC input current I_AC is substantially in phase with the AC input voltage V_AC, which corrects the power factor of the driving circuit 900 to improve the power quality.

FIG. 13 illustrates a flowchart 1300 of operations performed by a circuit for driving a load, e.g., the circuit 900 or 1000 for driving an LED string 208, in accordance with one embodiment of the present invention. FIG. 13 is described in combination with FIG. 9A-FIG. 12. Although specific steps are disclosed in FIG. 13, such steps are examples. That is, the present invention is well suited to performing various other steps or variations of the steps recited in FIG. 13.

In block 1302, an input voltage, e.g., the rectified AC voltage V_IN, and an input current, e.g., the rectified AC current I_IN, are received. In block 1304, the input voltage is converted to an output voltage to power a load, e.g., an LED light source. In block 1306, a current flowing through an energy storage element, e.g., the energy storage element 214, is controlled according to a driving signal, e.g., the driving signal 962, so as to regulate a current through said LED light source.

In block 1308, a first sensing signal, e.g., IAVG, indicating the current through said LED light source is received. In one embodiment, the first sense signal is generated by filtering a second sense signal indicating the current through the energy storage element. In block 1310, a saw-tooth signal is generated based on the driving signal.

In block 1312, the driving signal is controlled based on signals including the saw-tooth signal and the first sense signal to adjust the current through the LED light source to a target level and to correct a power factor of the driving circuit by controlling an average current of the input current to be substantially in phase with the input voltage. In one embodiment, an error signal indicating a difference between the first sense signal and a reference signal indicating the target level of the current through the LED light source is generated. The saw-tooth signal is compared to the error signal. A detection signal indicating an electric condition of the energy storage element is received. The driving signal is switched to a first state if the detection signal indicates that the current through the energy storage element decreases to a predetermined level and is switched to a second state according to a result of the comparison of the saw-tooth signal and the error signal. The current through the energy storage element is increased when the driving signal is in the first state and is decreased when the driving signal is in the second state. In one embodiment, a time duration for the saw-tooth signal to increase from a predetermined level to the error signal is constant if the current through the LED light source is maintained at the target level.

Embodiments in accordance with the present invention provide a driving circuit for driving a load, e.g., an LED light source. The driving circuit includes a power converter and a controller. The power converter converts an input voltage to an output voltage to power the load. The power converter provides a sense signal indicating a current flowing through the load. The driving circuit further includes a saw-tooth signal generator for generating a saw-tooth signal according to the driving signal. Advantageously, the controller generates a driving signal according to signals including the sense signal and the saw-tooth signal. The driving signal controls the current through the energy storage element, which further adjusts the current through the load to a target current level and corrects a power factor by controlling an AC input current to be substantially in phase with an AC input voltage of the driving circuit.

FIG. 14 shows an example for a schematic diagram of a driving circuit 1400, in accordance with one embodiment of the present invention. Elements labeled the same as in FIG. 2 and FIG. 3 have similar functions. In the example of FIG. 14, the driving circuit 1400 includes a rectifier 204, a power converter 1406, a filter 212, and a controller 1410. By way of example, the rectifier 204 is a bridge rectifier which includes diodes D1˜D4. The rectifier 204 rectifies an AC voltage from the power source 202. The power converter 1406 receives the rectified voltage from the rectifier 204 and provides output power for powering a load, e.g., an LED string 208.

In the example of FIG. 14, the power converter 1406 is a buck-boost converter, which receives an input voltage and generates an output voltage which can be greater or less than the input voltage. By using the buck-boost converter, the driving circuit 1400 can be more flexible to regulate the output voltage according to different load requirements. Furthermore, the driving circuit 1400 with the buck-boost converter has a relatively low total harmonic distortion and a relatively high power factor.

In one embodiment, the power converter 1406 includes a capacitor 1408, a switch 1416, a resistor 1420, an energy storage element 1414, a current sensor 1418 (e.g., a resistor), a diode 1412, and a capacitor 1424. The power converter 1406 receives an input voltage and an input current and powers the LED string 208. The switch 1416 is controlled by a driving signal. The controller 1410 receives a sensing signal IAVG indicating a current through the LED string 208 and generates the driving signal based on the sensing signal IAVG to control the switch 1416 and to adjust the current through the LED string 208.

More specifically, the energy storage element 1414 is coupled between the switch 1416 and a ground of the driving circuit 1400. The energy storage element 1414 is also coupled to a common node 1433 between the switch 1416 and the current sensor 1418. The common node 1433 provides a reference ground of the controller 1410. In one embodiment, the reference ground of the controller 1410 is different from the ground of the driving circuit 1400. In the example of FIG. 14, the energy storage element 1414 includes inductors 1402 and 1404. The inductor 1402 is coupled between the reference ground of the controller 1410 and the ground of the driving circuit 1400. The current of the energy storage element 1414 flows through the inductor 1402. The inductor 1404 electrically and magnetically coupled to the inductor 1402 is operable for sensing an electrical condition of the inductor 1402. More specifically, the inductor 1402 and the inductor 1404 are electrically coupled to the common node 1433.

The current of the energy storage element 1414 is controlled by the switch 1416. The resistor 1420, coupled between the switch 1416 and the energy storage element 1414, is operable for providing a sensing signal VSEN to the controller 1410, which indicates a status of the energy storage element 1414. The controller 1410 turns off the switch 1416 if the voltage of the sensing signal VSEN is greater than a predetermined voltage level (e.g. 1.1 V).

The current sensor 1418 has one end coupled to a node 1433, and the other end coupled to the diode 1412. The current sensor 1418 provides a sensing signal ISEN indicating an instant current flowing through the power converter 1406, for example, indicating an instant current flowing through the diode 1412 when the switch 1416 is off. When the switch 1416 is on, no current flows through the diode 1412 because the diode 1412 is reverse-biased. The sensing signal IAVG indicating the current through the LED string 208 is derived from the sensing signal ISEN. More specifically, the filter 212, coupled between the current sensor 1418 and the controller 1410, generates the sensing signal IAVG indicating the current through the LED string 208 based on the sensing signal ISEN. In one embodiment, the filter 212 includes a resistor 320 and a capacitor 322. In the example of FIG. 14, the sensing signal ISEN indicates an instant current flowing through the power converter 1406, e.g., an instant current flowing through the diode 1412. An average current flowing through the diode 1412 is substantially equal to the current through the LED string 208. However, in other alternative embodiments, the sensing signal ISEN may indicate an instant current flowing through other components of the buck-boost converter, and is not limited to the example shown in FIG. 14.

The controller 1410 receives the sensing signal IAVG and controls an average current flowing through the diode 1412 to a target current level by turning the switch 1416 on and off. A capacitor 1424 absorbs ripple current flowing through the LED string 208 such that the current flowing through the LED string 208 is smoothed and substantially equal to the average current flowing through the diode 1412. As such, the current flowing through the LED string 208 can have a level that is substantially equal to the target current level. As used herein, “substantially equal to the target current level” means that the current flowing through the LED string 208 may be slightly different from the target current level but within a range such that the current ripple caused by the non-ideality of the circuit components can be neglected.

In the example of FIG. 14, the controller 1410 has terminals ZCD, GND, DRV, VDD, CS, COMP and FB. The terminal FB is coupled to the current sensor 1418 through the filter 212 and is operable for receiving the sensing signal IAVG which indicates an average current flowing through the diode 1412. The average current flowing through the diode 1412 is substantially equal to the current through the LED string 208. As such, the terminal FB of controller 1410, coupled to the power converter 1406, is operable for receiving the sensing signal IAVG indicating the current flowing through the LED string 208. The terminal ZCD is coupled to the inductor 1404 for receiving a detection signal AUX indicating an electrical condition of the energy storage element 1414, for example, whether the current flowing through the inductor 1402 decreases to a first predetermined current level (e.g., zero ampere). The current of the energy storage element 1414 is controlled by the switch 1416. The controller 1410 turns on the switch 1416 if the current of the detection signal AUX decreases to the first predetermined current level (e.g., zero ampere). The detection signal AUX can also indicate whether the LED string 208 is in an open circuit condition. The terminal DRV is coupled to the switch 1416 and generates a driving signal, e.g., a pulse-width modulation signal PWM1, based on the sensing signal IAVG and the detection signal AUX. The pulse-width modulation signal PWM1 controls the instant current flowing through the power converter 1406, e.g., the current flowing through the diode 1412, so as to adjust the current through the LED string 208. In one embodiment, the pulse-width modulation signal PWM1 has a first state (e.g., logic 1) and a second state (e.g., logic 0). The switch 1416 is turned on if the pulse-width modulation signal PWM1 is in the first state, and is turned off if the pulse-width modulation signal PWM1 is in the second state. The current flowing through the inductor 1402 increases when the driving signal is in the first state, and decreases when the driving signal is in the second state. The terminal VDD is coupled to the inductor 1404 for receiving power from the inductor 1404. The terminal CS is coupled to the resistor 1420 and is operable for receiving the sensing signal VSEN indicating a status of the energy storage element 1414, for example, whether the energy stored in the energy storage element 1414 increases to a predetermined energy level. The sensing signal VSEN can also indicate whether the LED string 208 is in a short circuit condition. The terminal COMP is coupled to the reference ground of the controller 1410 through a capacitor 318. The terminal COMP provides an error signal. In the example of FIG. 14, the terminal GND, that is, the reference ground for the controller 1410, is coupled to the common node 1433 between the current sensor 1418, the inductor 1402, and the inductor 1404.

The switch 1416 can be an N channel metal oxide semiconductor field effect transistor (NMOSFET). The conductance status of the switch 1416 is determined based on a difference between the gate voltage of the switch 1416 and the voltage at the terminal GND (the voltage at the common node 1433). Therefore, the switch 1416 is turned on and turned off depending upon the pulse-width modulation signal PWM1 from the terminal DRV. When the switch 1416 is on, the reference ground of the controller 1410 is higher than the ground of the driving circuit 1400, making the invention suitable for power sources having relatively high voltages.

In operation, when the switch 1416 is turned on, a current flows through the switch 1416, the resistor 1420, the inductor 1402, to the ground of the driving circuit 1400. When the switch 1416 is turned off, a current flows through the inductor 1402, the LED string 208, the diode 1412, and the current sensor 1418. The current sensor 1418 provides the sensing signal ISEN indicating an instant current flowing through the diode 1412. The sensing signal IAVG indicating the current through the LED string 208 is derived from the sensing signal ISEN. Therefore, the controller 1410 controls the switch 1416 by a pulse-width modulation signal PWM1 according to the sensing signal IAVG so as to control an average current flowing through the diode 1412 to a target current level, in one embodiment. As such, the current flowing through the LED string 208, which is filtered by the capacitor 1424, can also be substantially equal to the target current level.

In one embodiment, the controller 1410 determines whether the LED string 208 is in an open circuit condition based on the detection signal AUX. If the LED string 208 is open, the voltage across the capacitor 1424 increases. When the switch 1416 is off, the voltage across the inductor 1402 increases and the voltage of the detection signal AUX increases accordingly. As a result, the current flowing through the terminal ZCD into the controller 1410 increases. Therefore, the controller 1410 monitors the detection signal AUX and if the current flowing through the inductor 1402 increases to a second predetermined current level (e.g., 300 uA) when the switch 1416 is off, the controller 1410 determines that the LED string 208 is in an open circuit condition.

In one embodiment, the controller 1410 determines whether the LED string 208 is in a short circuit condition based on the sensing signal VSEN. If the LED string 208 is in a short circuit condition, the energy stored in the energy storage element 1414 increases and the voltage of the sensing signal VSEN increases accordingly. As a result, the voltage at the terminal CS increases. Therefore, the controller 1410 monitors the sensing signal VSEN and if the voltage of the sensing signal VSEN is greater than a predetermined voltage level (e.g. 1.1 V), the controller 1410 determines that the LED string is in a short circuit condition.

FIG. 15 shows an example of the controller 1410 in FIG. 14, in accordance with one embodiment of the present invention. Elements labeled the same as in FIG. 4 have similar functions. FIG. 15 is described in combination with FIG. 14.

In the example of FIG. 15, the controller 1410 includes an error amplifier 402, a comparator 404, and a pulse-width modulation signal generator 408. The error amplifier 402 generates an error signal VEA at terminal COMP based on the sensing signal IAVG and a reference signal SET indicative of a target current level. The sensing signal IAVG is received at the terminal FB and can indicate an average current flowing through the diode 1412. The error signal VEA is used to adjust the average current flowing through the diode 1412 to the target current level. The comparator 404 is coupled to the error amplifier 402 and compares the error signal VEA with the signal VSEN. The signal VSEN is received at the terminal CS and indicates a status of the energy storage element 1414. The detection signal AUX is received at the terminal ZCD and indicates whether the current flowing through the inductor 1402 decreases to a first predetermined current level, e.g., zero ampere. The pulse-width modulation signal generator 408, coupled to the error amplifier 402 and the comparator 404, can generate a pulse-width modulation signal PWM1 based on the error signal VEA and the detection signal AUX. The pulse-width modulation signal PWM1 is applied to the switch 1416 via the terminal DRV to control a conductance status of the switch 1416.

In operation, the switch 1416 is on when the pulse-width modulation signal PWM1 has a first state (e.g., logic 1). When the switch 1416 is turned on, a current flows through the switch 1416, the resistor 1420, the inductor 1402, to the ground of the driving circuit 1400. The current flowing through the inductor 1402 increases such that the voltage of the sensing signal VSEN increases. The detection signal AUX has a negative voltage level when the switch 1416 is turned on, in one embodiment. The comparator 404 in the controller 1410 compares the error VEA with the signal VSEN. When the voltage of the signal VSEN increases above the voltage of the error signal VEA, the output of the comparator 404 is changed to logic 0. The pulse-width modulation signal generator 408 generates the pulse-width modulation signal PWM1 having a second state (e.g., logic 0) in response to a negative-going edge of the output of the comparator 404 to turn off the switch 1416. The detection signal AUX has a positive voltage level when the switch 1416 is turned off, in one embodiment. When the switch 1416 is turned off, a current flows through the inductor 1402, the LED string 208, the diode 1412, and the current sensor 1418. The current flowing through the inductor 1402 decreases such that the voltage of the signal VSEN decreases. The pulse-width modulation signal PWM1 is switched to the first state (e.g., logic 1) if the detection signal AUX indicates that the current through the inductor 1402 decreases to a first predetermined current level (e.g., zero ampere). More specifically, when the current flowing through the inductor 1402 decreases to the first predetermined current level (e.g., zero ampere), a negative-going edge occurs to the voltage of the detection signal AUX. Upon receiving a negative-going edge of the detection signal AUX, the pulse-width modulation signal generator 408 generates the pulse-width modulation signal PWM1 having the first state (e.g., logic 1) to turn on the switch 1416.

In one embodiment, the pulse-width modulation signal PWM1 remains at the second state (e.g., logic 0) if the detection signal AUX indicates that the current through the inductor 1402 increases to a second predetermined current level (e.g., 300 uA) when the switch 1416 is off. The controller 1410 determines that the LED string 208 is in an open circuit condition. In one embodiment, if the voltage of the sensing signal VSEN is greater than a predetermined voltage level (e.g., 1.1 V), the controller 1410 determines that the LED string is in a short circuit condition. When the controller 1410 determines that the LED string is in an open circuit condition or a short circuit condition, the pulse-width modulation signal PWM1 remains at the second state (e.g., logic 0) to turn off the switch 1416 until such abnormal condition no longer exists.

In one embodiment, a duty cycle of the pulse-width modulation signal PWM1 is determined by the error signal VEA. If the voltage of the sensing signal IAVG is less than the voltage of the reference signal SET, the error amplifier 402 increases the voltage of the error signal VEA so as to increase the duty cycle of the pulse-width modulation signal PWM1. Accordingly, the average current flowing through the diode 1412 increases until the voltage of the sensing signal IAVG reaches the voltage of the reference signal SET. If the voltage of the sensing signal IAVG is greater than the voltage of the reference signal SET, the error amplifier 402 decreases the voltage of the error signal VEA so as to decrease the duty cycle of the pulse-width modulation signal PWM1. Accordingly, the average current flowing through the diode 1412 decreases until the voltage of the sensing signal IAVG drops to the voltage of the reference signal SET. As such, the average current flowing through the diode 1412 can be maintained to be substantially equal to the target current level.

FIG. 16 shows another example for a schematic diagram of a driving circuit 1600, in accordance with one embodiment of the present invention. Elements labeled the same as in FIG. 14 have similar functions. The schematic diagram of the light source driving circuit 1600 in FIG. 16 is similar to the schematic diagram of the light source driving circuit 1400 in FIG. 14 except for the configuration of the power converter 1406. In the example of FIG. 16, the energy storage element 1414 includes the inductor 1402. In one embodiment, the power converter 1406 can further include a Zener diode D5 coupled between the inductor 1402 and the controller 1410. The Zener diode D5 forms a bias voltage level shifter which applies a level shift (voltage bias) to the power supply voltage of the controller 1410 so as to provide proper power from the inductor 1402 to the controller 1410 via the terminal VDD.

FIG. 17 shows an example for a schematic diagram of a driving circuit 1700, in accordance with one embodiment of the present invention. Elements labeled the same as in FIG. 9A, FIG. 10 and FIG. 14 have similar functions. The schematic diagram of the light source driving circuit 1700 in FIG. 17 is similar to the schematic diagram of the light source driving circuit 1000 in FIG. 10 except for the configuration of the power converter 1406.

In one embodiment, the power converter 1406 includes a capacitor 1408 coupled to the power line 912. The capacitor 1408 reduces ripples of the rectified AC voltage V_IN to smooth the waveform of the rectified AC voltage V_IN. In one embodiment, the capacitor 1408 has a relatively small capacitance to help eliminate or reduce distortion of the rectified AC voltage V_IN. Moreover, in one embodiment, a current flowing through the capacitor 1408 can be ignored due to the relatively small capacitance. Thus, the current flowing through the switch 1416 when the switch 1416 is on is approximately equal to the rectified AC current I_IN from the rectifier 204.

The power converter 1406 in FIG. 17 operates similarly as the power converter 1406 in FIG. 14. In one embodiment, a current I₁₄₁₂ flows through the diode 1412 and a current I₁₄₀₂ flows through the inductor 1402 according to the conductance status of the switch 1416. More specifically, the controller 910 generates the driving signal 962, e.g., a PWM signal, through the terminal DRV to switch the switch 1416 to an ON state or an OFF state. When the switch 1416 is turned on, the current I₁₄₀₂ flows through the switch 1416, the resistor 1420, the inductor 1402, to the ground of the driving circuit 1700. No current flows through the diode 1412 because the diode 1412 is reverse-biased. The current I₁₄₀₂ increases during the ON state of the switch 1416 according to equation (3):

ΔI₁₄₀₂=V_IN*T_ON/L₁₄₀₂,   (3)

where T_ON represents a time duration when the switch 1416 is turned on, ΔI₁₄₀₂ represents a change of the current I₁₄₀₂, L₁₄₀₂ represents the inductance of the inductor 1402, and the voltage drops across the switch 1416 and the resistor 1420 are ignored. In one embodiment, the controller 910 controls the driving signal 962 to maintain the time duration T_ON constant during each switching cycle of the switch 1416. Therefore, the change ΔI₁₄₀₂ of the current I₁₄₀₂ during the time T_ON is proportional to the rectified AC voltage V_IN. In one embodiment, the switch 1416 is turned on when the current I₁₄₀₂ decreases to a first predetermined current level, e.g., zero ampere. Accordingly, the peak level of the current I₁₄₀₂ is proportional to the rectified AC voltage V_IN.

In each switching cycle, the switch 1416 is turned off after being turned on for a time period of T_ON. If the switch 1416 is turned off, a current flows through the inductor 1402, the LED string 208, the diode 1412, and the current sensor 1418. Accordingly, the current I₁₄₁₂ decreases according to equation (4):

ΔI₁₄₁₂=ΔI₁₄₀₂=V_OUT*T_OFF/L₁₄₀₂.   (4)

where T_OFF represents a time duration when the switch 1416 is turned off, ΔI₁₄₁₂ represents a change of the current I₁₄₁₂, and the voltage drops across the diode 1412 and the current sensor 1418 are ignored. The rectified AC current I_IN is substantially equal to the current I₁₄₀₂ during an ON state of the switch 1416 and equal to zero ampere during an OFF state of the switch 1416, in one embodiment.

In one embodiment, the power converter 1406 includes a capacitor 1424. The capacitor 1424 can be a capacitor having a relatively large capacitance. As such, the current I_OUT through the LED string 208 represents an average level of the current I₁₄₁₂.

The controller 910 in FIG. 17 operates similarly as the controller 910 in FIG. 10. In the example of FIG. 17, the controller 910 has terminals ZCD, GND, DRV, VDD, CS, COMP and FB. The terminal ZCD is coupled to the inductor 1404 for receiving a detection signal AUX indicating an electrical condition of the inductor 1402, for example, whether the current flowing through the inductor 1402 decreases to a first predetermined current level(e.g., zero ampere). The detection signal AUX can also indicate whether the LED string 208 is in an open circuit condition. The terminal GND is coupled to the common node 1433 between the current sensor 1418, the inductor 1402, and the inductor 1404. The terminal DRV is coupled to the switch 1416 and generates a driving signal 962, e.g., a PWM signal, to turn the switch 1416 on and off. The terminal VDD is coupled to the inductor 1404 for receiving power from the inductor 1404. The terminal COMP is coupled to the reference ground of the controller 910 through a capacitor 318. The terminal FB is coupled to the current sensor 1418 through the filter 212 and is operable for receiving the sensing signal IAVG which indicates the current I_OUT through the LED string 208.

The saw-tooth signal generator 902 coupled to the controller 910 is operable for generating a saw-tooth signal 960 at the CS terminal based on the driving signal 962 at the DRV terminal. By way of example, the saw-tooth signal generator 902 includes a resistor 1016 and a diode 1018 coupled in parallel between the terminal DRV and the terminal CS, and further includes a resistor 1012 and a capacitor 1014 coupled in parallel between the CS terminal and ground. The saw-tooth signal 960 varies according to the driving signal 962. More specifically, in one embodiment, the driving signal 962 is a PWM signal. When the driving signal 962 is logic 1, a current I1 flows from the DRV terminal through the resistor 1016 to the capacitor 1014. Thus, the capacitor 1014 is charged and a voltage V₉₆₀ of the saw-tooth signal 960 increases. When the driving signal 962 is logic 0, a current I2 flows from the capacitor 1014 through the diode 1018 to the DRV terminal. Thus, the capacitor 1014 is discharged and the voltage V₉₆₀ decreases to zero volts. The saw-tooth signal generator 902 can include other components and is not limited to the example shown in FIG. 17.

Advantageously, the controller 910 generates the driving signal 962 based on the saw-tooth signal 960 and the sensing signal IAVG. The controller 910 adjusts the current I_OUT through the LED string 208 to a target current level and corrects a power factor of the driving circuit 1700 by controlling an average current I_IN—AVG of the rectified AC current I_IN to be substantially in phase with the input voltage V_IN.

FIG. 18 illustrates a waveform of signals generated or received by a driving circuit, e.g., the driving circuit 1700, in accordance with one embodiment of the present invention. FIG. 18 is described in relation to FIG. 4, FIG. 9A, FIG. 9B, and FIG. 17. FIG. 18 shows the rectified AC voltage V_IN, the rectified AC current I_IN, the average current I_IN—AVG of the rectified AC current I_IN, the current I₁₄₀₂ flowing through the inductor 1402, the current I_OUT flowing through the LED string 208, the sensing signal ISEN indicating the current I₁₄₁₂ flowing through the diode 1412, the error signal VEA, the saw-tooth signal 960, and the driving signal 962. The driving circuit 1700 with the buck-boost converter has a relatively low total harmonic distortion and a relatively high power factor.

As shown in the example of FIG. 18, the rectified AC voltage V_IN is a rectified sinusoidal waveform. At time t1, the driving signal 962 is changed to logic 1. Thus, the switch 1416 is turned on and the current I₁₄₀₂ flowing through the inductor 1402 increases. There is no current flowing through the diode 1412 because the diode 1412 is reverse-biased. Meanwhile, the saw-tooth signal 960 increases during the first state (e.g., logic 1) of the driving signal 962.

At time t2, when the saw-tooth signal 960 reaches the error signal VEA, the driving signal 962 is switched to the second state (e.g., logic 0). In response to the negative-going edge of the driving signal 962, the saw-tooth signal 960 drops to zero volts and the sensing signal ISEN increases to the peak level of the current I₁₄₀₂. The driving signal 962 turns off the switch 1416 and the current starts to flow through the inductor 1402 and the diode 1412, thereby decreasing the current I₁₄₀₂ and the sensing signal ISEN. In other words, the saw-tooth signal 960 and the error signal VEA determine the time period T_ON when the driving signal 962 is logic 1 to turn on the switch 1416.

At time t3, the current I₁₄₀₂ and the current I₁₄₁₂ decreases to the first predetermined current level, e.g., zero ampere. Thus, the controller 910 adjusts the driving signal 962 to logic 1 to turn on the switch 1416.

In one embodiment, the current I_OUT flowing through the LED string 208 is equal to or proportional to an average level of the current I₁₄₁₂ over a cycle period of the input voltage V_IN. As described in relation to FIG. 11, the current I_OUT is adjusted to the target current level which is determined by the reference signal SET. In addition, as shown in FIG. 18, the sensing signal ISEN indicating the current I₁₄₁₂ between t1 and t4 has same waveforms as those between t5 and t6. Thus, the average level of the current I₁₄₁₂ between t1 and t4 is equal to the average level of the current I₁₄₁₂ between t5 and t6. Accordingly, the current I_OUT is maintained at the target level. In one embodiment, the time period T_ON is determined by the saw-tooth signal 960 and the error signal VEA. In one embodiment, the time period T_ON is constant because the time period for the saw-tooth signal 960 to rise from zero volts to the error signal VEA is the same in each cycle of the driving signal 962. Based on equation (3), the change ΔI₁₄₀₂ of the current I₁₄₀₂ during the time period T_ON is proportional to the rectified AC voltage V_IN. Therefore, the peak level of the sensing signal ISEN (i.e., the peak level of the current I₁₄₀₂) is proportional to the rectified AC voltage V_IN as shown in FIG. 18.

The rectified AC current I_IN has a waveform similar to the waveform of the current I₁₄₀₂ when the switch 1416 is turned on, and is substantially equal to zero ampere when the switch 1416 is turned off, in one embodiment. The average current I_IN—AVG is approximately in phase with the rectified AC voltage V_IN between time t1 and t6. As described in relation to FIG. 9B, the controller 910 corrects the power factor of the driving circuit 1700 such that the AC input current I_AC is approximately in phase with the AC input voltage V_AC.

While the foregoing description and drawings represent embodiments of the present invention, it will be understood that various additions, modifications and substitutions may be made therein without departing from the spirit and scope of the principles of the present invention as defined in the accompanying claims. One skilled in the art will appreciate that the invention may be used with many modifications of form, structure, arrangement, proportions, materials, elements, and components and otherwise, used in the practice of the invention, which are particularly adapted to specific environments and operative requirements without departing from the principles of the present invention. The presently disclosed embodiments are therefore to be considered in all respects as illustrative and not restrictive, the scope of the invention being indicated by the appended claims and their legal equivalents, and not limited to the foregoing description.

Claims

1. A driving circuit for driving a light-emitting diode (LED) light source, said circuit comprising: a buck-boost converter that receives an input voltage and an input current and powers said LED light source, and that comprises a switch controlled by a driving signal; anda controller, coupled to said buck-boost converter, that receives a first signal indicating a current through said LED light source, and that generates said driving signal based on said first signal to control said switch and to adjust said current through said LED light source,wherein said buck-boost converter further comprises a current sensor coupled to said switch, wherein said current sensor provides a second signal indicating an instant current flowing through said buck-boost converter, wherein said first signal is derived from said second signal, and wherein a reference ground of said controller is different from a ground of said driving circuit.

2. The driving circuit of claim 1, wherein said buck-boost converter further comprises an energy storage element coupled between said switch and said ground of said driving circuit, wherein a current of said energy storage element is controlled by said switch, wherein said energy storage element is coupled to a common node between said switch and said current sensor, and wherein said common node provides said reference ground of said controller.

3. The driving circuit of claim 2, wherein said buck-boost converter further comprises a resistor, coupled between said switch and said energy storage element, that provides a voltage sensing signal to said controller, wherein said voltage sensing signal indicates a status of said energy storage element, and wherein said controller turns off said switch if a voltage of said voltage sensing signal is greater than a predetermined voltage level.

4. The driving circuit of claim 2, wherein said energy storage element comprises: a first inductor coupled between said reference ground of said controller and said ground of said driving circuit, wherein said current of said energy storage element flows through said first inductor; anda second inductor, electrically and magnetically coupled to said first inductor, that senses an electrical condition of said first inductor.

5. The driving circuit of claim 2, wherein said energy storage element comprises a first inductor coupled between said reference ground of said controller and said ground of said driving circuit, wherein said current of said energy storage element flows through said first inductor, and wherein said buck-boost converter further comprises a Zener diode coupled between said first inductor and said controller.

6. The driving circuit of claim 2, wherein said controller further receives a detection signal indicating an electrical condition of said energy storage element, wherein said driving signal has a first state and a second state, wherein said current through said energy storage element increases when said driving signal is in said first state, and decreases when said driving signal is in said second state, wherein said driving signal is switched to said first state if said detection signal indicates that said current through said energy storage element decreases to a first predetermined current level, and wherein said driving signal remains at said second state if said detection signal indicates that said current through said energy storage element increases to a second predetermined current level when said switch is off.

7. The driving circuit of claim 1, further comprising: a filter, coupled between said current sensor and said controller, that generates said first signal based on said second signal, wherein said instant current flowing through said buck-boost converter comprises an instant current flowing through a diode of said buck-boost converter, and wherein an average current flowing through said diode is substantially equal to said current through said LED light source; andan error amplifier that generates an error signal based on said first signal and a reference signal indicative of a target current level.

8. The driving circuit of claim 7, further comprising: a saw-tooth signal generator, coupled to said controller, that generates a saw-tooth signal based on said driving signal,wherein said controller generates said driving signal based on said saw-tooth signal and said error signal to adjust said current through said LED light source to said target current level and to correct a power factor of said driving circuit by controlling an average current of said input current to be substantially in phase with said input voltage.

9. The driving circuit of claim 8, wherein said driving signal has a first state and a second state, wherein said saw-tooth signal increases during said first state of said driving signal, and wherein when said saw-tooth signal reaches said error signal, said driving signal is switched to said second state.

10. The driving circuit of claim 8, wherein a time duration for said saw-tooth signal to increase from a predetermined level to said error signal is constant if said current through said LED light source is maintained at said target level.

11. The driving circuit of claim 8, wherein said saw-tooth signal generator comprises: a diode and a first resistor coupled in parallel between a first node and a second node; anda capacitor and a second resistor coupled in parallel between said second node and said reference ground of said controller, wherein said first node receives said driving signal, and said second node provides said saw-tooth signal.

12. The driving circuit of claim 1, further comprising: a rectifier that receives an alternating current (AC) input voltage and an AC input current and provides said input voltage and said input current,wherein said controller corrects a power factor of said driving circuit such that said AC input current is substantially in phase with said AC input voltage.

13. A controller for controlling a buck-boost converter that receives an input voltage and an input current and powers a light-emitting diode (LED) light source, said controller comprising: a first sensing pin that receives a first signal indicating a current flowing through said LED light source;a detection pin that receives a detection signal indicating an electrical condition of an energy storage element in said buck-boost converter, wherein a current of said energy storage element is controlled by a switch, and wherein said controller turns on said switch if a current of said detection signal decreases to a predetermined current level; anda driving pin that provides a driving signal to said switch based on said first signal and said detection signal, to control an instant current flowing through said buck-boost converter so as to adjust said current flowing through said LED light source,wherein said first signal is derived from a second signal indicating said instant current flowing through said buck-boost converter.

14. The controller of claim 13, further comprising: a compensation pin providing an error signal;wherein said driving signal has a first state and a second state, wherein said current through said energy storage element increases when said driving signal is in said first state, and decreases when said driving signal is in said second state.

15. The controller of claim 14, further comprising: an error amplifier generating said error signal at said compensation pin based on said first signal and a reference signal indicative of a target current level.

16. The controller of claim 15, further comprising: a pulse-width modulation signal generator, coupled to said error amplifier, that generates said driving signal based on said error signal and said detection signal.

17. The controller of claim 13, wherein said controller further receives a saw-tooth signal that varies according to said driving signal, and wherein said controller generates said driving signal based on said first signal and said saw-tooth signal to adjust said current through said LED light source to a target current level and to control an average current of said input current to be approximately in phase with said input voltage.

18. The controller of claim 17, wherein said driving signal has a first state and a second state, wherein said saw-tooth signal increases during said first state of said driving signal, wherein when said saw-tooth signal reaches an error signal, said driving signal is switched to said second state, and wherein said error signal is generated based on said first signal and a reference signal indicating a target current level.

19. The controller of claim 18, wherein a time duration for said saw-tooth signal to increase from a predetermined level to said error signal is constant if said current through said LED light source is maintained at said target level.

20. The controller of claim 13, wherein said controller further receives a voltage sensing signal that indicates a status of said energy storage element, and wherein said controller turns off said switch if a voltage of said voltage sensing signal is greater than a predetermined voltage level.