Complementary metal oxide semiconductor class AB amplifier

Abstract

A CMOS class AB amplifier has an adaptive level shift circuit, a compensated capacitor, and an output transistor pair. The adaptive level shift circuit includes a current mirror circuit, a diode transistor, a switch transistor, and a current source transistor. The diode transistor is utilized as a bias for driving the switch transistor so as to provide a relatively low linear resistor for being used as a feedback. Therefore, the Q (quality factor) value of the gyration inductance can be effectively reduced and the occurrence of the peak gain can be suppressed effectively, so as to maintain a desired gain margin.

Description

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to an amplifier circuit, and especially to a complementary metal-oxide semiconductor (CMOS) class AB amplifier.

2. Description of Related Art

FIG. 4 shows a conventional class AB amplifier, which includes an input-stage formed by a differential amplifier 41 , and an output stage formed by an class AB amplifier 42 , so as to perform a differential operation on an input signal, and amplify and drive the same for output.

The AB class amplifier can be analyzed as follows: The class AB amplifier 42 with a loading capacitor C L has an open loop voltage gain, from node A to node O, expressed as: $$ A AO ⁡ ( s ) = V O ⁡ ( s ) V A ⁡ ( s ) = ( gm 2 + gm 1 ⁢ gm 6 ⁢ r B sC B ⁢ r B + 1 ) ⁢ ( r O sC L ⁢ r O + 1 ) , ( 1 )

where r B =r ds4 //r ds5 is a DC resistance on node B; r o =r ds2 //r ds6 is an DC output resistance of the output stage; gm j represents transfer-conductance of a MOS transistor M j , j=1,2,3 . . . ; C B =C DB4 +C DB5 +C GD4 +C GD5 +gm 6 r O C GD6 is a parasitic capacitance of the output MOS transistor M 6 and the current mirror amplifier consisting of MOS transistors M 4 and M 5 .

When the class AB amplifier employs in an operational amplifier with a compensation capacitor C F , its output impedance with a loading capacitor C L becomes: $$ y O ⁡ ( s ) = 1 / z O ⁡ ( s ) = sC F ⁢ r A + 1 r A + sC F ⁢ r A sC F ⁢ r A + 1 ⁢ ( gm 2 + gm 1 ⁢ gm 6 ⁢ r B sC B ⁢ r B + 1 ) + 1 r O + sC L , ( 2 )

where r A is a DC resistance on node point A, i.e., an output resistance of the differential amplifier. If ω>>1/(r A C F ), and

$$ gm 2 ⪢ 1 r A + 1 r O ,

we have $$ y OH ⁡ ( s ) = 1 / z OH ⁡ ( s ) = gm 2 + gm 1 ⁢ gm 6 ⁢ r B ⁢ 1 sC B ⁢ r B + 1 + sC L , ( 3 )

herein, the term $$ gm 1 ⁢ gm 6 ⁢ r B ⁢ 1 sC B ⁢ r B + 1

represents a gyrated output inductor,

where,

L OEQ ≈C B /( gm 1 gm 6 ),  (4)

and with an equivalent series resistor:

R SEQ =1/( gm 1 gm 6 r B ).  (5)

Therefore, we have $$ z OH ⁡ ( s ) = sr B ⁢ C B + 1 s 2 ⁢ r B ⁢ C L ⁢ C B + sC L + sgm 2 ⁢ r B ⁢ C B + gm 1 ⁢ gm 6 ⁢ r B + gm 2 , ( 6 )

Due to gm 1 gm 6 r B >>gm 2 , z O (s) resonates at: $$ ω R = gm 1 ⁢ gm 6 ⁢ r B + gm 2 r B ⁢ C B ⁢ C L ≈ 1 r B ⁢ C B ⁢ gm 1 ⁢ r B ⁢ gm 6 C L = gm 1 ⁢ gm 6 C B ⁢ C L = 1 L OEQ ⁢ C L , ( 7 ) Q R = r B C L + gm 2 ⁢ r B ⁢ C B ⁢ gm 1 ⁢ gm 6 ⁢ C B ⁢ C L = gm 2 ⁢ r B ⁢ C B ⁢ C L C L + gm 2 ⁢ r B ⁢ C B ⁢ gm 1 ⁢ r B ⁢ gm 6 gm 2 , ( 8 )

and has a resonated impedance: $$ &LeftBracketingBar; z OH ⁡ ( j ⁢   ⁢ ω R ) &RightBracketingBar; = ( 1 C L + gm 2 ⁢ r B ⁢ C B ) ⁢ C B ⁢ C L gm 1 ⁢ gm 6 ⁢ gm 1 ⁢ gm 6 ⁢ r B 2 ⁢ C B C L + 1 . ( 9 ) $$ If ⁢   ⁢ gm 1 ⁢ gm 6 ⁢ r B 2 ⁢ C B C L ⪢ 1 ,

we have: $$ &LeftBracketingBar; z OH ⁡ ( j ⁢   ⁢ ω R ) &RightBracketingBar; ≈ r B ⁢ C B C L + gm 2 ⁢ r B ⁢ C B , ( 10 )

if gm 2 r B C B >>C L , we have $$ &LeftBracketingBar; z OH ⁡ ( j ⁢   ⁢ ω R ) &RightBracketingBar; ≈ 1 gm 2 . ( 11 )

From the above analysis, it is known that the compensation capacitor C F in the operational amplifier will cause a gyration effect, so that the internal parasitic capacitance C B =C DB4 +C DB5 +C GD4 +C GD5 +gm 6 r O C GD6 is gyrated to be the inductance in the output impedance. Therefore, the resonated output impedance |Z OH (jω R )| always induces a peak gain on a operational amplifier around several 1 MHz to several 10 MHz. As a result, the gain margin of the operational amplifier is insufficient, or even becomes negative, which will cause a serious problem of oscillating. In general, to avoid this problem, it is necessary to design a very low gain margin for the differential amplifier at the output stage of the operational amplifier. However, such a design will cause various problem such as greatly decreasing the gain bandwidth and increasing the DC offset.

In Japan Patent publication No. 09-01823, there is disclosed an operational amplifier circuit which may alleviate the aforementioned problem by reducing the equivalent value of r B . However, such a circuit will cause a problem of insufficient output driving ability, and thus the improvement is limited.

Therefore, it is desirable for the above conventional circuit to be improved so as to mitigate and/or obviate the aforementioned problems.

SUMMARY OF THE INVENTION

Accordingly, the object of the present invention is to provide a complementary metal oxide semiconductor class AB amplifier for achieving the purposes of saving power and having a large driving capability without sacrificing the gain bandwidth.

In accordance with one aspect of the present invention, there is provided a CMOS class AB amplifier having an adaptive level shift circuit, a compensation capacitor and an output transistor pair to receive an output of a differential amplifying circuit for being provided with a DC bias and a DC amplifying effect by the adaptive level shift circuit, phase compensated by the compensation capacitor, and driven by the output transistor pair. The adaptive level shift circuit comprises: a current mirror circuit having a first NMOS transistor and a second NMOS transistor, each having a gate connected to one another; a NMOS diode transistor having a gate and a drain connected to the gate; a NMOS switch transistor having a source connected to a drain of the second NMOS transistor of the current mirror circuit, a drain connected to a drain of the first NMOS transistor of the current mirror circuit and a source of the NMOS diode transistor, and a gate connected to the gate and drain of the NMOS diode transistor; a PMOS input amplifier transistor having a drain connected to the drain of the NMOS diode transistor; and, a PMOS current source transistor having a drain connected to the source of the NMOS switch transistor.

In accordance with another object of the present invention, there is provided a CMOS class AB amplifier having an adaptive level shift circuit, a compensation capacitor and an output transistor pair to receive an output of a differential amplifying circuit for being provided with a DC bias and a DC amplifying effect by the adaptive level shift circuit, phase compensated by the compensation capacitor, and driven by the output transistor pair. The adaptive level shift circuit comprises: a current mirror circuit having a first PMOS transistor and a second PMOS transistor, each having a gate connected to one another; a PMOS diode transistor having a gate and a drain connected to the gate; a PMOS switch transistor having a source connected to a drain of the second PMOS transistor of the current mirror circuit, a drain connected to a drain of the first PMOS transistor of the current mirror circuit and a source of the PMOS diode transistor, and a gate connected to the gate and drain of the PMOS diode transistor; a NMOS input amplifier transistor having a drain connected to the drain of the PMOS diode transistor; and, a NMOS current source transistor having a drain connected to the source of the PMOS switch transistor.

Other objects, advantages, and novel features of the invention will become more apparent from the following detailed description when taken in conjunction with the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is the circuit diagram of a CMOS AB class amplifier in accordance with a preferred embodiment of the present invention;

FIG. 2 shows a comparison of the phase curve of the present CMOS AB class amplifier with a load capacitor C L of 100 pF and the phase curve of the conventional amplifier;

FIG. 3 is the circuit diagram of a CMOS AB class amplifier in accordance with another preferred embodiment of the present invention; and

FIG. 4 is the circuit diagram of a conventional class AB amplifier.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

To acquire the CMOS AB class amplifier in accordance with the present invention, at first, from the analysis in the description of related art, it is known that C B can be assumed to zero to suppress the resonance, so as to obtain an idea output impedance:

y OH — Ideal =1/ z OH — Ideal =gm 2 +gm 1 gm 6 r B +sC L   (12)

Therefore, the following impedance variation can be derived: $$ z OH ⁡ ( s ) / z OH — ⁢ ideal ⁡ ( s ) =   ⁢ ( sC L + gm 2 + gm 1 ⁢ gm 6 ⁢ r B ) ⁢ ( sr B ⁢ C B + 1 ) s 2 ⁢ r B ⁢ C L ⁢ C B + sC L + sgm 2 ⁢ r B ⁢ C B + gm 1 ⁢ gm 6 ⁢ r B + gm 2 =   ⁢ 1 + sgm 1 ⁢ gm 6 ⁢ r B 2 ⁢ C B s 2 ⁢ r B ⁢ C L ⁢ C B + sC L + sgm 2 ⁢ r B ⁢ C B + gm 1 ⁢ gm 6 ⁢ r B + gm 2 . ( 13 )

Then, we have $$ z OH ⁡ ( j ⁢   ⁢ ω R ) z OH — ⁢ Ideal ⁡ ( j ⁢   ⁢ ω R ) = 1 + gm 1 ⁢ gm 6 ⁢ r B 2 ⁢ C B C L + gm 2 ⁢ r B ⁢ C B , ( 14 )

and if gm 2 r B C B >>C L , the maximum loss of gain margin is obtained: $$ z OH ⁡ ( j ⁢   ⁢ ω R ) z OH — ⁢ Ideal ⁡ ( j ⁢   ⁢ ω R ) = 1 + gm 1 ⁢ gm 6 ⁢ r B gm 2 ≈ gm 1 ⁢ gm 6 ⁢ r B gm 2 . ( 15 )

Referring to equations (14) and (15), it is known that the ideal method to suppress the resonation gain is to reduce the term C B to zero, but this is impossible. Therefore, a practical and preferred way is to reduce the value of r B . To achieve this object, FIG. 1 depicts the CMOS class AB amplifier in accordance the present invention, which includes an adaptive level shift circuit 11 , a compensation capacitor C F , and an output transistor pair 12 . This CMOS class AB amplifier receives an output from a differential amplifying circuit 13 , and the adaptive level shift circuit 11 will provide a DC bias required for amplifying and a DC amplifying effect in line with the differential input I/P. After being phase compensated by the compensation capacitor C F , the output is driven by the output transistor pair 12 .

Referring to FIG. 1 , the adaptive level shift circuit 11 includes a PMOS transistor M 1 , used as an input amplifier, a PMOS transistor M 5 , used as a current source, two NMOS transistors M 3 and M 4 , used as a current mirror, a NMOS transistor M 8 , used as a switch, and a NMOS transistor M 7 , used as a diode. The gate of the transistor M 1 is connected to an output (node A) of the differential amplifying circuit 13 . The gate of the transistor M 5 is connected to a bias terminalVB 1 . The sources of the transistors M 1 and M 5 are connected to a voltage source VDD, and the drains thereof are connected to the drain of the transistor M 7 and the source of the transistor M 8 , respectively. The gates of the transistor M 3 and M 4 are connected together and further connected to the drain of the transistor M 3 . The transistors M 1 , M 3 , M 4 and M 5 are utilized to provide the DC bias and DC amplifying effect to a known level shift circuit.

The output transistor pair 12 is formed by connecting the drain of the PMOS transistor M 2 to the drain of the NMOS transistor M 6 . The gate of the transistor M 2 is connected to the gate of the transistor M 1 . The gate of the transistor M 6 is connected to the source of the transistor M 8 . The compensation capacitor C F is connected between the drain of the transistor M 2 and the gate of the transistor M 1 .

In the CMOS class AB amplifier in accordance with the present invention, the transistor M 7 is connected between the output of the transistor M 1 and the transistor M 3 of the current mirror for providing a DC bias to control the transistor M 8 . As shown in the figure, the transistor M 7 has its gate and drain connected together so as to be formed as a diode. The source of the transistor M 8 is connected to the drain (node B) of the transistor M 4 . The drain of the transistor M 8 is connected to the drain of the transistor M 3 and the source of the transistor M 7 (node C). The gate of the transistor M 8 is connected to the connection point (node D) of the gate and drain of the transistor M 7 .

When the class AB amplifier in accordance with the present invention is working in small signal, the drain-source voltage Vds of the transistor M 8 is relatively small; i.e., the transistor M 8 is operating in a linear region (also known as a triode region), thereby providing a relatively small drain-source resistor r DS7 . Therefore, the output impedance of class AB amplifier is: $$ y O ⁡ ( s ) = 1 / z O ⁡ ( s ) = sC F ⁢ r A + 1 r A + sC F ⁢ r A sC F ⁢ r A + 1 ⁢ ( gm 2 + 1 2 ⁢ gm 1 ⁢ gm 6 ⁢ r DS8 sC B ⁢ r DS8 + 1 ) + 1 r O + sC L , ( 16 )

wherein C B =C GD8 +C DB8 +C DB4 +C DB5 +C GD4 +C GD5 +gm 6 r O C GD6 will gyrate an output inductance:

L OEQ ≈2 C B /( gm 1 gm 6 ),  (17)

which has an equivalent series resistance:

R SEQ =2/( gm 1 gm 6 r DS8 )  (18)

If ω>>1/(r A C F ) and $$ gm 2 ⪢ 1 r A + 1 r O ,

it can be simplified as: $$ z O ⁡ ( s ) = ( sC B ⁢ r DS8 + 1 ) / gm 2 s 2 ⁢ C L ⁢ C B ⁢ r DS8 / gm 2 + sC L / gm 2 + sC B ⁢ r DS8 + gm 1 ⁢ gm 6 ⁢ r DS8 / 2 ⁢ gm 2 + 1 , ( 19 )

and if gm 1 gm 6 r DS8 /2>>gm 2 , the output impedance has a resonation frequency: $$ ω R = gm 1 ⁢ gm 6 ⁢ r DS8 / 2 + gm 2 r DS8 ⁢ C B ⁢ C L ≈ 1 r DS8 ⁢ C B ⁢ gm 1 ⁢ r DS8 ⁢ gm 6 / 2 C L = gm 1 ⁢ gm 6 2 ⁢ C B ⁢ C L = 1 L OEQ ⁢ C L , ( 20 ) Q R = r DS8 C L + gm 2 ⁢ r DS8 ⁢ C B ⁢ gm 1 ⁢ gm 6 ⁢ C B ⁢ C L 2 , ( 21 )

and a resonated impedance: $$ &LeftBracketingBar; z OH ⁡ ( j ⁢   ⁢ ω R ) &RightBracketingBar; = ( 1 C L + gm 2 ⁢ r DS8 ⁢ C B ) ⁢ 2 ⁢ C B ⁢ C L gm 1 ⁢ gm 6 ⁢ gm 1 ⁢ gm 6 ⁢ r DS8 2 ⁢ C B 2 ⁢ C L + 1 . ( 22 )

Comparing with the ideal condition (C B approaches to zero to completely suppress the resonation) having an output impedance:

y OH — Ideal =1 /z OH — Ideal =gm 2 +gm 1 gm 6 r B +sC L ,  (23)

the following impedance variation can be derived: $$ z OH ⁡ ( s ) / z OH — ⁢ Ideal ⁡ ( s ) = 1 + sgm 1 ⁢ gm 6 ⁢ r DS8 2 ⁢ C B / 2 s 2 ⁢ r DS8 ⁢ C L ⁢ C B + sC L + sgm 2 ⁢ r DS8 ⁢ C B + gm 1 ⁢ gm 6 ⁢ r DS8 / 2 + gm 2 , ( 24 )

and therefore we have: $$ z OH ⁡ ( j ⁢   ⁢ ω R ) z OH — ⁢ Ideal ⁡ ( j ⁢   ⁢ ω R ) = 1 + 1 2 ⁢ gm 1 ⁢ gm 6 ⁢ r DS8 2 ⁢ C B C L + gm 2 ⁢ r DS8 ⁢ C B . ( 25 )

If gm 2 r DS8 C B >>C L , the maximum loss of gain margin can be obtained: $$ z OH ⁡ ( j ⁢   ⁢ ω R ) z OH — ⁢ Ideal ⁡ ( j ⁢   ⁢ ω R ) = 1 + gm 1 ⁢ gm 6 ⁢ r DS8 2 ⁢ gm 2 ≈ gm 1 ⁢ gm 6 ⁢ r DS8 2 ⁢ gm 2 . ( 26 )

Referring to the equations (14), (15), (25) and (26), it is known that the r DS8 of the class AB amplifier in -accordance with the present invention is a resistor operating in a linear region, and the resistance value if about 110 K Ω. Since the r DS8 is farther small than r B , the gain peak of the class AB amplifier in accordance with the present invention can be suppressed effectively.

When the class AB amplifier in accordance with the present invention is working in large signal to output a large source current, the node A at the input terminal will be pulled low. The transistor M 2 will output a large source current, and the transistor M 1 will conduct a transient current larger than a static current to flow through the transistor M 7 and transistor M 3 . Since the transistors M 4 and M 3 are formed as a current mirror, the transistor M 4 can also conduct such a large transient current. The transient current conducted by transistor M 4 is larger than the stable bias current provided by the transistor M 5 and therefore, the node B will be pulled down. The transistor M 6 will approach to be turned off. As a result, the amplifier is in the class B mode to output a large source out current. In this transient state, although the transistor M 8 and the transistor M 7 are connected as a current mirror, the gain thereof is farther smaller than that of the transistor M 3 to transistor M 4 . Consequently, the transistor M 7 only conducts a minor current without affecting the pull down of node B, so as not to restrict the source out capability of the class AB amplifier.

Moreover, as the AB class amplifier of the present invention is working in large signal to conduct a large sink current, the node A will be pulled high so that the conducting currents of the transistors M 1 and M 2 are relatively small. Therefore, the conducting currents of the transistors M 7 , M 8 , M 3 and M 4 are also very small and are smaller than the stable bias current provided by the transistor M 5 . Thus, the node B will be pulled high to be close to the voltage source VDD. Thus, the transistor M 6 can conduct a large current. As a result, the amplifier is in the class B mode to conduct a large sink current from the load terminal.

FIG. 2 shows the phase curve (A) of the present class AB amplifier with a load capacitor C L of 100 pF and the phase curve (B) of the conventional amplifier. It is known that the class AB amplifier in accordance with the present invention is effective in suppressing the resonated peak gain.

With reference to FIG. 3 , there is shown another preferred embodiment of the CMOS class AB amplifier in accordance with the present invention. The CMOS class AB amplifier includes an adaptive level shift circuit 31 , a compensation capacitor C F , and an output transistor pair 32 . As shown in the figure, the adaptive level shift circuit 31 includes a NMOS transistor M 1 , used as an input amplifier, a NMOS transistor M 5 , used as a current source, two PMOS transistors M 3 and M 4 , used as a current mirror, a PMOS transistor M 8 , used as a switch, and a PMOS transistor M 7 , used as a diode. The gate of the transistors M 1 is connected to an output terminal (node A) of the differential amplifying circuit 33 . The gate of the transistor M 5 is connected to a bias terminal VB 1 . The sources of the transistors M 1 and M 5 are connected to a system low voltage VSS. The drains of the transistors M 1 and M 5 are connected to the drain of the transistor M 7 and the source of the transistor M 8 , respectively. The gates of the transistors M 3 and M 4 are connected together and further connected to the drain of the transistor M 3 .

The output transistor pair 12 is formed by a NMOS transistor M 2 and a PMOS transistor M 6 , each having a drain connected to one another. The gate of the transistor M 2 is connected to the gate of the transistor M 1 . The gate of the transistor M 6 is connected to the source of the transistor M 8 . The compensation capacitor C F is connected between the drain of the transistor M 2 and the gate of the transistor M 1 . The transistor M 7 is arranged between the output of the transistor M 1 and the transistor M 3 of the current mirror for providing a DC bias to control the transistor M 8 . As shown in the figure. The transistor M 7 has its gate and drain connected together so as to be formed as a diode. The source of the transistor M 8 is connected to the drain (node B) of the transistor M 4 . The drain of the transistor M 8 is connected to the drain of the transistor M 3 and the source of the transistor M 7 (node C). The gate of the transistor M 8 is connected to the connection point (node D) of the gate and drain of the transistor M 7 .

This preferred embodiment is different from the previous one in that the PMOS and NMOS transistors are replaced by NMOS and PMOS transistors, respectively. The operations of the two circuits are identical. Therefore, a detailed description for the operation of this embodiment is deemed unnecessary.

In view of the foregoing, it is known that, in the present invention, the transistor M 7 is utilized as a bias for driving the transistor M 8 so as to provide a relatively low linear resistor r DS8 for being used as a feedback. Therefore, the Q value of the gyration inductance can be effectively reduced, and the occurrence of the peak gain can be suppressed effectively, so as to maintain a required gain margin. Moreover, the connections of the transistor M 7 and transistor M 8 still allow the driving voltage of the output transistor M 6 not to be clamped in large signal. Therefore, the amplifier can work in the class AB mode to achieve the object of saving power and having a large driving capability without sacrificing the gain bandwidth.

Although the present invention has been explained in relation to its preferred embodiment, it is to be understood that many other possible modifications and variations can be made without departing from the spirit and scope of the invention as hereinafter claimed.

Claims

1. A complementary metal oxide semiconductor (CMOS) class AB amplifier having an adaptive level shift circuit, a compensation capacitor and an output transistor pair to receive an output of a differential amplifying circuit for being provided with a DC bias and a DC amplifing effect by the adaptive level shift circuit, phase compensated by the compensation capacitor, and driven by the output transistor pair, the adaptive level shift circuit comprising:a current mirror circuit having a first NMOS transistor and a second NMOS transistor, each having a gate connected to one another; a NMOS diode transistor having a gate and a drain connected to the gate; a NMOS switch transistor having a source connected to a drain of the second NMOS transistor of the current mirror circuit, a drain connected to a drain of the first NMOS transistor of the current mirror circuit and a source of the NMOS diode transistor, and a gate connected to the gate and drain of the NMOS diode transistor; a PMOS input amplifier transistor having a drain connected to the drain of the NMOS diode transistor; and a PMOS current source transistor having a drain connected to the source of the NMOS switch transistor.

2. The CMOS class AB amplifier as claimed in claim 1, wherein the output transistor pair comprises:a PMOS transistor having a drain, a gate connected to the gate of the PMOS input amplifier transistor, and a source connected to a voltage source; and a NMOS transistor having a drain connected to the drain of the PMOS transistor, a gate connected to the source of the NMOS switch transistor, and a source connected to a system low voltage.

3. The CMOS class AB amplifier as claimed in claim 2, wherein the compensation capacitor is connected between the drain of the PMOS transistor and the gate of the PMOS input amplifier transistor.

4. The CMOS class AB amplifier as claimed in claim 1, wherein the PMOS input amplifier has a gate connected to an output terminal of the differential amplifying circuit and a source connected to a voltage source.

5. The CMOS class AB amplifier as claimed in claim 1, wherein the PMOS current source transistor has a gate connected to a bias terminal and a source connected to a voltage source.

6. A complementary metal oxide semiconductor (CMOS) class AB amplifier having an adaptive level shift circuit, a compensation capacitor and an output transistor pair to receive an output of a differential amplifying circuit for being provided with a DC bias and a DC amplifying effect by the adaptive level shift circuit, phase compensated by the compensation capacitor, and driven by the output transistor pair, the adaptive level shift circuit comprising:a current mirror circuit having a first PMOS transistor and a second PMOS transistor, each having a gate connected to one another; a PMOS diode transistor having a gate and a drain connected to the gate; a PMOS switch transistor having a source connected to a drain of the second PMOS transistor of the current mirror circuit, a drain connected to a drain of the first PMOS transistor of the current mirror circuit and a source of the PMOS diode transistor, and a gate connected to the gate and drain of the PMOS diode transistor; a NMOS input amplifier transistor having a drain connected to the drain of the PMOS diode transistor; and a NMOS current source transistor having a drain connected to the source of the PMOS switch transistor.

7. The CMOS class amplifier as claimed in claim 6, wherein the output transistor pair comprises:a NMOS transistor having a drain, a gate connected to a gate of the NMOS input amplifier transistor, and a source connected to a system low voltage; and a PMOS transistor having a drain connected to the drain of the NMOS transistor, a gate connected to the source of the PMOS switch transistor, and a source connected to a voltage source.

8. The CMOS class AB amplifier as claimed in claim 7, wherein the compensation capacitor is connected between the drain of the NMOS transistor and the gate of the NMOS input amplifier transistor.

9. The CMOS class AB amplifier as claimed in claim 6, wherein the NMOS input amplifier has a gate connected to an output terminal of the differential amplifying circuit, and a source connected to a system low voltage.

10. The CMOS class AB amplifier as claimed in claim 6, wherein the NMOS current source transistor has a gate connected to a bias terminal, and a source connected to a system low voltage.