COMPLEX SYNTHETIC METHOD AND SYSTEM FOR PHYSICAL AND INFORMATION SIGNAL CANALIZING

Abstract

The present invention focuses a system and or a complex synthetic method, which considering any not orthogonal, independent plurality signals with limited frequency bandwidth, not greater than any f0 central frequency, allows to develop a resultant complex signal with limited frequency bandwidth, not greater than any relative f0 central frequency instead of the one sum of all the relative plurality independent not orthogonal signals frequency bandwidths. The resultant complex signal is the linear combination of orthogonal complex signals plurality. Each one of such orthogonal complex signals is characterized by a limited frequency bandwidth, not greater than any relative f0, and each is in bijection with the relative independent not orthogonal signal considered in the beginning

Description

The present invention concerns a complex synthetic channeling and/or energy harvesting method and/or system for canalizing physical & information signals which characterize any occurring physical event.

In particular, this invention focuses on a complex synthetic method/system called Hurricane Spread & Compression System (HSCS).

HSCS allows to develop a resultant complex signal (complex vector), which is the linear combination of a multiplicity of L tri-dimensional (3-D) complex orthogonal signals (where L is an integer number, the value of which is included into an [1; ∞] interval). These orthogonal signals are all correlated to the same f₀ frequency. Each l-th of them (where l=1, . . . , L, is the related mode index) occupies a limited required frequency bandwidth BB_l≤f₀.

According to the method envisaged by our invention, each above mentioned orthogonal signal is generated in bijection with the one of the L independent physical signals.

Such independent physical signals are not orthogonal ones but simultaneously occupy overlapping independent frequency bands, BB_l.

At the same time, according to our method, each l-th above mentioned orthogonal complex signal is well represented as a linear combination of KM orthogonal complex signals (where K and Mare integer numbers, the value of which is included into an [1; ∞] interval, k=1, . . . , K is the related frequency step index and m=1, . . . , M is the related azimuth phase step index).

The resultant complex vector composed by the LKM orthogonal signals occupies a limited frequency bandwidth, BW_RF≤BW₀, (e.g. BW₀=f₀). The value of the latter equals the maximum one amongst L input signals BB_l with 1=1, . . . , L, BW_RF=max[BB_l].

The set of L orthogonal signals generated by the invention univocally characterizes a specific complex algebraic vector space which is a linear combination of 3LM scalar dimensions (3LKM-D) and 3LKM unit vectors.

Consequently, the complex vector space assigned by the method of the invention is an algebraic Hilbert space.

Conversely, in the complex vector space generated according to the invention by the method linear combination of 3LKM scalar dimensions, 3LKM-D, and 3LKM unit vectors, it is always possible to detect each and every L orthogonal complex vectors which the invention method generates in bijection with the same L physical signals.

This way, the invention allows to multiply by L the capacity of the available frequency band. The invention method sets the L value, irrespective of the available frequency band and the central frequency.

Our method also allows to allocate assigned not zero resultant Poynting vector value along propagation axis too. Otherwise the method of invention allows to measure the possible Poynting vector value when different from zero even all along the propagation axis.

It is to be underlined that such performances, which would be impossible to achieve without our invention, increase the efficiency of:

the Transfer channel

the Store space

the Signal

• • sent out across a link
  • Radiated towards any natural physical phenomena (i.e., atmospheric or climatic ones or else) in order to create an interference leading to an adequate control action.

BACKGROUND

Techniques earlier our invention only use mono dimensional (1-D) or planar (2-D) vector interferometer methods. For example, in electromagnetic field propagation such former techniques use TEM TE e TM real mode approximation only (see ref. [10]).

As it is well known (ref. [7] [10] [11], in such approximation former systems fail to consider negligible the second time derivatives of electric (E) and magnetic(B) fields phase, as well as the quadratic terms which have been derived by the first time derivatives of both electric (E) and magnetic(B) fields phases; furthermore, they neglect the longitudinal electromagnetic components of the fields themselves.

In former methods, e.g., in those one used for astrophysical observations of weak radiations coming from cosmos bodies, (see ref. [2], [3] e [13]) criticalities like the following ones can be traced:

• • It is possible to assign or detect the vector or scalar transverse field component only;
  • In far distance systems, a lot of the radiated energy (information) which is included into longitudinal Orbital Angular Momentum (OAM) components is lost.
  • Every longitudinal field components are ignored, therefore azimuthal phase ambiguities occur;
  • Serious destructive interferometric errors occur during detection. The destructive action increases in proportion to the propagation distance as well as to the L value [1], [2] e [3];
  • Far distance system sensitivity as well as detection of near radiations with very weak transversal components are seriously limited by destructive planar interferometric azimuthal phase errors;
  • Errors due to detection ambiguity restrict the validity of wave propagation mode approximations.
  • The value of L maximum quantity of OAM mode is strongly limited by azimuth phase interference.
  • Further reductions of L quantity of channels occur because of the strong increase in distance, e.g., in satellite communications applications;
  • The Poynting vector computation value along the longitudinal propagation axis is always equal to zero;
  • The approximation works for TEM, TE or TM real mode only (ref. [1], [2], [3], [10] and [11]);
  • As well known in Physics literature the far distance field measurements method is hugely different from the near distance field one.

Scope

The scope of the present invention is:

• • To find solutions to the problems and criticalities mentioned above;
  • To improve the Angular Momentum (AnM) and OAM approximation related to the observed wave beam equation complex modes;
  • To facilitate innovative applications.

Object

Object of the present invention is a method and/or a system, as mentioned in the attached responsibility declarations, which are integrating part of the present description. The present invention concerns physical signals processing.

A generic l-th (with l=1, . . . L) periodical physical signal, is represented by a periodical complex function SF_l(t). SF_l(t) describes a generic natural or artificial physical event in the time-space domain. It is defined in a Euclidean three-dimensional (3-D) space. SF_l(t) arrives at the HSCS input.

Examples of physical events are atomic radiations or acoustic, or climatic, or thermodynamic, or gravitational wave propagation, for example the one coming from any universe black hole, or electromagnetic wave, for example the one coming from the Early Universe etc. . . . events.

The following should be considered indicative only but not restrictive. First, the description of our invention, shows a generalized algebraic model which is valid for every proposed specific invention implementation, and following after it details each of that proposed specific invention implementation.

Each implementation model detailed below is to be considered as an integral part of the basic model and it is also to be considered as necessary to characterize the invention method.

Features and advantages of our invention are highlighted in the detailed description of some implementations provided for in the figures described below:

FIG. 1 shows a block diagram of the system operating principle according to our invention. The block diagram points out an invention implementation of the HSCS related to the requested complex vector space generation and/or detection;

FIG. 2 shows the details of a block diagram implementing the specific I/O input interfaces in the HSCS system. The block diagram points out an invention implementation which presents the signal data and “control data” generator.

FIG. 3 shows the details of the operating principle diagram of the OAM transmitter modulator described in FIG. 1. The block diagram points out an invention implementation of the chirp and shifted linear phase modulators with L=3, K=1 and M=4.

FIG. 4 shows a detail of FIG. 3. The block diagram points out an invention implementation which presents the complex (l and Q) generator (e.g. in the specific case: with a single side band signal, SSB, as carrier), the under-sampler and transducer, related to the lkm-th signal synthesis in the specific case, “a”: continuous chirp-continuous phase.

FIG. 5 shows a detail of FIG. 3. The block diagram points out an invention implementation which presents the complex (l and Q) generator (e.g. in the specific case: with a single side band signal, SSB, as carrier), the under-sampler and transducer, related to the lkm-th signal synthesis in the specific case, “b”: stepped chirp-continuous phase.

FIG. 6 shows a detail of FIG. 3. The block diagram points out an invention implementation which presents the complex (l and Q) generator (e.g. in the specific case: with a single side band signal, SSB, as carrier), the under-sampler and transducer, related to the lkm-th signal synthesis in the specific case, “c”: stepped chirp-stepped phase.

FIG. 7 shows a details of FIG. 3. The block diagram points out an invention implementation which presents the complex (l and Q) generator (e.g. in the specific case: with a single side band signal, SSB, as carrier), the under-sampler and transducer, related to the lkm-th signal synthesis in the specific case, “d”: stepped frequency-stepped phase.

FIG. 8 shows the details of FIG. 1. The block diagram points out an invention implementation which presents the detector, the complex (I and Q) extractor, the under-sampler HSCS_LM⁻¹ and the output interface, with L=3, K=1 and M=4.

FIG. 9 shows the details of block diagram of FIG. 8, the implementation related to the/km-th signal in all cases (a, b, c, d). The block diagram points out an invention implementation which presents the digital phase detector (DPD) and the extractor.

FIG. 10 shows the details of the block diagram of the specific HSCS I/O interface as defined by the invention.

FIG. 11 shows the details of FIG. 1. The block diagram points out the example of an invention implementation which presents the Analog wide bandwidth 2^nd order Power harvesting detector and complex (I and Q) extractor and compressor, using a depletion Mosfet bridge in specific Gate Source short-circuit configuration.

FIG. 12 shows the details of FIG. 1. The block diagram points out the example of an invention implementation which presents the Analog wide bandwidth 2^nd order Power harvesting detector and complex (I and Q) extractor and compressor, using a depletion Mosfet bridge in specific Darlington Gate 1 Source 2 as well Gate 2 Source 1 relatives short-circuit configuration.

FIG. 13 shows the details of FIG. 1. The block diagram points out tow examples of HSCS_LKM⁻¹ invention implementations which present two circular polarization Analog wide bandwidth 2^nd order Power harvesting detector and complex (I and Q) extractor and compressor configurations using a depletion Mosfet bridge in specific Darlington Gate 1 Source 2 as well Gate 2 Source 1 relatives short-circuit configuration.

FIG. 14 shows the details of FIG. 1. The block diagram points out an example of HSCS_LKM' invention implementations which present a linear polarization Analog wide bandwidth 2^nd order Power harvesting detector and complex (I and Q) extractor and compressor configuration using a depletion Mosfet bridge in specific Darlington Gate 1 Source 2 as well Gate 2 Source 1 relatives short-circuit configuration.

FIG. 15 shows the details of the block diagram of the Matlab (or others) code program which can be used to compute parameters of an HSCS system design as defined by the invention.

FIG. 16 points out the details of FIG. 15. It is a simplified example of a circular polarized Front End simulator block diagram in a Matlab (or other) code, for the verification of project parameters in a HSCS system according to the invention.

FIG. 17 points out the details of FIG. 15. It is a simplified example of a linear polarized Front End simulator block diagram in a Matlab (or other) code, for the verification of project parameters in a HSCS system according to the invention.

FIG. 18 points out the details of FIG. 15. It is a simplified example of an extractor and compressor simulator block diagram in a Mathlab (or other) code, for the verification of project parameters in a HSCS system according to the invention.

INVENTION DEFINITIONS

The HSCS of the invention system generates a 3LKM-D dimensional algebraic Hilbert space. It is in bijection with a plurality of three-dimensional (3-D) complex vectors which are defined in complex scalar Cartesian (x, y, z) and/or vector Euclidean (x, y, z) space reference systems.

Clearly, such space reference systems are related to scalar (ϕ, θ, z) and/or vector (ϕ, θ, z) ones, respectively.

Class of Signals in the Invention

• • Physical signal l-th is a complex periodic function, SF_l(t), which describes the specific time variation of the physical event (natural and artificial alike).
  • Electric signal, SE_l(0, is the output signal from a general transducer. Its variable (in the time and/or frequency domain) is in bijection with the one which pertains to the natural or artificial physical event SF_l(t).

The general l-th independent electric signal, SE_l(t), which transports the contents of information (or energy) to be processed, may appear in the following two algebraic form:

• • m_l(t): a periodic complex function, where its frequency is f_BBl≤f₀ and its frequency bandwidth is BB_l≤BW₀ and index l=1, . . . , L: m_l(t)=m_l exp(-j2πf_BBlt).
  • SI_l(t): a general composed periodic complex function. It is a composed function of m_l(t). SI_l(t) may assume forms two:
    • 1. Polynomial function, composed by R components (where R is an integer number, the value of which is included into an [1; ∞] interval). An FDM signals sequence e.g. each rth Polynomial (where r=1, . . . , R) presents a limited independent frequency bandwidth, BB_r≤BB_l, which is not overlapped and which is consecutive to the other one. The resultant composed by the R Polynomial frequency bandwidths is BB_l=Σ^R_r=1BB_r.
    • 2. Single general modulated function (any modulation, such as digital and/or continuous ones, angular and/or amplitude related ones) with a frequency carrier fo and a generic m_l(t) modulating signal. In the easiest case, e.g.:

SI _l(t)=F[m(t); f₀]=m_l exp[j2π(f₀-f_BBl)t].

Each l-th (where l=1, . . . , N) SI_l(t), electric signal function, has its own BB_l independent frequency bandwidth, Furthermore:

BB _l ≥f _sl ≥f _BBl; Σ^L_l=1BB_l≤Lf₀; Σ^L_l=1Σ^M_m=1BB_l≤LMf₀; and BW₀≥BB_l.

• • S_l(t)=S[m(t); f₀] is a complex l-th time function (or S_l(jΩ) in frequency domain) generated by the invention implementation. Its independent and limited frequency bandwidth is BW_Sl≤BB_l. Its central frequency is f_Sl≤f₀. In the easiest case: f_Sl=f₀-f_BBl

so that:

Σ^L_l=1BW_Sk≤(L)BW₀ and Σ^L_l=1Σ^M_m=1BW_Sl≤(L)MBW₀

• • S_lkm(t)=S[S_l(t); l; k; m] is a complex lkmth function generated by the invention method. It depends on S_l(t) and index l, k, and m. Its independent and limited frequency bandwidth, for example, is BW_Slkm≤BB_l[(l-1)K+k]m. Its central frequency is f_Slkm≤f₀[(l-1)+k/K]). In the easiest case: f_Slkm (f₀-f_BBl) [(l-1)+k/K]

The Invention Concept

HSCS system develops a linear combination of L assigned independent not orthogonal functions (or algebraic scalar polynomial functions), m_l(t) (or SI_l(t), in bijection with L spatial (3-D) orthogonal complex vectors functions. All the orthogonal complex vectors generated by the invention method are correlated to the frequencyf₀.

The HSCS (3-D) orthogonal complex vectors identify an univocal (3-D) complex vector space, which is an algebraic Hilbert space.

The complex vector space assigned frequency bandwidth is BW_RF if each of the independent m_l(t) (or SI_l(t)) frequency bandwidth BB_l is limited, too.

Each HSCS l-th orthogonal vector generated by the invention is equipped with the same requested temporal phase, f₀t, and an assigned l-th (3-D) specific spherical spatial phase, ψ_l(lf₀; lϕ).

ψ_l(lf₀; lϕ) is assigned by the invention method and is time independent. Once f₀ is set, ψ_l(lf₀; lϕ) belongs to the l relative index value only. The invention sets ψ_l(lf₀; lϕ) as composed by a pair of time independent planar phases:

• • Azimuth phase, the value of which depends on lϕ, is included into an [0; L2π] interval
  • Tilt phase, the value of which depends on If₀ ,is included into an [−(90°-π/2L), (90°-π/2L)] interval

Conversely, by the invention method HSCS it is possible to detect the same vector space orthogonal complex vectors (FIG. 8) from any 3-D complex vector space.

Separately or simultaneously HSCS, of the invention, extracts all the required 3-D orthogonal complex vectors which are necessary and sufficient to reconstruct all the required L m_l(t) or SI_l(t).

The operating principle of the invention method is based on the HSCS operator, which is an integral part of the invention as well as a characteristic one of it. According to the invention method, the HSCS is in turn composed by four basic operators, which are described as follows:

HSCS_LKM Transform Operators

The transform operator indicated as HSCS_LKM (see FIG. 1 block 12000) is an integral as well as a characteristic part of the invention. It is a linear operator. It is defined by the linear combination of all the LKM S_lkm(t). HSCS_LKM is represented as a linear combination of 3LKM complex orthogonal vectors. Each vector is in turn a linear combination of 3LKM unit vectors. The invention assigns LKM coefficient C_LMSlkmEach C_LMSlkm is calculated (e.g. by a Matlab code) according to the specific application and the l, k and m index value (data and frequency bandwidth compression/decompression, and/or information or energy canalizing/extraction, etc.). The HSCS_LM scalar expression is:

${\mathrm{HSCS}}_{\mathrm{LKM}}==\sum _{l=1}^L\sum _{k=1}^K\sum _{m=1}^MS_{\mathrm{lkm}}\left(t\right)=\sum _{l=1}^L\sum _{k=1}^K\sum _{m=1}^MC_{\mathrm{LMSlkm}}\exp \left\{-j\left[2\pi f_0t+{\psi }_{\mathrm{lkm}}\left({\mathrm{lf}}_0;l\phi ;k;m\right)\right]\right\}$

G_e(λ₀) Transducer Operators

The invention transducer operator is defined by its assigned transfer function G_e(λ₀).

G_e(λ₀) transforms the HSCS_LM output in physical signal (see FIG. 1 block 13000). HSCS uses G_e(λ₀) to generate the field related to the invention complex vector space. Its implementation depends on the specific invention application. In electromagnetic application, for example, G_e(λ₀) represents the transmitter antenna transfer function.

G_e(λ₀) Sensor Operators

The invention sensor operator is defined by its assigned transfer function G_c(λ₀)

G_c(λ₀) transforms the physical signal in a specific electrical signal (see FIG. 1 block 15000).

Usually, G_c(λ₀) is the inverse of the G_e(λ₀) or in many others applications G_e(λ₀)≠1/G_e(λ₀) is possible. Its implementation depends on the specific invention application. HSCS uses G_c(λ₀) to detect simultaneously or separately all the LKM orthogonal complex vectors which are correlated to f₀.

HSCS_LKM⁻¹ Anti-Transform Operators

HSCS_LM⁻¹ anti-transform operator (see block 16000 of FIG. 1), is an integral as well as a characteristic part of the invention. It is a linear operator. It is defined by the inverse linear combination of HSCS_LM operator. HSCS_LM⁻¹ allows to extract simultaneously or separately all the LKM orthogonal complex vectors which are correlated to f₀.

HSCS_LKM⁻¹[HSCS_LM]=Σ^L_l=1Σ^K_k=1Σ^M_m=1HSCS_lkm⁻¹[S_lkm(t)].

HSCS_LKM⁻¹ of the invention reconstructs all the L requested m_l(t) or SI_l(t) independent functions.

Therefore, HSCS of the invention is composed by two different parts which are considered as two separate subsystems:

• 1. Emitter: G_e(λ₀)*HSCS_LM=G_e(λ₀*Σ^L_l=1Σ^K_k=1Σ^M_m=1S_lkm(t)], related to an HSCS_LM linear transform operator and G_e(λ₀) transducer operator process. The subsystem generates a complex vector space (Hilbert space) and a requested related complex field. The subsystem is characterized by a vector complex scalar coefficients 3 LKM×3LKM matrix. The complex matrix developed by the invention method, e.g. in the electromagnetic case, describes the complex scalar field components amplitude distribution which satisfies the wave equation in the paraxial approximation (see ref. [10]).
• 2. Collector: G_c(λ₀)*HSCS_LKM⁻¹=G_c(λ₀)*HSCS_LKM⁻¹[Σ^L_l=1Σ^K_k=1Σ^M_m=1S_lkm(t)], related to G_e(λ0) sensor operator and HSCS_LM⁻¹ linear anti-transform. The subsystem detects and extracts the complex orthogonal vector correlated to f₀. The subsystem is characterized by a vector complex scalar coefficients 3LKM×3LKM matrix. The matrix of the subsystem, developed by the invention method, is the inverse of the first one above.

OAM Spatial Phase Resultant

OAM spatial phase resultant generated by invention assigned complex vector 3LKM-dimensional space is indicated in the following:

Ψ_LM(Lf₀; Lϕ)=Ψ[ψ_lkm(lf₀; lϕ; k; m); L, K, M]

Once f₀ is set, the invention Ψ_LM(Lf₀; Lϕ) belongs to the L, K and M parameters values which are assigned by the invention method. Ψ_LM(Lf₀; Lϕ) is time invariant and is assigned by the invention method.

l-th Harmonic Phase

l-th harmonic phase of invention is: f_lt=lf₀t.

Relative Carrier Frequency of the I-th OAM Mode

The relative carrier frequency of the l-th OAM mode is generated by the invention method. It is related to the fundamental frequencyf₀=2πΩ₀. It depends on the specific application, f₀ and l index. Its expression, e.g. in the simplest case, is:

Ω_RFSl=Ω₀-Ω_BBSl; and dϕ/T₀=Ω_RFSl.

Besides, according to the invention method what results is the following:

• l-th azimuth phase: φ_RFSl=−l dϕ; and ϕ=2π-φ_RFSl.
• l-th tilt phase: θ_RFSl=φ_RFSl/(4l); and θ=90°-θ_RFSl; dθ=−θ_RFSl

channel index: l=1/tgθ_RFSl.

Development Surrounding of the Invention

Three of the possible development surroundings of the invention are considered here. Without excluding any other development surroundings, they are listed below in illustrative but not restrictive way:

• • 1. Physical: natural and/or not natural (artificial) and/or Synthetic (electronics, and/or mechanics, etc . . . . );
  • 2. Theoretical Numerical;
  • 3. Theoretical Algebraic.

Modality of Implementation

The invention implementation modalities are:

• Complete system, HSCS (see FIG. 1);
• Single independent subsystem, one of two separately:

Emitter: [HSCS_LKM G_e(λ₀)] (see FIG. 3);

Collector: [G_c(λ₀) HSCS_LKM⁻¹] (see FIG. 8).

Typology of Implementation

Five possible typologies of implementation are considered. Each of them is to be considered as an integral part of the invention.

They are identified by their specific level of implementation complexity. The invention method implements each of them by using a different architecture of the HSCS. Each typology allows a different improving factor of the related phase error margin. That phase error margin factor is inversely proportional to the specific level of implementation complexity. They are all feasible and listed below:

• a. Continuous chirp and continuous phase. Its implementation complexity level is high. It is valid in every condition, especially suitable in mentioned theoretical algebraic environments applications;
• b. Stepped chirp and continuous phase. Its implementation complexity level is more than medium. It is valid in every condition, especially suitable in mentioned theoretical numerical environments applications;
• c. Stepped chirp and stepped phase. Its implementation complexity level is optimal. It is valid in every condition, especially suitable in mentioned natural physical environments applications;
• d. Stepped frequency and stepped phase. Its implementation complexity level is low. It is valid in every condition, especially suitable in mentioned physical environments applications
• e. Analog wide frequency bandwidth. Its implementation complexity level is low. It is valid in every condition, especially suitable in mentioned physical as well as power transferring and energy harvesting and measuring environments applications.

The invention method uses only one functional scheme (see FIG. 1) to implement any of the four typologies highlighted above. The scheme of the invention method consists of seven equivalent functional blocks. The outputs of one block are the inputs of the following one. Each functional block defined by the invention method will be detailed according to the required specific implementation typology.

General Description of Invention Implementation

With reference to the figures, according to the invention, a method is supplied to canalize and transfer a multiplicity of periodical independent physical signals represented by SF_l(t), complex functions in the time domain t, where L is an integer number, the value of which is included into an [1; ∞] interval and l=1, . . . L, each one occupying a BB_l, independent frequency band, e.g. BB_l=[(f₀-f_BBl), f₀], Some of such independent signals are e.g.: electrical, electromagnetic, digital and/or analog, mechanical oscillations, pressure waves, optical signals. Starting from these independent signals the invention method generates a single complex signal, which is the resultant signal of L orthogonal components. Each l-th of that orthogonal components is in bijection with the respective SF_l(t). The single complex signal is correlated at a frequency f₀ and occupies a single frequency bandwidth equal to BW_RF=max [BB_l], the largest band among said BB_l bands. The invention method performs the following steps:

• • A. acquiring (11000) the physical signals SF(t) and transforming them (11000) into electrical signals by means of one or more transducers;
  • B. extracting (11000) from each electrical signal of the previous step the relevant information content as modulation function m_l(t);
  • C. performing (11000) the Fourier Transform of m_l(t), obtaining the function m_l(jΩ);
  • D. generating (12100), for each modulation function m,(/co), a corresponding periodic digital signal P_l(jΩ)=S_l(jΩ) having carrier frequencyfo and modulation function m_l(jΩ);
  • E. generating (12100), for each periodic digital signal P_l(jΩ), a pair of signals in quadrature;
  • F. generating (12100), for each pair of signals in quadrature, mentioned in step E, K*M pairs of signals PP_lkm(jΩ)=S_lkm(jΩ), where K and M are integer numbers, the value of which is included into an [1; ∞] interval and k=1, . . . , K as well as m=1, . . . , M. Each pair of signals PP_lkm(jΩ) is generated through the following steps performed subsequently in any order or at the same time:
    • performing KM chirp modulations, each characterized by a corresponding frequency increment Δf_lkm=[(l-1)+k/K] (f₀--f_BBl) with varying k, in a time interval ΔT=T₀ at a clock frequency f_c≥2LKMf₀;
    • performing KM phase linear modulations. Each of K*M phase linear modulations is characterized by a corresponding phase increment Δϕ_lkm=-l[1-(K-k)/K] (m/M)2π with varying k and m, at a clock frequency of e.g. f_ck≥2LKMf₀;
  • G. under sampling (12200) said pair of signals PP_lkm(jΩ) at a frequency f₀ thus obtaining a pair of canalized signals PP⁰_lm(jΩ) which by construction have a spatial phase (3-D) orthogonal to each of the other (L*K*M-1) signal pairs and moreover they are correlated to f₀;
  • H . vector summing (12200) all the L*K*M pairs of orthogonal PP⁰_lkm(jΩ) components thus obtaining a pair of LKM-dimension signals PP⁰(jΩ) occupying band BW_RF≤f₀;
  • I . sending the LKM-dimensional signal PP⁰(jΩ) to a signals emitter-transducer (13000) tuned to f₀, which performs whatever polarization.

According to the invention, step H mey be also realized by a simple collection node of all the orthogonal signals produced in step G.

According to the invention, steps E and F may be also inverted.

According to the invention, steps E and/or F and/or G and/or H and/or I may be also performed at the same time.

According to the invention, the transducer of step I may be also a signal transmission means with unitary transfer function (e.g. a cable connector).

According to the invention, it is further provided a method for the extraction of a plurality of periodical physical signals represented by complex functions in the time domain t SF_l(t) where L is an integer number the value of which is included into an [1; ∞] interval and l=1, . . . L, each of which occupy an independent frequency band BA (e.g. BB_l=[(f₀-f_BBl), f₀]), such as electric, electromagnetic, digital, mechanical oscillation, pressure waves optical signals.

The extraction starts from a single signal correlated to a frequency f₀ occupying a band whose width is equal to the largest band among said bands RB_l, BW_RF=max[BB_l], wherein the following steps are performed:

• • J. detecting (15000) by means of a sensor tuned tof₀, which performs whatever polarization, said complex single signal to obtain PP⁰ (jΩ)=HSCS_LM;
  • K. applying (16100) a complex correlator, at frequency f₀, with a clock frequency f_ck, to the signal detected by said sensor, thus obtaining LKM pair of signals, PP⁰_lkm(jΩ)=HSCS_LKM⁻¹(S_lkm);
  • L. performing (16200), for each one or single one values of l=1, . . . L, KM contemporary convolutions, with a clock frequency f_ck, of each signal pair PP⁰_lkm(jΩ), the KM convolutions being configured to extract KM respective chirps with frequency shift Δf_lkm=[(l-1)+k/K] f₀-f_BBl) and an azimuth phase delay Δϕ_lkm=-l[(K-k)/K] (m/M)2πin an interval T₀ from said signals pair, obtaining corresponding pairs of signals with k=1, . . . K and m=1, . . . M for each l;
  • M. summing (16200) the KM components of each l value, PP_l(jΩ), obtaining all the independent modulation functions m_l(jΩ) from said single signal with l varying;
  • N. performing an Fourier Anti-Transform (16100) of m_l(jΩ) and thus obtaining corresponding modulation functions m_l(t);
  • O. transforming (17000) if required m_l(t) into SF_l(t) by one or more transducer.

According to the invention, said an only signal may be also the signal emitted by said emitter-transducer 13000) of signals tuned to f₀, according to any claims 1 to 5.

According to the invention, the polarization of step I may be also the same or is different from the polarization of step J.

According to an aspect of the invention, in step F and/or correspondingly in step L, the value of M and/or K and/or L is infinite, thus obtaining infinite continuous chirps and linear phase modulation Components, wherein the summation is replaced by integral operation.

According to an aspect of the invention, the starting periodic physical signal set is subdivided into a plurality of periodic physical signals, to each element of the plurality being applied the steps of the method.

According to an aspect of the invention, it is provided a computer program, comprising code tools set up in such a way that, when running on said computer, perform steps B to H and/or K to N according to a method of any previous claim.

According to an aspect of the invention, it is provided a system 10000 for the canalization and/or extraction of a plurality of physical signals, wherein the system comprises:

• • acquisition devices 11000 of a set of original physical signals;
  • one or more transducers 11000 for the transformation of the original physical signals into electric signals;
  • a computerized sub-system 12000 to process said electric signals according to steps B to H of the method according to any claims 1 to 5, thus obtaining one or more final electric signals;
  • one or more transducers 13000 for the transformation of said final electric signals into final physical signals;
  • emission devices 13000 for the emission of said one or more final physical signals; and/or
  • sensors 15000 for the acquisition of said one or more final physical signals;
  • one or more transducers 15000 for the transformation of physical signals into electric signals;
  • a computerized sub-system 16000 for processing said electric signals according to steps K to N of the method according the invention, obtaining one or more electric signals;
  • one or more transducers 17000 for obtaining said original physical systems from said electric signals.

Description of an Implementation of the Generalized Algebraic Model According to the Invention

The following implementation example can be applied in the most general case (therefore in all a, b, c, d, e, cases already described). Such invention implementation is specifically best for the more complex case of theoretical algebraic Continuous Spectrum (Continuous Chirp-Continuous Phase, already mentioned as “a” case).

Actually its implementation requires a greater processes' precision and reiteration technological level than the other four cases (b, c, d and e) according to requested approximation degree.

Once the following values have been fixed:

• • f₀;
  • f_RFl, the l-th carrier (e.g. in the easiest case: at f_RFl=f₀-f_BBl);
  • L, the number of components
  • LKM, phase steps [e.g. the easiest case, with K=1, phase steps varying value: from 0 to ±L2it (azimuth); from 0 to ±(n/2-7e2L) (Tilt)].the spatial coefficients and the parameters related to the project are assigned, according to the invention, also using e.g. the already mentioned Matlab Code as a programming tool which is described in the following.

Input Interface 11000 of FIG. 1

Here we refer to the flow diagram in FIG. 2 that illustrates a 10000 system portion implementation example according to the invention.

The generic signal m_l(t)=m_l exp(j2πf_BBlt) , or SI_l(t) as an alternative, reaches the input of a standard interface 11001. After being filtered by the anti-alias Polyphase one of 11002 block, the signal is sent to the sampler of 11003 block which could be a Serial Digital to Digital or an Analog to Digital standard converter, at a frequency rate of f=f_CK. The frequency rate of the example has been set to f_CK≥2LKMf₀. The output of 11003 block is sampled and, in case conveniently de-modulated by 11004 block. The output of 11004 reaches the entry of the optional stage 11005. The 11005 output, corresponding to complex vector m_l(t)=m_l exp[j(2πf_BBlt)] sampled at f_CK, is the output of this process phase of the invention.

Optionally, this output can be further processed by a 11006 FFT stage to extract the data related to the m_l(t) message frequency spectrum. The aim of this optional process is to generate control data to be sent in cascade to a standard controller of 11007 block.

The 11007 controller puts the “Register control data” into a proper digital format. Each specific lkm-th “Register control data” is conveniently transmitted to the lkm-th related register of the successive stage (12108 block of FIG. 4).

Expander (I & Q) Complex Generator 12100 of FIG. 1

System generator is implemented in this case with a bench of L*K*M direct digital synthesis (D.D.S.) generator devices (see the implementation block diagram of FIG. 3 where L=3, K=1, L=4).

FIG. 4 shows a specific detailed development of a lkm-th single elementary DDS generator.

The lkm-th “Register control data”, related to the m_l(t) signal, obtained as output of 11007 block of FIG. 2, go into 12108 register of FIG. 4, dedicated to the carrier, S_l(t), specific information (e.g., “SSB delta phase”) in the corresponding DDS having l, k and m indexes. The lkm-th “Register control data” is requested to digital synthesize the carrier signal, S_l(t), with an assigned frequency f_RFl. Each of the 12100 generators (see FIG. 3) is realized as a D.D.S. and it is highlighted in FIG. 4. The l-th control data corresponding to the m_l(t) signal, coming from the input interface, entering the HSCS system invention, is essential to digitally synthesize the S_l(t) signal having f₀ carrier and m_l(t) modulating, with any appropriate modulation required, according to the specific application of the invention (in the example here the single side band, SSB is considered).

The “Register control data” designated to the specific information of the “SSB delta phase”, assigned by the invention, and contained in the 12108 block, enter the combiner, 12110 block of FIG. 4. Therefore 12110 block provides the data related to the instant incremental phase value of S_l(t) that are to be sent to 12111 block, which is a phase accumulator. The 12111 block output data enter the 12113 block which includes an algebraic structure (lookup table) originating the I and Q pair (see FIG. 4) of the SSB signal digital direct synthesis associated to m_l(t).

Then, each element of the I & Q pair is auto-correlated by a first resonant stage 12114, having a multiplier tuned to f_RF as algebraic model. This tuned multiplier is built by a couple of synchronous digital resonators put in parallel and f₀ correlated (at least for a small phase error of ε proportional to Ω_BBl/Ω₀). Those synchronous digital resonators work simultaneously, at the same temporal and angular conditions, and on the same signal m_l(t)) linked.

The two I and Q pairs out bounding the 12114 block are respectively represented by two synchronized polynomials. Those synchronized polynomials identify an infinite number of simultaneous intermodulation products related to the same typical single carrier frequency f=f_RF ∀m_l(t) (having a small proportional phase error less than the corresponding Ω_Sl/Ω₀). The two I and Q pairs enter the second “resonant” 12115 stage. This second “resonant” 12115 stage has an algebraic model of a multiplier. Inside the 12115 stage, each simultaneous component related to each of the two input sequences, is modulated by each of the identical and simultaneous components of the other one sequence present there.

The I and Q pair output from 12115 block enters a digital low-pass filter FIR 12116 having f_−1dB=(f₀-f_BBl)[(l-1)+k/K]. The low-pass filter FIR 12116 is in charge of harmonic suppressing having f>(f₀-f_BBl)[(l-1)+k/K] frequency, which is typical of the l-th sequence couple I and Q.

Complex Under-Sampler 12200 of FIG. 1

The output sequence pair (I and Q) of 12216 block of FIG. 4 are added and under-sampled through a sequence of two blocks in cascade:

• • Sum that performs a vector addition (12217 block) of all the LKM pairs outgoing from the filters (12216 block) in a 3-D complex space;
  • A digital f₀ multiplier (or mixer) that uses a complex correlation f₀ based, to assign the polarity to 12216 block output signal. Depending on the polarization to be assigned, the polarization can be implemented in two different manners.
    • 1. A cylindrical structure having a circular, or even elliptic, polarization (with typical complex Divergent mode).

Here a 12218 block complex double mixer (one cosf₀t, and the other sinf₀t) generates two quadrature outputs, for each I and Q. These output signal pairs have been generated with a general complex vertical and horizontal polarization, designated as x^VI and x^HI (for 1) and as x^VQ and x^HQ (for Q), each respectively with cosf₀t or sinf₀t carrier.

Combiner, 12219 block, follows in cascade, and performs separately two vector additions in a 3-D space. It follows that, those two resultants vectors (V and H respectively indicated) are both correlated to _thefo carrier, and each in quadrature (90° out of phase) to the other one:

• • The one V as: x^VI +x^VQ
  • The other H as: x^HI +x^HQ.
  • 2. A linear polarization (with typical complex Non-TEM mode).
  • In this case, each of all the LKM pairs outgoing from the previous 12217 stage, is sent to the 12220 block. Here the LKM components are, first of all, respectively correlated with a f₀ frequency by two quadrature digital complex mixer (one cosf₀t, and the other senf₀t). The respective output components of the two mixers are here in-phase. Because they are orthogonal too, then here they are one to each other added and correlated with a f₀ frequency.
  • Optionally this purpose is implemented by 13023 block of FIG. 1. Such purpose, e.g. in the electromagnetic transmission system, which performs a linear polarization field, is carried out by a simple λ₀ tuned antenna. In this case 12220 stage is skipped and does not participate to the invention process. Furthermore, the block 12217 can be just realized by sending every of the 12116 outputs (each one orthogonal to the other) directly to the 12218 or 12220 stage's entries: then stage 12117 becomes merely functional symbol.

Optionally, each of the LKM output components outgoing from 12119 or 12120 blocks, enter in a D/A converter (12221) (in case an Analogic field to be generated is required). Output 12221 block is the input of 12222 block, a low-pass or a pass-bandwidth filter (also optional). In the example explained here, it is a pass-bandwidth filter BW_HSCS=BW_RF in charge of letting go each orthogonal component overlapping in the BW₀ band.

Emitter Transducer 13000 FIG. 1

For simplicity, here in the most general case, it is an Operator that, based on the polarity whatever assigned (linear, circular or even elliptic) by the previous stage, generates the algebraic field depending on each I and Q pair outputs from 12119 or 12120 preceding stages (see FIG. 3-4).

The output signal from 12222 block enters 13023 block. This block provides a transfer faction equal to G_e(λ₀) already described.

Actually, the transducer is optional: for the easiest case like a data transfer over a data network (wired network) the transducer might not be used. The equivalent is obtained setting the transferring functions =1(G_e(λ₀)=G_c(λ₀)=1).

Field, 14000 of FIG. 1

13023 block output is an algebraic expression representing a field that occupies a BW₀ bandwidth allocated by HSCS_LKM, in an infinite complex 3-D space.

Optionally the output from 13023 block (see FIG. 1) enters a combiner 14024 block (see FIG. 1). 13023 block executes a sum of HSCS_LKM generated field and all the vector elements belonging to other different complex vector spaces, that is with all the possible algebraic elements, within or outside the BW₀ frequency bandwidth, existing in the same complex infinite 3-D space.

Collector Sensor 15000 of FIG. 1

This Collector Sensor performs the inverse transducer 13000 block operation and it fulfills the role of a sensor (15025 block in FIG. 1). It is implemented according to the field polarization and it is characterized by a transfer function G_c(λ₀) already described. It reveals each of the LKM 3-D orthogonal vectors of the complex vector space assigned by HSCS_LM together with any other different vector space signal existing on its input, and correlates them to a f₀ frequency.

All those complex 3-D modes occupy the same assigned bandwidth BW₀ defined by the frequencies falling in the interval (e.g. here f=[(f₀-f_BWo), f₀], and all are f₀ correlated.

The sensor, like the transducer above, is actually optional. In effect, in the easiest case of a simple data transfer over a data network (e.g.: wired network), the sensor is not needed. This is the same as putting the sensor and transducer transfer functions to 1 (G_e(λ₀)=G_c(λ₀)=1).

Complex Extractor 16000 of FIG. 1, HSCS_LM⁻¹

Here the activated L modes HSCS_LM are extracted (individually or simultaneously according to the invention implementations) from any point of the field and at every t instant. Each l-th linear combination can be continuous or discrete, depending on application cases. Each l-th linear combination is represented by a complex periodic vector algebraic expression Ω₀ correlated, and it is composed by 3LKM orthogonal vectors having 3LKM unit-vectors. Referring to FIGS. 1, 8 and 9 an invention implementation is described as follows.

Optionally, a proper filter (16126 block of FIG. 8) selects the components coming out from 15025 block (of FIG. 1 or 8). This filter selects the frequencies included in the bandwidth interval, Δf=[(f₀-f_BWo), lf₀], in the event of a single l-th mode is of interest.

In case all the L modes together are the matter of interest, then the optional filter will be a low-pass one, in order to select the frequencies contained in the [0, Lf₀] interval. It is not needed, and can be skipped, in case of a data transfer on a digital data network.

The output signal of 16126 block in FIG. 8, is then converted into a Digital one by a single large bandwidth Sampler (16127 block of FIG. 8) having a proper clock frequency (in the example here f_ck≥2LKM//f₀). This stage is optional, and it is inserted only in the event of an Analogic input.

Then, this stage sends the samples to 16128 stage (see FIGS. 8 and 9) which is a complex mixer (I e Q). The Digital Phase Detector (DPD) 16128 (see FIG. 8 or FIG. 9) detects the phase of each one of the input LKM samples, having a f_CK (or optionally a f₀) frequency. The DPD translates each lkm-th sample to the corresponding central frequency of f_Slkm=[(l- l) +k/K]f₀. The DPD phase error is equal to an ε as small as requested (programmable) and proportional to the corresponding ratio Ω_sklm/Ω_ck.

The output signal (I e Q) is sent to a LKM Polyphase filters bank. Each LKM filter is built by three stages in cascade that works at a frequency clock of f_ck:

• • 16229 block of FIG. 9 has a function of properly decimating each lkm-th component of the signal;
  • 16230 stage that introduces a specific incremental azimuthal phase delay l [1-(K-k)/K] (m/M)2π on each orthogonal lkm-th signal;
  • 16231 block produces a chirp compression with a assigned slope of μ=π[(l-1)+k/K]f₀/T₀, and defined pulsed response.

The last two cascade blocks (16230 e 16231) together realize a convolution (matched Filter) that selects each lkm-th component of the orthogonal LKM ones. The sequence order of these stages can be optionally changed.

The LKM Polyphase filters bank can be realized merging the 16229, 16230 and 16231 stages in a single decimation—compressor bank made up of one Polyphase filter that processes the LKM modes together (all or some of them).

The 16231 outputs are then combined by L combiners 16232 (optionally only 1), grouping them conveniently based on the corresponding l, k and m indexes. Each combiner sums the related KM outputs, having l index, to rebuild the l-th signal m_l(t) used as the input of 16233 stage. 16233 stage is a bank that executes a Fourier transform, FFT. Here (FIG. 9) all the L m_l(t) are finally rebuilt (individually or simultaneously) in a digital requested format.

The L combiners 16232 stages can be integrated with the FFT bank 16233 to realize a single bank

Output Interface 17000 of FIG. 1

The output interface 17000 transforms at a frequency of f_CK, with a process corresponding to the one described above for the 11000 blocks of FIG. 1.

The L outputs requested (optionally a single output) coming out from 16233 block are sent to 17034 stage of FIG. 10. Here are converted into the proper standard format required. In the case at hand, for the sake of simplicity, an optional 17034 modulator is in place, and a D/A converter (17035 block) follows in cascade.

After a proper 17137 follows.

The 17137 block outputs of FIG. 10 are the rebuilding of the m_l(t) expressions, which are the input of HSCS device, subject of the invention (FIG. 1 or FIG. 2.)

Stepped Chirp Continuous Shifted Phase Case: Detailed Implementation Description

Input Interface 11000 of FIG. 1

All that has been previously described for the Continuous Chirp Continuous Phase case (see the detailed blocks 11001-11007 in FIG. 2) is valid also for the Stepped Chirp Continuous Phase one.

Complex Generator (I and Q) Expander 12100 in FIG. 1

System generator is implemented in this case with a bench of L*K*M D.D.S. generator devices (see the implementation block diagram of FIG. 3 where L=3, K=1, L=4).

FIG. 5 shows a specific detailed development of a lkm-th single elementary DDS generator.

The lkm-th “Register data control” related to each l-th m_l(t) output signal of 11007 block of FIG. 2, simultaneously enter the registers 12108 and 12109 blocks of FIG. 5, of the corresponding DDS having l, k and m indexes. As an alternative, the generator control data can be directly supplied as input of the generator registers without going through the interface block.

Each of the 12100 generators (see FIG. 3) is realized as a D.D.S. and it is highlighted in FIG. 5. The l-th control data corresponding to the m_l(t) signal, coming from the input interface, entering the HSCS system invention, is essential to digitally synthesize the S_l(t) signal with an assigned frequency f_RFl. S_l(t) is composed by a f₀ carrier and m_l(t) modulating, with any appropriate modulation required, according to the specific application of the invention. In the example here the S_l(t) carrier is generated as a single side band, SSB.

The specific carrier information lkm-th “Register control data” assigned by the invention are collected by 12108 block, here e.g. the “SSB delta phase”.

The “chirp delta phase” specific information lkm-th “Register control data”, assigned by the invention, is collected by 12109 block. The “Register control data” is used to synthesize the lkm-th chirp signal into the DDS of the invention.

Each lkm-th chirp signal is f₀ correlated and it is produced by a S_l(t) carrier having a frequency of f_RF defined by the invention for a period T=T₀.

In 12109 stage the following values are set:

• • slope μlkm=π[(l-1)+k/K](f₀-f_BBl/T₀
  • initial phase ϕ_loss=0

This chirp modulation causes S_l(t) a frequency linear shift of Δf_lkm=[(l-1)+k/K] (f₀-f_BBl).

The value of Δf_lkm is established by HSCS invention method, and e.g. it is computed using a MatLab code program.

The whole process of direct digital synthesis, is implemented by the invention, providing a clock frequency of f_cK, the same one of all LKM expressions of HSCS, S_lkm.

12108 and 12109 stages outputs enter the combiner (see stage 12110 of FIG. 5).

The 12110 block output of data added together are sent to 12111 block, which represents the invention phase accumulator.

Therefore, based on the data provided by 12108 and 12109 stages, each lkm-th phase accumulator 12111 outcomes the incremental phase value to be sent to 12113 block which is a “look up table”.

For every clock pulse, stage 12113 synthesizes a pair of digital sample, a real part I and an orthogonal imaginary part Q, of an assigned complex periodic function.

Please note that the accumulator input data bit number is N (where N is conveniently selected in order to achieve the resolution on the basis of the assigned frequency).

In turn, stage 12111 has a data bit number=P≤N as output.

The pair related to the 12113 block output digital signal S_lkm=S_lkm[f_RF, ψ_lkm(lf₀; lϕ; m)], enters 12115 block.

12115 block is a linear phase modulator that produces the lkm-th I and Q pair f₀ related.

The lkm-th pair is the sub mode component having 3LKM unit vectors of the sequence.

The stage 12115 performs an algebraic model multiplier tuned to f_RF. Each element of the I & Q pair is auto-correlated by the resonant stage 12115. In the stage 12115 each one of the two identical replicas of each I & Q pair, entering the two input stage, is modulated by other simultaneous one.

The I and Q pair output from 12115 block enters a digital low-pass filter FIR 12116. For example, a filter with f_1dB=f₀-f_BBl[(l-1)+k/K]. is in charge of harmonic suppressing having f>(f₀-f_BBl)[(l-1)+k/K] frequency, which is typical of the l-th sequence couple I and Q.

The 12115 output represents a direct synthesis of a signal, S_lkm(t), linearly modulated both in phase (linear Shift) and in frequency (chirp). S_lkm(t), generated by the invention, is provided with a real part I and with an imaginary part Q squared. S_lkm(t) is related to f₀ frequency and turns out to be orthogonal to all the other (LKM-1) similarly synthetized by the invention (FIG. 3).

The output generated by the two identical filters (I and Q) 12116 are synchronous signals (apart from a little phase error ε proportional to f_BBl/f₀) algebraically represented by a linear combination having lkm order compared with a fundamental frequency f_RFl.

Complex Under-Sampler 12200 of FIG. 1

All that has been previously described for the Continuous Chirp Continuous Phase case is valid also for the Stepped Chirp Continuous Phase one (see the FIG. 5).

Emitter Transducer 13000 FIG. 1

All that has been previously described for the Continuous Chirp Continuous Phase case is valid also for the Stepped Chirp Continuous Phase one (see the FIG. 1)

Field, 14000 of FIG. 1

All that has been previously described for the Continuous Chirp Continuous Phase case is valid also for the Stepped Chirp Continuous Phase one (see the FIG. 1)

Collector Sensor 15000 of FIG. 1

All that has been previously described for the Continuous Chirp Continuous Phase case is valid also for the Stepped Chirp Continuous Phase one (see the FIG. 1)

Complex Extractor 16000 of FIG. 1, HSCS_L⁻¹

All that has been previously described for the Continuous Chirp Continuous Phase case is valid also for the Stepped Chirp Continuous Phase one (see the FIG. 8 and FIG. 9)

Output Interface 17000 of FIG. 1

All that has been previously described for the Continuous Chirp Continuous Phase case is valid also for the Stepped Chirp Continuous Phase one (see the FIG. 10)

Stepped Chirp Stepped Shifted Phase Case: Detailed Implementation Description

Input Interface 11000 of FIG. 1

All that has been previously described for the Continuous Chirp Continuous Phase case (see the detailed blocks 11001-11007 in FIG. 2) is valid also for the Stepped Chirp Stepped Phase one.

Complex Generator (I and Q) Expander 12100 in FIG. 1

System generator is implemented in this case with a bench of L*K*M D.D.S. generator devices (see the implementation block diagram of FIG. 3 where L=3, K=1, L=4).

FIG. 6 shows a specific detailed development of a lkm-th single elementary DDS generator.

The “Register control data” related to each l-th m_l(t), output of 11007 block of FIG. 2, simultaneously enter the registers (see 12108 and 12109 blocks of FIG. 6) of the corresponding DDS having l, k and m indexes. As an alternative, the generator control data can be directly supplied as input of the generator registers without going through the interface block.

Each of the 12100 generators (see FIG. 3) is realized as a D.D.S. and it is highlighted in FIG. 5. The l-th control data corresponding to the m_l(t) signal, coming from the input interface entering the HSCS system invention, is essential to digitally synthesize the S_l(t) signal, with an assigned frequency f_RFl, having f₀ carrier and m_l(t) modulating, with any appropriate modulation required, according to the specific application of the invention. Here in the example, the S_l(t) carrier is generated as a “single side band”, SSB.

The “Register control data” specific for the “SSB delta phase” information, assigned by the invention, are dedicated to the invention and are collected by 12108 block. They are used to synthesize a S_l(t) carrier within the DDS of the invention.

The “Register control data” specific of the “chirp delta phase” information, assigned by the invention, are dedicated to the invention and are collected by 12109 block. The “Register control data” are used to synthesize the lkm-th chirp signal into the DDS of the invention.

Each lkm-th chirp signal is f₀ related and it is produced by a S_l(t) carrier having a frequency of f_RF defined by the invention for a period T=T₀.

In 12109 stage the following values are set:

• • slope μlkm=π[(l-1)+k/K](f₀-f_BBl)/T₀
  • initial phase ϕ_loss=0

This chirp modulation causes S_l(t) a frequency linear shift of Δf_lkm=f(l-1)+k/K] (f₀-f_BBl).

The value of Δf_lkm is established by HSCS invention method, and it is computed using a MatLab code program.

The whole process of direct digital synthesis, is implemented by the invention, providing a clock frequency of f_CK, the same one of all S_lkm LKM expressions of HSCS.

12108 and 12109 stages outputs enter the combiner (see stage 12110 of FIG. 6).

The 12110 block output data are added together and are sent to 12111 block, which represents the invention phase accumulator. Therefore, based on the data provided by 12108 and 12109 stages, each lkm-th phase accumulator 12111 outcomes the incremental phase value to be sent to 12112 block.

12112 block collects the “phase shift Register control data” too, which are dedicated to the specific linear phase shift modulation assigned by the method. They are used to synthesize the lkm-th phase shift modulation of the carrier S_l(t) f₀ correlated signal with the specific phase shift, Δϕ_lkm, into the DDS of the invention. The invention defines the lkm-th azimuthal incremental phase shift value. E.g. in the case considered in FIG. 5, it is:

Δϕ_lkm=-[(l-1)+Δϕ_lk](m/M)2π=-l[1-(K-k)/K](m/M)2π

where Δϕ_lm and Δϕ_lmk are related to the instantaneous phase linear term of the lkm-th component S_lmk(t) and both are computed by the invention method (e.g. using a Matlab code program).

The 12112 block sums the data related to the output 12111 block with the one collected by its “phase shift Register” section.

The 12112 block output data are added together and are sent to 12113 block which is a “look up table”.

For each clock impulse, stage 12113 synthesizes a pair of digital samples, one for the real part I and the other for the orthogonal imaginary part Q, of a corresponding digital complex signal of an assigned complex periodic function.

Please note that the accumulator has a number of bit data N as input from registers (where N is conveniently selected in order to achieve the resolution on the basis of the assigned frequency).

In turn, stage 12111 has a number of bit data=P≤N as output.

Stepped Frequency Stepped Shifted Phase Case: Detailed Implementation Description

Input Interface 11000 of FIG. 1

All that has been previously described for the Stepped Chirp Stepped Phase case (see the detailed blocks 11001-11007 in FIG. 2) is valid also for the Stepped Frequency Stepped Phase one.

Complex Generator (I and Q) Expander 12100 in FIG. 1

All that has been previously described for the Stepped Chirp Stepped Phase case case is valid also for the Stepped Frequency Continuous Phase one (see the FIG. 3 and FIG. 7) considering K=1 or K≥1 and M=1.

Complex Under-Sampler 12200 of FIG. 1

All that has been previously described for the Stepped Chirp Stepped Phase case case is valid also for the Stepped Frequency Continuous Phase one (see the FIG. 7).

Emitter Transducer 13000 FIG. 1

All that has been previously described for the Continuous Chirp Continuous Phase case is valid also for the Stepped Chirp Continuous Phase one (see the FIG. 1)

Field, 14000 of FIG. 1

All that has been previously described for the Continuous Chirp Continuous Phase case is valid also for the Stepped Chirp Continuous Phase one (see the FIG. 1)

E.g., for the Electromagnetic propagation case, the 13000 output expression is:

{EH}=F{Re[G_c(λ₀)HSCS_LM)]+αjImm[G_c(λ₀)(HSCS_LM)]}

where:

α=1 if the circular or elliptic {EH} field polarization

α=0 if linear {EH} field polarization

Collector Sensor 15000 of FIG. 1

All that has been previously described for the Continuous Chirp Continuous Phase case is valid also for the Stepped Chirp Continuous Phase one (see the FIG. 1)

Complex Extractor 16000 of FIG. 1,

All that has been previously described for the Continuous Chirp Continuous Phase case is valid also for the Stepped Chirp Continuous Phase one (see the FIG. 8 and FIG. 9)

Output Interface 17000 of FIG. 1

All that has been previously described for the Continuous Chirp Continuous Phase case is valid also for the Stepped Chirp Continuous Phase one (see the FIG. 10)

Analog Wide Frequency Bandwidth Power Transferring and Harvesting: Detailed Implementation Description

Input Interface 11000 of FIG. 1

All that has been previously described for the Continuous Chirp Continuous Phase or Continuous Chirp Stepped Phase or Stepped Chirp Stepped Phase or Stepped Frequency Stepped Phase cases (see also the detailed blocks 11001-11007 in FIG. 2) is valid also for the Analog wide frequency bandwidth power transferring one.

Complex Generator (I and Q) Expander 12100 in FIG. 1

All that has been previously described for the Continuous Chirp Continuous Phase or Continuous Chirp Stepped Phase or Stepped Chirp Stepped Phase or Stepped Frequency Stepped Phase cases is valid also for the Analog wide frequency bandwidth power transferring one (see the FIG. 3 up to FIG. 7) considering K=1 or K≥1 and M=1.

Complex Under-Sampler 12200 of FIG. 1

All that has been previously described for the Continuous Chirp Continuous Phase or Continuous Chirp Stepped Phase or Stepped Chirp Stepped Phase or Stepped Frequency Stepped Phase cases is valid also for the Analog wide frequency bandwidth power transferring one (see the FIG. 7).

Emitter Transducer 13000 FIG. 1

All that has been previously described for the Continuous Chirp Continuous Phase or Continuous Chirp Stepped Phase or Stepped Chirp Stepped Phase or Stepped Frequency Stepped Phase cases is valid also for the Analog wide frequency bandwidth power transferring one (see the FIG. 1)

Field, 14000 of FIG. 1

All that has been previously described for the Continuous Chirp Continuous Phase or Continuous Chirp Stepped Phase or Stepped Chirp Stepped Phase or Stepped Frequency Stepped Phase cases is valid also for the Analog wide frequency bandwidth power transferring one (see the FIG. 1).

E.g., for the Electromagnetic propagation case, the 13000 output expression is:

{EH}=F{Re[G_c(λ₀)HSCS_LM)]+jαImm[G_c(λ₀)(HSCS_LM)]}

where:

α=1 if the circular or elliptic {EH} field polarization

α=0 if linear {EH} field polarization

Collector Sensor 15000 of FIG. 1

All that has been previously described for the Continuous Chirp Continuous Phase or Continuous Chirp Stepped Phase or Stepped Chirp Stepped Phase or Stepped Frequency Stepped Phase cases is valid also for the Analog wide frequency bandwidth power r transferring one (see the FIG. 1).

Complex Extractor 16000 of FIG. 1, HSCS_LM⁻¹

Continuous Chirp Continuous Phase o Continuous Chirp Stepped Phase or Continuous Chirp Stepped Phase or Stepped Chirp Stepped Phase or Stepped Frequency Stepped Phase cases is valid also for the Analog wide frequency bandwidth power transferring one (see figures from 11 up to FIG. 14)

Output Interface 17000 of FIG. 1

All that has been previously described for the Continuous Chirp Continuous Phase or Continuous Chirp Stepped Phase or Stepped Chirp Stepped Phase or Stepped Frequency Stepped Phase cases is valid also for the Analog wide frequency bandwidth power transferring one (see the FIG. 10)

Description of a Code Program Tool for Invention Method Parameters Computation

The program (eg. in Matlab code) used to compute the method parameter, according to the invention, is valid for all the implementation environments considered above (see FIG. 15 up to FIG. 18). It simulates the mathematical model of the invention. It makes possible to set and optimize the invention implementation design parameters depending on the the value input: L, K, M, f₀ and f_CK.

Following the computed system HSCS design parameters, according to the invention: the elementary spherical spatial phase increment, ψ₀(M; f₀; ϕ₀); all the lkm-th vectors coefficients; all the lkm-th spherical spatial phase increments, ψ_lkm(lf₀; lϕ; k; m); all the lkm-th azimuthal spatial phase increments, Olk_m; all the lkm-th tilt spatial phase increments, θ_lkm.

Furthermore, if required, the tool emulates the complete emitter subsystem HSCS_LKM. For instance, for the electromagnetic case, the tool computes the Poynting Vector and the specific OAM.

The single resulting complex signal, generated by the invention, identifies a specific three-dimensional (3-D) complex vector space and 3LKM unit vectors, algebraically represented by the invention. e.g. as a general linear combination like:

PP ⁰(jΩ)=Σ^L_l=1Σ^K_k=1Σ^M_m=1S_lkm=Σ^L_l=1Σ^K_k=1Σ^M_m=1C_LMSlkm exp[-jψ_lkm(lf₀; lϕ; k; m)].

In the electromagnetic (EH) case eg., the tool computes the Poynting Vector in every related complex vector space EH field point, then in every propagation axis point too. That is possible simply solving the Maxwell equations with respect of the block 12200 (FIG. 3) output expression and imposing the invention specific surrounding conditions to obtain G_e(λ₀)*HSCS_LKM. Both E and B fields belong to the algebraic function PP⁰(jΩ) (like the linear combination defined before). The generated E(PP⁰(jΩ)) and B(PP⁰(jΩ) fields represent the required general complex radiation TEM, non-TEM or Divergent assigned modes by the invention. Once computed, or detected by the invention method, the x, y, z values related to E(PP⁰(jΩ) and B(PP⁰(jΩ), then it is possible, as well known in Physics, for every point (∀P also on the propagation axis) to compute: the Poynting Vector, the specific Linear Moment and OAM values.

Properties of the Invention

The HSCS of the invention system generates an 3LKM dimensional algebraic Hilbert space with 3LKM unit vectors. The complex vector space is characterized by an L independent limited occupied frequency bandwidth, BW₀.

In the most general case, the HSCS complex vector space generated by the invention is characterized by 3LKM complex component vectors. All of them are:

each other orthogonal

• • related to a single central frequency typical of the resultant apart a smallest phase error ε belonging to Ω_sl/Ω₀ ratio.

Each Complex Vector is Provided with:

• • • a spatial spheric phase ψ_lkm(lf₀; lϕ; k; m) assigned by the invention and to which corresponds a characteristic spatial azimuthal phase Δϕ_lkm and a tilt spatial pahse θ_lk. Those phases are detected by HSCS depending exclusively on l, k and m indexes (in addition to f₀, and f_BBl).
    • The partial derivative of the first order of the spatial speric phase both for t and z (wave propagation axis) turns out to be null, in other words it does not belong to t not to z, actually ∀t and ∀z ∂ψ_lkm(lf₀; lϕ; k; m)/∂ z=costant and furthermore ∂ψ_lkm(lf₀; lϕ; k; m)/∂ t=costant (if the event occurs in an homogeneous medium).
  • ψ_lm(lf₀; lϕ; k; m) is not variant along propagation axis z as well as with respect to time.

Once provided specific f_cK, L, K and M which are assigned by the invention method with the requested f₀, the HSCS system allows:

• • No Processing Gain losses
  • Processing Gain exclusively proportional belong to L, K and M.
  • Channel Capacity gain proportionally increasing with the L, K and M values.

Such gains occur because the system is capable to process also the longitudinal component power (or informative) contributions (originated by not zero value of ∂E/∂z as well as ∂H/∂z). As well known in technical literature the last mentioned properties should be impossible to achieve with former invention method.

The invention HSCS develops a field that, e.g. in the electromagnetic, is equipped with assigned OAM for any required value of:

• • L Number, and/or central frequency, and/or occupied frequency band BW_O, and/or transducer physical dimension and topology, and/or sensor in use.

Such OAM is stable, invariant and differs from zero independently of:

• • A Time
  • Distance between the two blocks 13000 and 15000 of FIG. 1.

The actual channel capacity (setting, for instance for the easiest case: K=M=1) provided by the invention method HSCS, C_HSCS, as well known in technical literature, is computed by the expression:

C _HSCS=2(L)BW₀.

Provided the same available bandwidth, BW₀, the channel capacity C_HSCS is greater than the one C_BWo, allowed by former invention systems. As a matter of fact,

C _HSCS >C _BWo=2BW₀, furthermore Lim_{per L→∞}C_HSCS=∞.

The last expression guarantees that invention method generates a whole defined and algebraically continuum system, which is valid in an unlimited complex scalar as well as vector space.

The HSCS structure, both algebraic and implemented, does not present any singularity species, neither at its extreme, nor along its boundary.

Therefore, HSCS generated by the invention is a continuum at the infinite and along the assigned complex vector space longitudinal axis. On the contrary, in the former invention methods, it is impossible to achieve that last property.

Such complete HSCS system generated by the invention, transfers the power which is allocated using L orthogonal vectors. The L orthogonal vectors frequency bandwidth BW_RF of complex vector resultant is equal to the maximum independent one, BB_', related to the L input signals 1=1, . . . , L, BW_RF=max[BB_l].

Therefore, the L≥1 orthogonal complex mode resultant occupies a frequency bandwidth, BW_occupied≤BW_RF≤BW₀, while the utilized actual bandwidth, BW_effective, effective, according to the invention method is always greater than the resultant one BW_occupied:

BW _effective=Σ^L_l=1BB_Sl>BW_occupied

It is to be underlined that such performances, which would be impossible to achieve without our invention, increase the efficiency of:

• • the Transfer channel
  • the Store space
  • the Signal
    • sent out across a link
    • Radiated towards any natural physical phenomena (i.e., atmospheric or climatic ones or else) in order to create an interference leading to an adequate control action.

Furthermore the invention method implementation, as high efficiency 2^nd order wide frequency bandwidth power channeling and energy harvesting directly generates voltaic energy. It directly converts the available harvested energy, the one related to any TEM+DEM 1^st order linear moment (LM) and Angular moment (AnM) (Electromagnetic, acoustic, gravitational, thermodynamics, pressure, optic, mechanic) detected along the wave propagation axis. As well-known AnM spherical spatial 2^nd order wave beams power is about 80 dB greater than LM azimuthal spatial 1^st order waves or wave beams.

The invention method allows to direct detect a Black Hole Gravitational signals with a 2^nd order visibility degree >1/2. Usually the gravitational signal intensity is about 10⁻²². As well-known in literature, the complex mutual (in a space-time domain) coherence measurement of such Gravitational wave requires to analyze an extremely short bandwidth (in the ideal case it is =0), and an instantaneous time resolution, τ, that in the most favorable case, turns out to be inversely proportional to the square frequency observed bandwidth value (BW₀) or the wave coherence period, τ₀. Such direct measurement especially when τ>>τ₀>1/BW₀ could be impossible using any former method, because its 1^st order visibility degree <<1/2.

The invention method allows to allocate or to measure any generated not zero resultant Poynting vector value along propagation axis of any twisted swirling field equipped with complex TEM, non-TEM, or divergent mode too.

The complex vector space is immune from any possible interferometry error due to destructive planar (2-D) interferences. Actually each of all its orthogonal complex 3LKM-D vectors components is equipped with one characteristic orthogonal spherical (3-D) spatial (time invariant) phase, ψ_lkm(lf₀; lϕ; k; m).

The invention method has three degrees of freedom available to set ψ_lkm(lf₀; lϕ; k; m). Such degrees of freedom are: the fundamental frequency f₀; the lkm-th frequency mode lf₀=f_lkm (or tilt plane θ); the azimuth spatial phase lϕ=Δϕ_lkm (or azimuth plane ϕ). Then, because of its 3-D spatial phase, ψ_lkm(lf₀; lϕ; k; m), the invention method is immune from destructive planar phase ambiguity.

Such design three degrees of freedom available are the ones requested, and sufficient, to guarantee the generation of the LKM complex vectors space component being:

• • 3-D spatial orthogonal;
  • Each correlated whatever required frequency value f₀;
  • Each extractable without processing method losses.

On the contrary, the former invention methods and/or systems have less degrees of freedom available (e.g. at the most only: f₀ and lϕ): thus, they are not able to eliminate planar phase ambiguity with any f₀ and lϕ values because they lose the longitudinal component.

The invention method, HSCS, is in the same way efficiently applicable to measure near as well far distance fields, because of its planar interferometry error immunity.

The operator, according to the invention, develops both linear and quadratic, spherical wave time—spatial (mutual) phases equation terms. Such terms are originated setting or detecting both spherical wave phases first and second time space derivatives. They contribute to generate the required complex or eventually full imaginary 3LKM-D TEM, non-TEM, or Divergent spherical (3-D) modes which satisfies the wave equation with a “Paraxial approximation”. Usually former methods neglect such terms as well as the longitudinal modes components.

The invention HSCS performs a greater approximation degree because it processes the complex or full imaginary OAM mode which are neglected or ignored by all the former invention methods. In Physics literature, such specific modes are well known as non-TEM, or Divergent mode [Electromagnetic (DEM) or Electrical (DE) or Magnetically (DM) Polarization].

As well known in technical literature, all the last-mentioned properties should be impossible to be achieved at once with the former invention methods and/or systems (see ref. [10] and [13]).

HSCS is applicable for any periodic complex function with frequency f=f₀ which is included into a Low Frequency and Millimeter Wave (MMW) range interval.

HSCS is applicable for any guided or free light or laser light (e.g. Laguerre Gaussian Beam model) propagation space.

HSCS is applicable for any guided or free EM wave propagation Rx/Tx system, which TEM field is or not equipped with OAM and whatever polarization (e.g. “twisted and/or skewed” and/or Rx polarization different from Tx one).

HSCS is applicable for any guided or free EM wave propagation Rx/Tx system, which non-TEM TE or TM complex field mode are equipped with longitudinal not zero E or M component.

HSCS is applicable for any guided or free EM wave propagation, which divergent complex or full imaginary field mode are equipped with different Rx polarization kind compared with the Tx one.

HSCS is applicable for any “base band” electric signal.

HSCS is applicable for any signal which has been whatever modulated. E.g. as following:

• • Analog modulation: AM, PM, FM;
  • Digital modulation: QPSK, MSK, and so on . . . ;
  • TDM, FDM;

HSCS develops systems for: information signal OAM modulation and de-modulation (mode-demod), from both base band and already up converted.

HSCS develops systems for: Spread Spectrum (SS) modulations, e.g. “Direct sequence” (SS-DS), see ref. [1].

HSCS develops a complex scalar function describing the distribution of a twisted field amplitude which satisfies the wave equation applicable to light and/or laser beam. It could be one of the following:

• • Pseudo Gaussian
  • Spherical
  • Hyperboloid

Such distribution is developed independently by the topological characteristics of the specific sensor.

The method of the invention, e.g. by single contemporaneously observation of the frequency wide spectrum of each one of the array antennas, develops a three-dimensional, 3-D, Early universe “Tomographic interferometer process” with 3LKM unit vectors. Such Early universe quadratic order radio telescope observation turns out to be without planar (azimuth) phase error and ambiguity. Therefore, it is full compliant with the requirements of the innovative quadratic models used in the Astrophysics research post processors new generation, which elaborate the data from radio telescope observation.

The method of the invention, besides the space requirement, power consumption and costs decrease, empowers the radio telescope performances efficiency. On the contrary all the last-mentioned properties should be impossible to achieve with former methods and/or systems preceding the invention.

Invention Advantages and Applications Following some examples of invention applications:

• 1. Application example of Collector subsystem for detection, observation and measure of radiation coming from any evenly bodies, see ref. [13], which are far hundreds of billions of head and shoulders above us. Here as e.g. is considered the invention application specific case in Square Kilometer Array (SKA) program. In that case, HSCS according with the invention method, uses its ability to correlate and process, in a single concentrated local processor, both the linear and orbital angular momentum in order to extract the requested tomography information data from the whole early universe radiation frequency spectrum (for SKA mission: from the low “band one” 50 MHz up to the highest “band five” frequency, 25 GHz), which is detected by whatever single assigned local % tuned antenna of the SKA antennas arrays stations. On the contrary all the lasts mentioned properties should be impossible to achieve with former SKA methods and/or systems preceding the invention. λ₀ is assigned according the invention method. In this case the invention method performs a 3LKM unit vectors interferometry process which allows an early universe tomography without planar phase (both azimuth and tilt) or planar ambiguity detection (destructive multi path or Doppler, etc., that are eventually highlighted by grating lobes increasing or by a very low mutual visibility degree value, <1/2).

Such 3-D “Early Universe Tomography” approximation order, according with the invention, is quadratic. That is why it is compliant with the innovative last generation quadratic numerical simulator research models, which are already available to elaborate the data from astrophysics SKA observations to reconstruct a required coherent early universe scale. In that example the method of the invention, in addition to the decreased dimension, power consumption and costs, is capable to improve the SKA radio telescope performances efficiency. The efficiency improving factor is proportional to the following values:

• • LKM
  • A Saved Antennas (not used) composing the array which, therefore, might be used for other missions or services.

On the other end all the last mentioned properties should not be achieved with former methods and/or systems preceding the invention.

• 2. Application example of emitter subsystem e.g. for weather and hurricanes forecasts, prevention and control. Here is a selection of a suitable number of frequencies, e.g. N (N is a integer number ≠ ∞) f_n (with n=1, . . . , N). N and f_n are selected based on statistical analysis of previous opportune climatic observations, preferably developed according to the invention Extractor sub system (above) too.

An emitter sub system according with the invention here develops a required radiation composed by LKM (or NLKM) pressure waves. Each one is characterized by a central frequency value f₀ (or f_n) and opposite phase value respect to the phase one of the LKM (or NLKM) statistical previously observed e.g. with the invention method. The intensity, the central frequency, and the spatial phase characterizing each LKM (or NLKM) component radiation conveniently emitted by HSCS according with the invention allows a softening action on the physic natural components before statistically selected which possibly are present. Such radiated LKM (or NLKM) components may invert the natural event (here is considered e.g. Hurricane event) by softening its action depending on the values assigned to N, L, K, and M which are separately or assembly used to implement the N subsystems, according with the invention. Such implementation method is the same of electromagnetic case, except for introducing a specific electric/pressure signal transducer rather than electromagnetic/pressure one.

• 3. Application example of emitter subsystem e.g. emitter and extraction system for Satellite Communication which is equivalent to the one described to implement EM one with circular polarized transmitter and receiver antennas. The possible suitable different satellite station signals LKM number to be assigned, compared with the ground control one, belongs by specific compatibility and requirements of the spaceship or space orbited station typology. Therefore, it is possible reduce wasteful and eliminate the not required receiving and transmitting satellite channels. The invention method improves sensitivity of former systems preceding the invention too.
• 4. Setting LKM high value (not acceptable in former systems preceding the invention) the method allows to detect the complex OAM mode included in weak fundamental with real mode far distant radiations which usually should be neglected or lost (as e.g. in Astrophysics radio telescope observations)
• 5. The invention allows to assign the gauge transverse related to the spherical OAM of the complex field. It allows to improve the energy related the emitted and/or extracted linear moment by a real TEM mode planar system, according with the invention.
• 6. The invention is applicable to implement systems or models both analogue and digital e.g. in: Physics; Mechanics; Dynamics; Acoustics; Geology; Clinics; Metrology; Interferometry; Meteorology; Astrophysics; Electronics; Radar; Navigation Assistance Services (Space, Atmospheric, Ground, Naval, Satellite); Communications; energy harvesting; Optoelectronics; Bio-Medical and Bio-Engineering; Monitoring; Security; Civil; industrial, military, data transmission; numerical data and/or video and/or audio signals compression and spreading in any multi points link multi ends emitters and users.
• 7. The invention allows observation and measurement of the partial second time derivative of all the variable describing any physic complex event which occurs in any environment. On the contrary all the lasts mentioned properties should be impossible to achieve with former methods and/or systems preceding the invention.
• 8. Furthermore, the invention method allows opportunities to open new applications fields e.g. the following:
  • I. Nuclear Physics;
  • II. Gravitational Physics;
  • III. Energy harvesting;
  • IV. Nuclear Fusion “clean energy” generation;
  • V. Sustainable nuclear not radioactive nuclear energy;
  • VI. Astrophysics see ref.[13];
  • VII. Medicine, e.g. in DNA particles computation;
  • VIII. Wing Power in nautical as well in aeronautics (e.g. that occurring in high vortex state or located into so called “Wing vortex extremity”).
  • IX. Dynamics Fluidity of ships outline in Navigation as well in Aeronautics;
  • X. Eco environment and climatology: prevention measurements and control of macro and micro climatic and meteorological phenomena due to vortex motions. E.g.: pressure wave and air masses or magmatic fluid or solid mass motions (earthquakes, Vulcans, landslide, tornado, whirlwind, etc.)
  • XI. Geology: pressure quadratic variation measuring in order to prevent ground earthquakes and landslide;
  • XII. Data compression to memories storing;
  • XIII. Communication: channelization and storage of any analogue and digital data information signal as well as the resultant channelization of a multi sub channelization components developing according with the invention both physically and numerically.
• 9. Application example as electric wide frequency bandwidth wave beam Power generator, transmission, energy harvesting and converter, HSCS system is implemented using: the Analogic wide frequency bandwidth HSCS_LKM transform operator configuration of the invention as power transmitter shown in figures from 1 to 7, and the Analogic wide frequency bandwidth HSCS_LKM⁻¹ as power extractor and voltaic converter shown in figures from 10 to 15.
• 10.Application example as electric Analogic wide frequency bandwidth Power extractor and converter, HSCS system is implemented using simply: the Analogic HSCS_LKM⁻¹ as power extractor and voltaic converter shown in figures from 10 to 15. Such configuration is valid to extract and convert and or measure any typology of linear (LM) and/or quadratic (2^nd order AOM) energy (electromagnetic, and/or gravitational, and/or thermodynamic, and/or mechanics, and/or optic, and/or etc. . . . ). Such HSCS_LKM⁻¹ invention implementation named 2^nd order -Wave-Volt converter allows to an 80 dB efficiency gain value greater than the former method one.
• 11.Application example as 2^nd order Bio-power frequency meter and/or 2^nd order Bio-impedance meter to perform any kind of human health check about: blood, hart, lang, DNA, RNA, tumor or cancer cells, etc. . . . In such application, according to the invention are to use both the invention operators: the transform operator HSCS_LKM (as reference radiation generator) and the anti-transform operator HSCS_LKM⁻¹ (as power spectrum meter).
• 12.Application example as 2^nd order Seismic sensor, according to the invention, is to use the anti-transform operator HSCS_LKM' (as power spectrum meter).
• 13.Application example s 2^nd order Thermodynamic sensor for example in Meteo phenomena control and/or prevention such as Hurricane, is to use, according to the invention, the anti-transform operator HSCS_LKM⁻¹ (as power spectrum meter). It allows a time spatial analysis of any environmental observed event complex 2^nd order 3-D mutual visibility.
• 14.According to the invention HSCS_LKM performs a radio electromagnetic health therapy system. It radiates any short bandwidth at any extremely high wave beam frequency at any time-space (period-distance) value.
• 15.Application example as 2^nd order gravitational time-space wave beam detector system for black hole analysis and studies applications. In such cases, as in Ligo (USA) or VIRGO (Italy), is to be used the full HSCS system composed by both the invention transform operator HSCS_LKM and anti-transform operator HSCS_LKM⁻¹. Such system performs a required time space high frequency and extremely short bandwidth wave beam radiation as well as detects the time space 2^nd order complex 3-D visibility of observed physic event
• 16.Application example as 2^nd order gravitational time-space wave beam direct detector system. As well-known gravitational wave beam intensity is about 10e-22 eV. Such value is extremely weak vs. space noise. In that case, according to the invention is to use the anti-transform operator HSCS_LKM⁻¹. It detects both the first and second time space derivatives of spherical (3-D) spatial phases (azimuth and tilt) related to the observed gravitational 3-D wave beam. It allows to extract a complex 2^nd order mutual visibility using any requested sample rate to independently analyze any frequency bandwidth in any coherence period (generally is requested a very long coherence period)

Invention Feasibility

The present invention feasibility occurs because of the following.

First of all, a suitable “very large scale integration (VLSI) technological state of art is already available which is useful to implement present invention method. E.g. innovative and consolidate architectures and microelectronic numerical synthesis techniques are already available to develop generalized form of the transfer function algorithms according to the invention. Such architectures are possible because the state of art of ASIC as well FPGA as well GPU technology.

In addition to that, there are available programmable and reconfigurable mixed signal ASIC technology tools innovative architecture and numerical technique in order to process very high speed (MMW) complex periodic functions and signals.

Invention Applicability Limits

The invention method generates and processes infinites Hilbert algebraic spaces, I valid for every L value integer number comprised in [1; ∞] interval, for every K value integer number comprised in [1; ∞] interval, for every M value integer number included in [1; ∞] interval. Obviously actual invention applicability limits belong to the trade off between implementation costs and advantages.

Furthermore, the trade off critical point belongs to application typology and environment.

• • A. E.g. in Emitter (Canalization) and/or Extractor (Universe evenly Bodies Radiations Detection and Measure) System Applicability Belongs:
• I. Acceptable LKM signal phase error value due to the respective clock signals mismatching phase.
• II. Specific implementation technology (ASIC, FPGA, GPU or other) used to develop the complex modulators (Cirp and linear phase shift). The implementation technology defines optimum clock frequency, data rate, frequency bandwidth, consumption and power dissipation values
  • B. Limits in Data Storing and Compression Belong:
• I. Specific implementation technology (ASIC, FPGA, GPU or other) used to develop the complex modulators (Cirp and linear phase shift). The implementation technology defines optimum clock frequency, data rate, frequency bandwidth, consumption and power dissipation values
• II. Trade-off between specific temporary and storing memory sizes requested in the different applications. Anyway the canalized data compressing and transferring requires data memories, then invention benefits are a result of a specific case by case trade-off. Furthermore, it is to consider in the receiver the possibility to select or not the interest required channels only.In both A and B below already it may be considered that the optimum LKM value is: LKM≤10¹⁵

Furthermore, e.g. in specific hurricane climatic control and prevention emitters subsystems(e.g.: Stepped Chirp and Continuous Linear Phase Shift of FIG. 5) it may be considered that the optimum LK value is LK≤10¹⁵ with M value is M≤∞.

BIBLIOGRAFIA

• 1. “Spread Spectrum Systems” Robert C. Dixon 1984 Ed John Wiley & Sons, Inc.
• 2. “Angular Momentum of Electromagnetic Radiation”—Autori: Johan Sjoholm & Kristoffer Palmer—UPTEC F07 056 Examensarbete 20 p April 2007—Autori: Johan Sjöholm Kristoffer Palmer
• 3. “Encoding many channels in the same frequency through radio vorticity: first experimental test.”—Venice, on Jun. 24 2011—Autori: Fabrizio Tamburinil, Elettra Mari, Anna Sponselli, Filippo Romanato, Bo Thidé, Antonio Bianchini, Luca Palmieri, Carlo G.—35131 Padova, Italy, EU.
• 4. “Beamforming: A Versatile Approach to Spatial Filtering”—IEEE ASSP MAGAZINE APRIL 1988—Autors: Barry D.Van Veen & Kevin M. Buckeley
• 5. “Angular momentum of light” A. M. Stewart Research School of Physical Sciences and Engineering, The Australian National University Canberra, Australia 0200. e-mail: andrew.stewart@anu.edu.au
• 6. “ELECTROMAGNETICS ” author : E. J. Rothwell (Michigan State University East Lansing, Mich.); Michael J. Cloud Lawrence Technological University Southfield, Mich.—CRC Press Boca Raton London New York Washington, D.C.
• 7. “Appunti di Campi elettromagnetici “S.Barbato
• 8. “Microonde” dispense Prof. M Farina DIE Bioingegneria UNI delle Marche
• 9. “Waves and Fields in Inhomogeneous Media Chew, W. C.”, Van Nostrand Reinhold, New York, 1990.
• 10. “Orbital angular momentum of light and the transformation of Laguerre-Gaussian laser modes”, Physical Review A, 45, 11 (2006), pp. 8185-8189 Author: L. ALLEN, M. W. BEIJERSBERGEN, R. J. C. SPREEUW, AND J. P. WOERDMAN,
• 11. “Classical Electrodynamics” J. D. Jackson professor of Physics, University of Illinois
• 12.“From Maxwell to paraxial wave optics” M. Lax, W. H. Louisell and W. B. MacKnight, Phys. Rev. A11,1365(1975). Melvin Lax Department of Physics, City College of New York, * New York, York 10031 and Bell Telephone Laboratories, Murray Hill, New Jersey 07974
• 13.“Italian SKA White Book”, June 2013, Editors: L. Ferretti & I. Prandoni—On behalf of SKA-Italy WG-Endorsed by Scientific Council-, INAF Press.
• 14.“Campi Elettromagnetici II” Fabrizio Frezza 2005

Claims

1. A method for canalizing and transferring a plurality of periodical physical signals represented in the time domain, t, by contemporary or not contemporary complex functions, SFl(t), with l=1, . . . L, and L is an integer number comprised in [1; ∞] each occupying an independent frequency band, e.g. BBl=[(f0-fBBl), f0], such as electrical, electromagnetic, gravitational wave, analog and/or digital, mechanical oscillations, pressure waves, optical signals, thermodynamic signals, into a single complex signal, resultant of L orthogonal components each l-th being in bijection with the respective SFl(t), correlated at a frequency f0 with a single occupied frequency bandwidth having width equal to the largest band among said bands BBl, BWRF=max[BBl], wherein the following steps are executed shown in the FIGS. 1 and 4: A. acquiring if required the physical signals SFl(t) and transforming them into electrical signals by means of one or more transducers;B. extracting from each electrical signal of the previous step the relevant information content as modulation function ml(t);C. performing the Fourier Transform of mi(t), obtaining the function ml(jΩ);D. generating, for each modulation function ml(jΩ), a corresponding periodic digital signal Pl(jΩ)=Sl(jΩ) having carrier frequency f0 and assigned modulation function ml(jΩ);E. generating, for each periodic digital signal Pl(jΩ), a pair of signals in quadrature;F. generating, for each pair of signals in quadrature of step E, K*M pairs of signals PPlkm(jΩ)=Sklm(jΩ), wherein Kand M are integer numbers comprised in [1; ∞] and k=1, . . . , K as well as m=1, . . . , M each pair being generated through the following steps performed subsequently in any order or at the same time: performing KAI chirp modulations, each characterized by a corresponding frequency increment Δflkm=[(l-1)+k/K](f0-fBBl), which is the same for each of the M with k varying, in a time interval ΔT=T0 at a clock frequency e.g. fc≥2LKM f0;performing M phase linear modulations for each of K chirp modulations; each of K*M phase linear modulations is characterized by a own phase increment Δϕ_lkm=[l-1)+l(k-1)/K+(lm/M)]2π with varying k and m, at a clock frequency e.g. fc≥2LKM f0;G. under sampling said pair of signals PPlkm(jΩ) at a frequency f0 thus obtaining a pair of canalized signals PP0lm(jΩ) which by construction have a spatial phase (3-D) orthogonal to each of the other (L * K*M-1) signal pairs and moreover they are correlated to f0;H. vector summing all the L*K*Mpairs of orthogonal PP0lm(jΩ) components thus obtaining a pair of LKM-dimension signals PP0(jΩ) occupying a frequency bandwidth BWRF≤f0;I. sending the LKM-dimensional signal PP0(jΩ) to a signals emitter-transducer tuned to f0; which performs whatever polarization.

2. The method according to claim 1, wherein step H is realized by a simple collection node of all the orthogonal signals produced in step G.

3. The method according to claim 1, wherein steps E and F are inverted.

4. The method according to claims 1, wherein steps E e/o F e/o G e/o H e/o I are performed at the same time.

5. The method according to claim 1, wherein the transducer of step I is a signals transmission media with unitary transfer function, eg. a cable connector.

6. A method for the extraction of a plurality of periodical physical signals represented in the time domain, 4 by contemporary or not contemporary complex functions SFl(t) with L is an integer number comprised in [1; ∞] and l=1, . . . L, each of which occupies an independent frequency band BBl=[(f0-fBBl), f0], such as electric, electromagnetic, digital, mechanical oscillation, pressure waves optical signals, starting from a single complex signal correlated to a frequency f0 occupying a frequency band whose width is equal to the largest band among said bands BBl, BWRF=max[BBl], wherein the following steps are performed, shown in the FIGS. 1 and 8: J. detecting by means of a tuned to f0 sensor, which performs whatever polarization, said complex single signal to obtain PP0(jΩ)=KSCSLM;K. applying a complex correlation, at frequency f0, to the signal detected by said sensor, thus obtaining LKM pairs of signals PP0lkm(jΩ)=HSCSLKM−1(Slkm);L. performing, for each one or single one values of l=1, . . . L, KM contemporary convolutions, at a clock frequency fck, of each signals pair PP0lkm(jΩ), the KM convolutions being configured to extract KM respective chirps with frequency shift Δf_lkm=[(l-1) +k/K](f0-fBBl) and a azimuth phase delay Δϕlkm=-l[1-(K-k)/K](m/M)2π in an interval T0, from said signals pair, obtaining corresponding pairs of signals with k=1, . . . K and m=1, . . . M for each l;M. summing the KMcomponents of each l value, PPl(jΩ), obtaining all the independent modulation functions ml(jΩ) from said single signal with l varying;N. performing a Fourier Anti-Transform of ml(jΩ) and thus obtaining corresponding time domain modulation functions mi(t);O. transforming if required ml(t) into SFl(t) by one or more transducer.

7. The method according to claim 6, wherein said single signal is the signal emitted by said emitter-transducer of signals tuned to f0, according to claim 1.

8. The method according to claim 7, wherein the polarization of step I is the same or is different from the polarization of step J.

9. The method according to claim 1, wherein in step F, the value of M and/or K and/or L is infinite, thus obtaining continuous chirps and linear phase modulation.

10. The method according to any claim 1, wherein the starting periodic physical signal set is subdivided into a plurality of periodic physical signals, to each element of the plurality being applied the steps of the method.

11. A computer program, comprising code tools set up in such a way that, when running on said computer, perform steps B to H according to the method claim 1.

12. A system for the canalization and/or extraction of a plurality of physical signals, wherein the system comprises: acquisition devices of a set of original physical signals;one or more transducers for the transformation of the original physical signals into electric signals;a computerized sub-system to process said electric signals according to steps B to H of the method according to claims 1, thus obtaining one or more final electric signals;one or more transducers for the transformation of said final electric signals into final physical signals;emission devices for the emission of said one or more final physical signals;