Constant current supply circuit

BACKGROUND OF THE INVENTION

The present invention relates to a constant current supply circuit for supplying constant current to an electric load via a bipolar transistor.

FIG. 5

shows a constant current supply circuit disclosed in the unexamined Japanese patent publication No. 5-60623. An electric load L has one end connected to a ground terminal and the other end connected to an emitter of a primary transistor Q

100

. The primary transistor Q

100

is an NPN-type transistor which controls the current supplied to the electric load L. A first resistor

101

, a second resistor

102

, a third resistor

103

, and a fourth resistor

104

are serially connected between a high-potential terminal VCC and a ground-potential terminal of a power source. A secondary transistor Q

200

, being an NPN-type transistor, has a collector connected to a connecting point of the second resistor

102

and the third resistor

103

, a base connected to a connecting point of the third resistor

103

and the fourth resistor

104

, and an emitter connected to the ground-potential terminal of the power source. A base of the primary transistor Q

100

is connected to a connecting point of the first resistor

101

and the second resistor

102

. A positive voltage terminal

105

, having an electric potential VD higher than the ground potential, is connected to a collector of the primary transistor Q

100

.

According to this conventional constant current supply circuit, the electric load L receives a constant current (i.e., load current) I from the terminal

105

via the primary transistor Q

100

. The electric load L has temperature characteristics in its resistance value R. To compensate such temperature characteristics, the relationship among resistance values of the first to fourth resistors

101

to

104

(especially, a resistance ratio of the third resistor

103

to the fourth resistor

104

) is determined in such a manner that a voltage E applied between both ends of the electric load L adequately varies in accordance with the temperature. With this setting, the load current I is maintained at a constant value irrespective of temperature change.

However, the above-described conventional constant current supply circuit has the following problems.

It is now assumed that TCRL represents a resistance temperature coefficient of the electric load L, Rtyp represents a typical resistance value of the load resistance R at a reference temperature Ttyp, and ΔT represents a temperature deviation from the reference temperature Ttyp.

Using the above, the load resistance R can be expressed by the formula Rtyp(1+TCRL×ΔT). The temperature characteristics of the load resistance R is Rtyp×TCRL×ΔT. In other words, the load resistance R causes a variation equivalent to Rtyp×TCRL×ΔT in response to the temperature deviation ΔT from the reference temperature Ttyp. Thus, the temperature characteristics of the electric load L varies in accordance with a change of the typical resistance value Rtyp of the load resistance R.

However, according to the above-described conventional constant current supply circuit, the resistance values of the resistors

101

to

104

are determined in such a manner that the voltage E applied between the both ends of the electric load L varies in accordance with the temperature so as to compensate the temperature characteristics of the load resistance R. Accordingly, the optimum resistance values of the resistors

101

to

104

vary in response to the deviation of the typical value Rtyp of the load resistance R.

Hence, the resistance values of the resistors

101

to

104

need to be adjusted for each electric load L. This forces the workers to perform very complicated adjustment which is not practically feasible.

The electric load L may be a pressure sensing element of a Wheatstone bridge circuit consisting of four strain gauges made of diffused resistors. The resistance value of each diffused resistor has a dispersion range of approximately ±10˜20% due to manufacturing error of the diffusion density of impurities or the width of resistor wire. It is therefore difficult to adopt the above-described conventional constant current supply circuit to this kind of pressure sensor.

SUMMARY OF THE INVENTION

In view of the foregoing problems, the present invention has an object to provide a constant current supply circuit capable of supplying constant current to an electric load irrespective of temperature, even when the resistance value of the electric load is not constant due to the manufacturing error.

To accomplish the above and other related objects, the present invention provides a constant current supply circuit for supplying constant current to an electric load. The constant current supply circuit comprises a primary transistor having a collector connected to one end of the electric load for controlling current supplied to the electric load. A current path resistor is connected between an emitter of the primary transistor and a reference voltage terminal for forming an electric path supplying the current to the electric load via the primary transistor. First, second, third and fourth resistors are serially connected in this order between one potential terminal of an electric power source and the other potential terminal of the electric power source. A secondary transistor, being identical in type with the primary transistor, has a collector connected to a connecting point of the second resistor and the third resistor, a base connected to a connecting point of the third resistor and the fourth resistor, and an emitter connected to the other potential terminal of the electric power source. The primary transistor has a base connected to a connecting point of the first resistor and the second resistor.

According to this arrangement, load current I is supplied via the primary transistor to the electric load. The load current I is substantially identical with current I′ flowing across the current path resistor. The type and the resistance value of respective first to fourth resistors can be optimized so that a voltage applied between both ends of the current path resistor is maintained at a constant value irrespective of temperature. Thus, it becomes possible to supply constant load current I to the electric load even when the resistance value of a manufactured electric load (i.e., actual load resistance) is different from a designated value.

Preferably, the first and second resistors are identical in type with the third and fourth resistors, and resistance values of the first, second, third and fourth resistors satisfy the following relationship:

0.5 < γ = \frac{R1}{R1 + R2} \cdot \frac{R3 + R4}{R4} < 1.5

where R

1

represents a resistance value of the first resistor, R

2

represents a resistance value of the second resistor, R

3

represents a resistance value of the third resistor, and R

4

represents a resistance value of the fourth resistor.

For example, to obtain preferable characteristics, the resistance values R

1

to R

4

of the first to fourth resistors are set to satisfy γ=1.

Preferably, the current path resistor is a thin-film resistor.

Preferably, a direct-current amplification factor of the primary transistor to the secondary transistor is equal to or larger than 50.

Preferably, each of the primary transistor and the secondary transistor is constituted by a pair of transistor elements connected in a Darlington pattern.

BRIEF DESCRIPTION OF THE DRAWINGS

The above and other objects, features and advantages of the present invention will become more apparent from the following detailed description which is to be read in conjunction with the accompanying drawings, in which:

FIG. 1

is a circuit diagram showing a constant current supply circuit in accordance with a first embodiment of the present invention;

FIG. 2

is a circuit diagram showing a constant current supply circuit in accordance with a second embodiment of the present invention;

FIG. 3

is a circuit diagram showing a constant current supply circuit in accordance with a third embodiment of the present invention;

FIG. 4

is a circuit diagram showing a constant current supply circuit in accordance with a fourth embodiment of the present invention; and

FIG. 5

is a circuit diagram showing a conventional constant current supply.

DESCRIPTION OF PREFERRED EMBODIMENTS

FIG. 1

is a circuit diagram showing a constant current supply circuit in accordance with a first embodiment of the present invention.

A current path resistor

15

has one end connected to a reference voltage terminal which is a ground terminal (=0V). An electric power source has a high-potential terminal (=VCC) and a low-potential terminal (=0V). Thus, the high-potential terminal of the electric power source supplies a positive voltage VCC. One end of first resistor

11

is connected to the high-potential terminal (=VCC) of the electric power source. One end of fourth resistor

14

and an emitter of secondary transistor Q

2

are connected to the low-potential terminal (i.e., ground terminal) of the electric power source.

A primary transistor Q

1

, which is an NPN type transistor, has a collector connected to an electric load L. The primary transistor Q

1

controls current supply to the electric load L. The other end of the electric load L is connected to a current supply terminal

16

which has a positive potential VD higher than the ground potential.

The current path resistor

15

, connected between an emitter of the primary transistor Q

1

and the ground terminal of the electric power source, has a resistance temperature coefficient substantially equal to 0. In other words, the current path resistor

15

has no temperature characteristics. The current path resistor

15

forms a current path for supplying current to the electric load L via the primary transistor Q

1

.

The current path resistor

15

is a thin-film resistor which is formed on a semiconductor integrated circuit. This kind of thin-film resistor is free from temperature characteristics. The resistance temperature coefficient of the thin-film resistor is substantially equal to 0.

The first resistor

11

, second resistor

12

, third resistor

13

, and fourth resistor

14

are sequentially connected in this order between the high-potential terminal and the low-potential terminal of the electric power source.

The secondary transistor Q

2

, being an NPN transistor identical in type with the primary transistor Q

1

, has a collector connected to a connecting point of the second resistor

12

and the third resistor

13

. A base of the secondary transistor Q

2

is connected to a connecting point of the third resistor

13

and the fourth resistor

14

. An emitter of the secondary transistor Q

2

is connected to the ground terminal.

R

1

, R

2

, R

3

, R

4

and R

5

represent resistance values of the first to fourth resistors

11

,

12

,

13

and

14

, and the current path resistor

15

. VBE

1

represents a base-emitter voltage of the primary transistor Q

1

. VBE

2

represents a base-emitter voltage of the secondary transistor Q

2

. A direct-current amplification factor (hfe) of the primary transistor Q

1

to the secondary transistor Q

2

is sufficiently large. Each base current of the transistors Q

1

and Q

2

is negligible.

The following formula 1 shows a relationship between an electric potential Vx of the connecting point of the first resistor

11

and the second resistor

12

and an electric potential Vy of the connecting point of the second resistor

12

and the third resistor

13

.

\begin{matrix} \begin{matrix} V x = V y + \frac{R2}{R1 + R2} (VCC - Vy) \\ = \frac{R3 + R4}{R4} VBE2 + \frac{R2}{R1 + R2} (VCC - \frac{R3 + R4}{R4} VBE2) \\ = \frac{R2}{R1 + R2} VCC + \frac{R1}{R1 + R2} \cdot \frac{R3 + R4}{R4} VBE2 \end{matrix} & (1) \end{matrix}

When the direct-current amplification factor (hfe) of the primary transistor Q

1

to the secondary transistor Q

2

is sufficiently large, the load current I flowing across the electric load L can be regarded as substantially equal to the current I′ flowing across the current path resistor

15

. The electric potential Vx of the connecting point of the first resistor

11

and the second resistor

12

is expressed by the following equation 2.

Vx=R

5

·

′+VBE

1

=R

5

·

I+

VBE

1

(2)

From the above equations (1) and (2), the load current I flowing across the electric load L is expressed by the following formula 3.

\begin{matrix} \begin{matrix} I = \frac{1}{R5} (\frac{R2}{R1 + R2} VCC + γ \cdot VBE2 - VBE1) \\ γ = \frac{R1}{R1 + R2} \cdot \frac{R3 + R4}{R4} \end{matrix}} & (3) \end{matrix}

In the above formula 3, the value in parentheses represents a voltage V applied between both ends of the current path resistor

15

.

The primary transistor Q

1

is identical in type (i.e., NPN-type transistor) with the secondary transistor Q

2

. Thus, the base-emitter voltage VBE

1

of the primary transistor Q

1

is substantially equal to the base-emitter voltage VBE

2

of the secondary transistor Q

2

irrespective of temperature. The first resistor

11

and the second resistor

12

are identical in type with the third resistor

13

and the fourth resistor

14

. And,, the resistance values of first, second, third and fourth resistors

11

˜

14

satisfy the following relationship:

γ = \frac{R1}{R1 + R2} \cdot \frac{R3 + R4}{R4} = 1

In other words., the resistance values of first, second, third and fourth resistors

11

˜

14

are set so as to cancel the VBE

1

and VBE

2

each varying largely in accordance with a temperature change.

The voltage V applied between both ends of the current path resistor

15

becomes V=VCC×R

2

/(R

1

+R

2

). The load current I is expressed by the following formula 4.

\begin{matrix} I = \frac{1}{R5} \cdot \frac{R2}{R1 + R2} VCC & (4) \end{matrix}

When the first resistor

11

and the second resistor

12

are identical in type with the third resistor

13

and the fourth resistor

14

, the mutual resistance temperature coefficient of the first resistor

11

and the second resistor

12

is identical with the mutual resistance temperature coefficient of the third resistor

13

and the fourth resistor

14

. Thus, the above equation γ=1 is established irrespective of temperature. The voltage V (=VCC×R

2

/(R

1

+R

2

)) applied between the both ends of the current path resistor

15

is maintained at a constant value irrespective of the temperature and the resistance value R of the electric load L.

The resistance temperature coefficient of the current path resistor

15

is substantially 0. As can be understood from the above-equation (4), the load current I flowing across the electric load L is maintained at a constant value irrespective of the temperature and the load resistance R.

In this manner, according to the constant current supply circuit of the present invention, the load current I supplied to the electric load L can be maintained at a constant value irrespective of temperature change even if the resistance value R of the electric load L varies. Thus, no adjustment of resistors is necessary for each electric load L.

In the above-described constant current supply circuit, it is preferable that the direct-current amplification factor of the primary transistor Q

1

to the secondary transistor Q

2

is large because the base current reduces to a negligible level. The collector current becomes substantially identical with the emitter current. The above formulas (1) to (4) are surely established. Practically, it is referable that the direct-current amplification factor of the primary transistor Q

1

to the secondary transistor Q

2

is equal to or larger than 50. It will be more preferable that the direct-current amplification factor is equal to or larger than 100. More specifically, when the direct-current amplification factor is in a level equivalent to 50, the error of the load current I due to influence of the base current becomes approximately 2%. This gives no bad influence to the accuracy. When the direct-current amplification factor is larger than 50, more preferable performance will be obtained.

From the foregoing, each of the primary transistor and the secondary transistor is constituted by a pair of transistor elements connected in a Darlington pattern, so that the direct-current amplification factor of the primary transistor Q

1

to the secondary transistor Q

2

becomes an extremely large value. Regarding the resistance values R

1

˜R

4

of the first to fourth resistors

11

to

14

, it is preferable that the following relationship is established.

\begin{matrix} 0.5 < γ = \frac{R1}{R1 + R2} \cdot \frac{R3 + R4}{R4} < 1.5 & (5) \end{matrix}

The reason for setting the above relationship 5 is as follows.

First, resistors for the integrated circuits are usually subjected to large manufacturing errors. The dispersion range of resistance value is approximately ±20% in case of diffused resistors or thin-film resistors. The dispersion range of the ratio of resistance values in a same chip is generally ±1˜2% depending on line width. Accordingly, the value of γ in the equations (3) and (5) is believed to be 1±several %.

An allowable error amount of γ can be estimated based on the calculation of the value in the parentheses of the above formula (3).

For example, the result will slightly vary depending on the setting of the temperature range (temperature variation) ΔT, the allowable variation of the load current I, and setting of VCC×R

2

/(R

1

+R

2

). It is now assumed that ΔT is 50° C., the allowable variation of the load current I is ±5%, and VCC×R

2

/(R

1

+R

2

) is 1.0V. In this condition, the temperature variation of the base-emitter voltage VBE of the bipolar transistor is approximately −2 mV/° C. From the formula (3), ±5%=(γ−1)×|−2×10

−3

|×50/1.0×100%

Thus, 0.5<γ<1.5

When the value of γ is within ±50% with respect to an ideal value

1

, the value in the parentheses of the formula (3) varies within a range of ±5% in response to the temperature.

The constant current supply circuit of the present invention is constituted by a semiconductor integrated circuit. In this case, the characteristics of the primary transistor Q

1

and the secondary transistor Q

2

can be equalized by configuring them in the same shape and disposing them in the same direction.

When the primary transistor Q

1

and the secondary transistor Q

2

have the same characteristics, the base-emitter voltage VBE

1

of the primary transistor Q

1

is surely equalized with the base-emitter voltage VIBE

2

of the secondary transistor Q

2

. Thus, the above formulas (1) to (5) can be established.

In the setting of resistance values R

1

˜R

5

of the resistors

11

˜

15

, it is desirable that the current flowing across the primary transistor Q

1

becomes equal to the current flowing across the secondary transistor Q

2

at the reference temperature Rtyp. In other words, it is easy to equalize the base-emitter voltage VBE

1

of the primary transistor Q

1

with the base-emitter voltage VBE

2

of the secondary transistor Q

2

.

Only the ratio of the resistance values R

1

˜R

4

of the first to fourth resistors

11

˜

14

gives influence to the constant current supply circuit of the present invention. The resistance temperature coefficient of each resistor needs not be 0. The first to fourth resistors

11

˜

14

can be made of diffused resistors or base resistors which have large temperature characteristics. It is preferable that the first resistor

11

and the second resistor

12

have the same resistance temperature coefficient while the third resistor

13

and the fourth resistor

14

have the same resistance temperature coefficient. The first to fourth resistors

11

˜

14

are identical in type. The first to fourth resistors

11

˜

14

can be made of thin-film resistors having no temperature characteristics.

According to the constant current supply circuit of the present invention, as understood from the formula (4), the load current I varies in proportion to the power source voltage VCC. Thus, the constant current supply circuit of the present invention has power source ratio characteristics.

In

FIG. 1

, the electric load L is a pressure sensing element of a Wheatstone bridge circuit consisting of four strain gauges made of diffused resistors.

The constant current supply circuit shown in

FIG. 1

is constituted by a semiconductor integrated circuit. The current path resistor

15

is a thin-film resistor made of Cr·Si having no temperature characteristics or other comparable metal. The first to fourth resistors

11

˜

14

are diffused resistors having substantially the same temperature characteristics.

The primary transistor Q

1

and the secondary transistor Q

2

are identical in type with each other and are disposed in the same direction, so that the transistors Q

1

and Q

2

have the same characteristics. The direct-current amplification factor of the primary transistor Q

1

to the secondary transistor Q

2

is equal to or larger than 100, so that the base current is negligible.

The first to fourth resistors

11

˜

14

have resistance values R

1

˜R

4

satisfying the relationship γ=1 at the reference temperature Ttyp.

According to the constant-current supply circuit in accordance with the first embodiment, the load current I supplied to the pressure sensing element can be maintained at a constant value irrespective of temperature even if the resistance value of respective gauge resistors constituting the pressure sensing element varies.

To supply a constant current to the pressure sensing element, it is desirable that the voltage VR applied to the pressure sensing element is large enough. The pressure sensing element can produce a large output voltage sufficient to accurately detect the pressure.

According to the constant current supply circuit of the first embodiment, the voltage VR applied to the pressure sensing element (i.e., electric load L) is defined by the following formula 6.

\begin{matrix} \begin{matrix} VR = VD - VCEQ1 - V \\ = VD - VCEQ1 - VCC \times R2 / (R1 + R2) \end{matrix} & (6) \end{matrix}

The following formula 7 defines a maximum value VRMAX of the above voltage VR.

\begin{matrix} \begin{matrix} VRMAX = VD - VCEQ1 (sat) - VCC \times R2 / (R1 + R2) \\ = VD - VCEQ1 (sat) - VCC \times R3 / (R3 + R4) \end{matrix} & (7) \end{matrix}

where VCEQ

1

(sat) represents a collector-emitter voltage of the primary transistor Q

1

in a saturated state.

From the foregoing, it is possible to adjust the maximum value VRMAX by adequately setting the resistance ratio of the first resistor

11

to the second resistor

12

and the resistance ratio of the third resistor

13

to the fourth resistor

14

. It is however necessary to consider the operating temperature range in determining the above each resistance ratio because VCEQ

1

(sat) and VRMAX vary in accordance with the temperature.

VR=load current I×load resistance R. The load current I is proportional to a reciprocal of the resistance value R

5

of the current path resistor

15

. It is therefore possible to change the value of VR by adjusting the resistance value R

5

of the current path resistor

15

. In this case, laser trimming the current path resistor

15

is preferable when the resistor

15

is made of a thin-film resistor.

FIG. 2

shows a constant current supply circuit in accordance with a second embodiment. The constant current supply circuit of the second embodiment differs from the constant current supply circuit of the first embodiment in that the primary transistor Q

1

consists of a pair of transistor elements Q

11

and Q

12

which are connected in a Darlington pattern. Similarly, the secondary transistor Q

2

consists of a pair of transistor elements Q

21

and Q

22

which are connected in a Darlington pattern.

According to the circuit arrangement of the second embodiment, the direct-current amplification factor of the primary transistor Q

1

to the secondary transistor Q

2

is extremely large. The substantial base current of respective transistors Q

1

and Q

2

can be reduced greatly. The above formulas 1 to 5 can be surely established. The accuracy of the constant current increases.

It is now assumed that, in the primary transistor Q

1

of Darlington type, VBE

11

represents a base-emitter voltage of the first-stage transistor Q

11

and VBE

12

represents a base-emitter voltage of the second-stage transistor Q

12

. In the secondary transistor Q

2

of Darlington type, VBE

21

represents a base-emitter voltage of the first-stage transistor Q

21

and VBE

22

represents a base-emitter voltage of the second-stage transistor Q

22

.

VBE

1

=

VBE

11

+

VBE

12

VBE

2

=

VBE

21

+

VBE

22

FIG. 3

shows a constant current supply circuit in accordance with a third embodiment. The constant current supply circuit of the third embodiment differs from the constant current supply circuit of the first embodiment in the following points (I) to (III).

(I) A low-potential terminal (having a ground potential) of the electric power source is connected to one end of the first resistor

11

. A high-potential terminal (having a power source voltage VCC) of the electric power source is connected to one end of the fourth resistor

14

and the emitter of the secondary transistor Q

2

. Namely, the constant current supply circuit of the third embodiment differs from the constant current supply circuit of the first embodiment in that the first to fourth resistors

11

to

14

are serially connected in the opposite order between the high-potential and low-potential terminals of the electric power source.

(II) Furthermore, one end of the current path resistor

15

is connected to the high-potential terminal of the electric power source. In other words, the reference potential of the current path resistor

15

is VCC. The other end of the current path resistor

15

is connected to the emitter of the primary transistor Q

1

. The base of the primary transistor Q

1

is connected to the connecting point of the first resistor R

1

and the second resistor R

2

. The collector of the primary transistor Q

1

is connected to one end of the electric load L. The other end of the electric load L is connected to the current supply terminal

16

having an electric potential VD which is lower than the power source potential VCC.

(III) The primary transistor Q

1

and the secondary transistor Q

2

are PNP transistors.

The constant current supply circuit of the third embodiment brings substantially the same function and effect as those of the constant current supply circuit of the first embodiment.

FIG. 4

shows a constant current supply circuit in accordance with a fourth embodiment. The constant current supply circuit of the fourth embodiment differs from the constant current supply circuit of the third embodiment in that the primary transistor Q

1

of PNP type consists of a pair of transistor elements Q

11

and Q

12

which are connected in a Darlington pattern. Similarly, the secondary transistor Q

2

of PNP type consists of a pair of transistor elements Q

21

and Q

22

which are connected in a Darlington pattern.

The constant current supply circuit of the fourth embodiment brings substantially the same function and effect as those of the constant current supply circuit of the third embodiment. Namely, the direct-current amplification factor of the primary transistor Q

1

to the secondary transistor Q

2

is extremely large. The substantial base current of respective transistors Q

1

and Q

2

can be reduced greatly. As a result, the accuracy of the constant current increases

The PNP transistor tends to have a small direct-current amplification factor compared with the NPN transistor. Thus, it is preferable to employ Darlington transistors as shown in FIG.

4

.

The present invention is not limited to the above-described embodiments, and therefore can be variously modified.

For example, it is preferable to provide a predetermined potential difference, which is not temperature dependent, between the emitter of the secondary transistor Q

2

and the other end of the current path resistor

15

.

The electric load is not limited to the pressure sensing element. And, therefore, the constant current supply circuit of the present invention can be applied to various resistive loads.

The present embodiments as described are therefore intended to be only illustrative and not restrictive, since the scope of the invention is defined by the appended claims rather than by the description preceding them. All changes that fall within the metes and bounds of the claims, or equivalents of such metes and bounds, are therefore intended to be embraced by the claims.

Number	Name	Date
3956661	Sakamoto et al.	May 1976
4591778	Holland	May 1986
4990846	Buck et al.	Feb 1991
5780921	Mitsuishi	Jul 1998

Constant current supply circuit

Information

Patent Number

Date Filed

Date Issued

Inventors

Original Assignees

Examiners

Agents

CPC

US Classifications

Field of Search

US

International Classifications

Abstract

Description

Claims

Priority Claims (1)

US Referenced Citations (4)

Foreign Referenced Citations (1)