Constant level-shift buffer amplifier circuits

Abstract

A push-pull dynamic amplifier is operable in reset and amplification phases. The amplifier includes first NMOS and PMOS input transistors that are electrically coupled to a first input terminal and a first output terminal. Second NMOS and PMOS input transistors are electrically coupled to a second input terminal and a second output terminal. First and second reset switches are electrically coupled to the first and second output terminals, respectively. A power supply switch is electrically coupled to the first and the second PMOS transistors, and a ground switch is electrically coupled to the first and the second NMOS transistors. During the reset phase, the reset switches are closed and the power supply switch and the ground switch are opened. During the amplification phase, the reset switches are opened and the power supply switch and the ground switch are closed.

Description

TECHNICAL FIELD

This application generally relates to buffer amplifier circuits.

BACKGROUND

Buffer amplifiers are widely used in electronic systems. They provide high input impedance, low output impedance, and voltage gain close to 1. For example, level-shifting buffer amplifiers (LSBAs) are employed in switched-capacitor circuits to bootstrap the virtual ground node, thereby improving the circuit performance. See, for example, U.S. Pat. No. 9,214,912, which is hereby incorporated by reference.

FIG. 1 shows a prior art buffer amplifier circuit 10 comprising a source follower. The gate of the NMOS transistor M1 is the input terminal of the buffer amplifier and the source of M1 is the output terminal. The current source I provides the bias current for M1, and the current l_o is the load current, the current delivered to the load. The current source I is typically implemented by a transistor current source, for example, an NMOS transistor. The gate-to-source voltage of M1 provides the level shift V_LS between the input and the output voltages:

$\begin{array}{cc}{V}_{\mathrm{LS}}=V_{\mathrm{GS}1}=V_T+\sqrt{\frac{2I_{D1}}{{\mu }_n{C_{\mathrm{OX}}\left(\frac{W}{L}\right)}_1}}=V_T+\sqrt{\frac{2\left(I+I_O\right)}{{\mu }_n{C_{\mathrm{OX}}\left(\frac{W}{L}\right)}_1}}& \left(1\right)\end{array}$ where I_D1 is the drain current through M₁, and

${\left(\frac{W}{L}\right)}_1$ is W over L ration of M1. V_T is the threshold voltage, μ_n the electron mobility, and C_OX, the oxide capacitance per unit area.

The incremental gain of the buffer amplifier is given by

$\begin{array}{cc}{a}_v=\frac{1}{1+\frac{1}{g_{m1}}\left(\frac{1}{r_{o1}}+\frac{1}{R_{\mathrm{out}}}\right)}& \left(2\right)\end{array}$ where g_m1 and r_o1 are the transconductance and the output resistance of M1, respectively, and R_out is the incremental output resistance of the current source I. The incremental gain in Equation (2) is typically slightly less than 1, and depends on the device geometry, bias condition, and R_out.

The output resistance R_o of the source follower buffer amplifier is approximately the inverse of the transconductance of M1:

$\begin{array}{cc}{R}_o\approx \frac{1}{g_{m1}}& \left(3\right)\end{array}$

The level shift given by Equation (1) varies considerably, due to process, temperature, and supply voltage (PVT) variations of V_T, μ_n, C_OX, and I. The variation of V_T alone can be as much as 250-350 mV across PVT. The variations of μ_n, C_OX, and I significantly increase the level shift variation. In addition, the load current I_O also affects the level shift as indicated in Equation (1).

The large amount of variation in the level shift in a source-follower buffer amplifier poses a difficult challenge in systems where a precise, constant level shift is required. For example, in virtual ground bootstrapping circuits in U.S. Pat. No. 9,214,912, the level shift determines the reference voltage of the system, thus needs to be kept constant. It is possible to keep the level shift V_GS1 constant by adjusting the current I as can be seen in Equation (1). However, given the large variation of V_T and variations of μ_n and C_OX, the current I needs to be adjusted by a large factor. As an example, if M1 is biased in weak inversion, more than three orders of magnitude current adjustment would be required just to compensate for V_T variation. This amount of current adjustment is highly undesirable, because the important parameters for the buffer amplifier such as the bandwidth and the output resistance will vary accordingly.

FIG. 2 illustrates another prior art LSBA 20, also called a flipped source follower (FSF). As in the source follower in FIG. 1, the level shift is given by V_GS1:

$\begin{array}{cc}{V}_{LS}=V_{GS1}=V_T+\sqrt{\frac{2I}{{\mu }_n{C_{\mathrm{OX}}\left(\frac{W}{L}\right)}_1}}& \left(4\right)\end{array}$

The transistor M2 provides negative feedback to keep the current through M1 constant at I independent of the load current I_O. Therefore, the variation of the level shift due to the load current is eliminated. Another advantage of the FSF compared with the standard source follower in FIG. 1 is that the negative feedback provided by M2 reduces the output resistance by a factor of g_m2r_o1, where g_m2 is the transconductance of M2, and r_o1 is the output resistance of M1. However, as in the LSBA in FIG. 1, the level shift is subject to the variations of V_T, μ_n and C_OX, thus is sensitive to PVT variations.

SUMMARY

Example embodiments described herein have innovative features, no single one of which is indispensable or solely responsible for their desirable attributes. The following description and drawings set forth certain illustrative implementations of the disclosure in detail, which are indicative of several exemplary ways in which the various principles of the disclosure may be carried out. The illustrative examples, however, are not exhaustive of the many possible embodiments of the disclosure. Without limiting the scope of the claims, some of the advantageous features will now be summarized. Other objects, advantages and novel features of the disclosure will be set forth in the following detailed description of the disclosure when considered in conjunction with the drawings, which are intended to illustrate, not limit, the invention.

One or more embodiments are directed to a level shifting buffer amplifier circuit having an input terminal and an output terminal, a first transistor, a current source, a variable resistance electrically coupled to the first transistor, wherein a resistance of the variable resistance is a function of a voltage at a control terminal, and wherein the buffer amplifier provides a constant level shift between the input and the output terminals.

The variable resistance may comprise a variable resistor in some embodiments. In other embodiments, the variable resistor may comprise a transistor, for example a PMOS transistor or an NMOS transistor. The transistor(s) may further comprise drain, source and gate terminals. In some aspects, said transistor(s) may have a variable resistance between their drain and source terminals. In other aspects, said transistor(s) may have a gate terminal acting as said control terminal.

One or more embodiments are directed to a level shifting buffer amplifier producing a level shift between an input and output terminals having a first transistor and a second transistor, a current source, and a variable resistance electrically coupled to the first transistor, wherein a resistance of the variable resistance is a first function of a voltage at a control terminal, wherein the level shift is a second function of the voltage at the control terminal.

BRIEF DESCRIPTION OF THE DRAWINGS

For a fuller understanding of the nature and advantages of the present concepts, reference is made to the following detailed description of preferred embodiments and in connection with the accompanying drawings. In the drawings, like reference characters generally refer to like features (e.g., functionally-similar and/or structurally-similar elements).

FIG. 1 is a schematic diagram of a source follower buffer amplifier circuit according to the prior art.

FIG. 2 is a schematic diagram of a flipped source follower buffer amplifier circuit according to the prior art.

FIG. 3 is a schematic diagram of a level shifting buffer amplifier circuit according to one or more embodiments.

FIG. 4 is a schematic diagram of a level shifting buffer amplifier circuit according to the first embodiment of the invention using an NMOS transistor as a variable resistor.

FIG. 5 is a schematic diagram of the first embodiment of a level shifting buffer amplifier circuit according to the first embodiment of the invention using a PMOS transistor as a variable resistor.

FIG. 6 is a schematic diagram of a level shifting buffer amplifier circuit according to a second embodiment of the invention.

FIG. 7 is a schematic diagram of a level shifting buffer amplifier circuit according to a third embodiment of the invention.

FIG. 8 is a schematic diagram of a level shifting buffer amplifier circuit according to a fourth embodiment of the invention.

FIG. 9 is a schematic diagram of a level shifting buffer amplifier circuit according to a fifth embodiment of the invention.

FIG. 10 is a schematic diagram of a level shifting buffer amplifier circuit according to a sixth embodiment of the invention.

FIG. 11 is a diagram of a control loop to keep the level shift constant in level shifting buffer amplifier circuits according to various embodiments of the invention.

FIG. 12 is an illustration of one embodiment of a combination of a difference amplifier and an integrator in a control loop shown in FIG. 11.

FIG. 13 illustrates an embodiment of an amplifier with a variable resistance implemented in a NMOS transistor.

FIG. 14 illustrates an embodiment of an amplifier with a variable resistance implemented in a PMOS transistor.

FIG. 15 illustrates an embodiment comprising a source follower with a variable threshold transistor.

FIG. 16 illustrates an embodiment comprising a FSF with a variable threshold transistor.

FIG. 17 illustrates an embodiment comprising a feedback control loop.

FIG. 18 illustrates a switching circuit portion comprising a differential switched-capacitor integrator.

DETAILED DESCRIPTION

The following discussion illustrates detailed descriptions of various concepts related to, and embodiments of, inventive apparatus relating to constant level-shift buffer amplifier circuits. It should be appreciated that various concepts introduced above and discussed in greater detail below may be implemented in any of numerous ways, as the disclosed concepts are not limited to any particular manner of implementation. Examples of specific implementations and applications are provided primarily for illustrative purposes.

As is evident from Equations (1) and (4), prior art source follower buffer amplifiers' level shift varies a great deal across PVT. The inventors have recognized that it is advantageous to provide a control for the level shift to make it constant (or less variable), for example, over PVT variation.

FIG. 3. shows an exemplary buffer amplifier 30 comprising an NMOS input transistor M1, a current source I, and a variable resistor R_var.

The variable resistor R_var provides an IR voltage drop such that the level shift is given by:

$\begin{array}{cc}{V}_{LS}=V_{GS1}+\left(I+I_O\right)R_{var}=V_T+\sqrt{\frac{2\left(I+I_O\right)}{{\mu }_n{C_{\mathrm{OX}}\left(\frac{W}{L}\right)}_1}}+\left(I+I_O\right)R_{\mathrm{var}}& \left(5\right)\end{array}$ where V_GS1 is the gate-to-source voltage of M1,

${\left(\frac{W}{L}\right)}_1$ is the width (W) over length (L) ratio of M1.

As can be seen in Equation (5), it will be possible to adjust the value of R_var to counter the variations of V_T and

$\sqrt{\frac{2\left(I+I_O\right)}{{\mu }_n{C_{\mathrm{OX}}\left(\frac{W}{L}\right)}_1}}$ without changing the bias current I.

In some applications, it may be advantageous to adjust both R_var and I for more flexibility in the adjustment.

FIG. 4 illustrates a level shifting buffer amplifier according to one or more embodiments wherein the variable resistor R_var of the previous example is substituted by an NMOS transistor MR1. We note that in this and other present embodiments and examples, the substitution or replacement of a transistor in place of a resistor, e.g., a variable resistance, is intended broadly as would be understood by those skilled in the art. Therefore, the description of a resistor or variable resistor to comprise such other components (e.g., NMOS or PMOS transistors) is intended to convey a general ability to insert or replace the resistance or variable resistance of one such element with the other as best suits a given implementation.

The NMOS transistor MR1 is biased in the triode region so that its ON resistance R_ON functions as R_var, which is controlled by the control voltage V_CONT applied at the gate of MR1.

$\begin{array}{cc}{R}_{var}=R_{ON}\approx \frac{1}{{\mu }_n{C_{\mathrm{OX}}\left(\frac{W}{L}\right)}_{R1}\left(V_{\mathrm{CONT}}-V_S-V_T\right)}& \left(6\right)\end{array}$ where

${\left(\frac{W}{L}\right)}_{R1}$ is the W over L ratio of MR1, V_CONT is the control voltage applied to the gate of MR1, and V_S the voltage at the source of MR1, which is the output voltage V_O.

The variable-resistance embodiments of FIGS. 3 and 4 can result in higher output resistances and lower voltage gains than the prior art LSBA of FIG. 1. The variable resistor R_var is in series with the resistance into the source of M1, increasing the output resistance to:

$\begin{array}{cc}{R}_o\approx \frac{1}{g_{m1}}+R_{var}& \left(7\right)\end{array}$ Since R_var is a function of the output voltage according to Equation (6), it can be shown that the incremental gain is further reduced from Equation (2) by the factor of

$1+\frac{R_{\mathrm{var}}I}{V_{\mathrm{GSR}1}-V_T}$ giving:

$\begin{array}{cc}{a}_v=\frac{1}{1+\frac{1}{g_{m1}}\left(\frac{1}{r_{o1}}+\frac{1}{R_{out}}\right)}·\frac{1}{1+\frac{R_{\mathrm{var}}I}{V_{\mathrm{GSR}1}-V_T}}& \left(8\right)\end{array}$ where V_GR1 is the gate-to-source voltage of MR1.

FIG. 5 illustrates a level shifting buffer amplifier according to one or more embodiments wherein the former variable resistor R_var is substituted by a PMOS transistor MR2. The PMOS transistor MR2 is biased in the triode region so that its ON resistance R_ON functions as R_var, which is controlled by the control voltage V_CONT:

$\begin{array}{cc}{R}_{var}=R_{ON}\approx \frac{1}{{\mu }_n{C_{\mathrm{OX}}\left(\frac{W}{L}\right)}_{R2}❘V_{\mathrm{CONT}}-V_S-V_T❘}& \left(9\right)\end{array}$ where

${\left(\frac{W}{L}\right)}_{R2}$ is the W over L ratio of MR2, V_CONT is the control voltage applied to the gate of MR2, and V_S the voltage at the source of MR2.

The output resistance of the embodiment of the present invention in FIG. 3 with the variable resistance in FIG. 5 is given by:

$\begin{array}{cc}{R}_o\approx \frac{1}{g_{m1}}+R_{var}& \left(10\right)\end{array}$ The incremental gain in this case, however, is increased by a factor of

$1+\frac{R_{\mathrm{var}}I}{❘V_{\mathrm{GSR}2}-V_T❘}:$

$\begin{array}{cc}{a}_v=\frac{1+\frac{R_{\mathrm{var}}I}{❘V_{\mathrm{GSR}2}-V_T❘}}{1+\frac{1}{g_{m1}}\left(\frac{1}{r_{o1}}+\frac{1}{R_{out}}\right)}& \left(11\right)\end{array}$ where V_GSR2 is the gate-to-source voltage of MR2.

Since this gain is closer to 1 than the LSBA circuit using the variable resistor in FIG. 4, the embodiment of the variable resistor in FIG. 5 may be a preferred embodiment for some applications.

FIG. 6 shows another embodiment of the present invention, based on the FSF in FIG. 2, further including a variable resistor R_var. The level shift is given by

$\begin{array}{cc}{V}_{LS}=V_{GS1}+{\mathrm{IR}}_{var}=V_T+\sqrt{\frac{2I}{{\mu }_n{C_{\mathrm{OX}}\left(\frac{W}{L}\right)}_1}}+{\mathrm{IR}}_{var}& \left(12\right)\end{array}$ where V_GS1 is the gate-to-source voltage of M1,

${\left(\frac{W}{L}\right)}_1$ is the W over L ratio of M1.

Again, the variable resistor R_var can be substituted or implemented by an NMOS transistor as in FIG. 7, or by a PMOS transistor as in FIG. 8, where the resistance R_var is given by Equation (6) and Equation (9), respectively.

In some aspects, the embodiments in FIGS. 6-8 compared with the embodiments in in FIGS. 3-5 eliminate or reduce the effect of the load current, and the output resistance is also reduced due to the negative feedback provided by M2. The output resistance is given by:

$\begin{array}{cc}{R}_o\approx \frac{\frac{1}{g_{m1}}+R_{var}}{g_{m2}{r}_{o1}}& \left(13\right)\end{array}$ where g_m1 and g_m2 are the transconductance of M1 and M2, respectively, and r_o1 is the output resistance of M1. The incremental gains of the LSBA in FIG. 7 is given by

$\begin{array}{cc}{a}_v=\frac{1}{1+\frac{1}{g_{m1}{r}_{o1}}}·\frac{1}{1+\frac{R_{\mathrm{var}}I}{V_{\mathrm{GSR}1}-V_T}}& \left(14\right)\end{array}$ where V_GSAR1 is the gate-to-source voltage of MR1.

The incremental gain of the LSBA in FIG. 8 is given by

$\begin{array}{cc}{a}_v=\frac{1+\frac{R_{\mathrm{var}}I}{V_{\mathrm{GSR}2}-V_T}}{1+\frac{1}{g_{m1}{r}_{o1}}}& \left(15\right)\end{array}$ where V_GSR2 is the gate-to-source voltage of MR2.

The incremental gain of the circuit in FIG. 8 is higher than that in FIG. 7, and may be preferred for some applications.

FIG. 9 illustrates another embodiment of the present invention. It comprises a parallel connection of a first source follower M1 and a second source follower M2 with a series variable resistor R_VAR. Preferably, M1 and M2 have two different threshold voltages such that V_T1>V_T2, where V_T1 is the threshold voltage of M1 and V_T2 is the threshold voltage of M2. In one example, M1 may be a standard V_T device and M2 may be a low V_T device. In another example, M1 may be a high V_T device and M2 may be a standard V_T device. In yet another example, M1 may be a high V_T device and M2 maybe a low V_T device. As R_var is varied, the amount of level shift is varied. When R_var is very small, the voltage drop IR_var across R_var is small. In this case, most of the current I+I_O flows through M2 because V_T2 is lower, and M1 is nearly OFF. In this case, the level shift is determined by the gate-to-source voltage V_GS2 of M2:

$\begin{array}{cc}{V}_{LS}=V_{GS2}+{\mathrm{IR}}_{var}\approx V_{GS2}=V_{T2}+\sqrt{\frac{2\left(I+I_O\right)}{{\mu }_n{C_{\mathrm{OX}}\left(\frac{W}{L}\right)}_2}}& \left(16\right)\end{array}$ where V_GS2 is the gate-to-source voltage of M2, and

${\left(\frac{W}{L}\right)}_2$ is tne W over L ratio of M2. The output resistance is determined by g_m2 of M2:

$\begin{array}{cc}{R}_o\approx \frac{1}{g_{m2}}+R_{var}\approx \frac{1}{g_{m2}}& \left(17\right)\end{array}$

On the other hand, if R_var is very large, most of the bias current I is steered to M1, and M2 is nearly OFF. In this case, the level shift is determined by the gate-to-source voltage V_GS1 of M1:

$\begin{array}{cc}{V}_{LS}\approx V_{GS1}=V_{T1}+\sqrt{\frac{2\left(I+I_O\right)}{{\mu }_n{C_{\mathrm{OX}}\left(\frac{W}{L}\right)}_1}}& \left(18\right)\end{array}$ where V_GS1 is the gate-to-source voltage of M1,

${\left(\frac{W}{L}\right)}_1$ is the W over L ratio of M1. The output resistance is determined by g_m1 of M2:

$\begin{array}{cc}{R}_o\approx \frac{1}{g_{m1}}& \left(19\right)\end{array}$ By varying R_var in between these two extremes, the level shift can be varied continuously between the values given in Equations (14) and (16). Therefore, the range of level shift is given by

$\begin{array}{cc}{V}_{T2}+\sqrt{\frac{2\left(I+I_O\right)}{{\mu }_n{C_{\mathrm{OX}}\left(\frac{W}{L}\right)}_2}}\le V_{\mathrm{LS}}\le V_{T1}+\sqrt{\frac{2\left(I+I_O\right)}{{\mu }_n{C_{\mathrm{OX}}\left(\frac{W}{L}\right)}_1}}& \left(20\right)\end{array}$ Again, the variable resistor R_var can be implemented by an NMOS transistor as shown in FIG. 10 or by a PMOS transistor as shown in and FIG. 11.

The adjustability, which is the difference between the upper and lower bounds of the level shift is given by

$\begin{array}{cc}\Delta V_{\mathrm{LS}}\approx V_{T1}-V_{T2}+\sqrt{\frac{2\left(I+I_O\right)}{{\mu }_n{C_{\mathrm{OX}}\left(\frac{W}{L}\right)}_1}}-\sqrt{\frac{2\left(I+I_O\right)}{{\mu }_n{C_{\mathrm{OX}}\left(\frac{W}{L}\right)}_2}}& \left(21\right)\end{array}$ If sizes of M1 and M2 are equal, the adjustability reduces to the difference between the threshold voltages:

$\begin{array}{cc}\Delta V_{\mathrm{LS}}\approx V_{T1}-V_{T2}& \left(22\right)\end{array}$ However, making

${\left(\frac{W}{L}\right)}_2>{\left(\frac{W}{L}\right)}_1$ gives wider adjustability. In addition, if transistors with different threshold voltages are not available, V_T1 and V_T2 are equal, unequal sizing between M1 and M2 gives the adjustability of

$\begin{array}{cc}\Delta V_{\mathrm{LS}}\approx \sqrt{\frac{2\left(I+I_O\right)}{{\mu }_n{C}_{\mathrm{OX}}\left\{{\left(\frac{W}{L}\right)}_1+{\left(\frac{W}{L}\right)}_2\right\}}}-\sqrt{\frac{2\left(I+I_O\right)}{{\mu }_n{C_{\mathrm{OX}}\left(\frac{W}{L}\right)}_2}}& \left(23\right)\end{array}$

FIG. 12 illustrates another embodiment of the present invention. It comprises a FSF with a parallel connection of a first source follower M1 and a second source follower M2 with a series variable resistor R_VAR. Preferably, M1 and M2 have two different threshold voltages such that V_T1>V_T2, where V_T1 is the threshold voltage of M1 and V_T2 is the threshold voltage of M2. In one example, M1 may be a standard V_T device and M2 may be a low V_T device. In another example, M1 may be a high V_T device and M2 may be a standard V_T device. In yet another example, M1 may be a high V_T device and M2 maybe a low V_T device. As R_var is varied, the amount of level shift is varied. When R_var is very small, the voltage drop IR_var across R_var is small. In this case, most of the current I flows through M2, and M1 is nearly OFF. In this case, the level shift is determined by the gate-to-source voltage V_GS2 of M2:

$\begin{array}{cc}{V}_{\mathrm{LS}}=V_{\mathrm{GS}2}+{\mathrm{IR}}_{\mathrm{var}}\approx V_{\mathrm{GS}2}=V_{T2}+\sqrt{\frac{2I}{{\mu }_n{C_{OX}\left(\frac{W}{L}\right)}_2}}& \left(24\right)\end{array}$ where V_GS2 is the gate-to-source voltage of M2,

${\left(\frac{W}{L}\right)}_2$ is the W over L ratio of M2. The output resistance is significantly reduced by the factor of g_m2r_o1 by the negative feedback provided by M3 as in the FSF, and is given by:

$\begin{array}{cc}{R}_o\approx \frac{\frac{1}{g_{m2}}+R_{var}}{g_{m3}{r}_{o1}}\approx \frac{1}{g_{m3}{r}_{o1}{g}_{m2}}& \left(25\right)\end{array}$

On the other hand, if R_var is very large, most of the bias current I is steered to M1, and M2 is nearly OFF. In this case, the level shift is determined by the gate-to-source voltage V_GS1 of M1:

$\begin{array}{cc}{V}_{\mathrm{LS}}\approx V_{\mathrm{GS}1}=V_{T1}+\sqrt{\frac{2I}{{\mu }_n{C_{\mathrm{OX}}\left(\frac{W}{L}\right)}_1}}& \left(26\right)\end{array}$ where V_GS1 is the gate-to-source voltage of M1,

${\left(\frac{W}{L}\right)}_1$ is the W over L ratio of M1. The output resistance is given by:

$\begin{array}{cc}{R}_o\approx \frac{1}{g_{m3}{r}_{o1}{g}_{m1}}& \left(27\right)\end{array}$

By varying R_var in between these two extremes, the level shift can be varied continuously between the values given in Equations (14) and (16). Therefore, the range of level shift is given by

$\begin{array}{cc}{V}_{T2}+\sqrt{\frac{2I}{{\mu }_n{C_{\mathrm{OX}}\left(\frac{W}{L}\right)}_2}}\le V_{\mathrm{LS}}\le V_{T1}+\sqrt{\frac{2I}{{\mu }_n{C_{\mathrm{OX}}\left(\frac{W}{L}\right)}_1}}& \left(28\right)\end{array}$ Again, the variable resistor R_var can be implemented by an NMOS transistor or a PMOS transistor, as shown in FIG. 13 and FIG. 14, respectively.

The adjustability, which is the difference between the upper and lower bounds of the level shift is given by

$\begin{array}{cc}\Delta V_{\mathrm{LS}}\approx V_{T1}-V_{T2}+\sqrt{\frac{2I}{{\mu }_n{C_{\mathrm{OX}}\left(\frac{W}{L}\right)}_1}}-\sqrt{\frac{2I}{{\mu }_n{C_{\mathrm{OX}}\left(\frac{W}{L}\right)}_2}}& \left(29\right)\end{array}$

If sizes of M1 and M2 are equal, the adjustability reduces to the difference between the threshold voltages:ΔV_LS=V_T1−V_T2  (30)However, making

${\left(\frac{W}{L}\right)}_2>{\left(\frac{W}{L}\right)}_1$ gives wider adjustability. In addition, if transistors with different threshold voltages are not available, V_T1 and V_T2 are equal, unequal sizing between M1 and M2 gives the adjustability of

$\begin{array}{cc}\Delta V_{\mathrm{LS}}\approx \sqrt{\frac{2I}{{\mu }_n{C}_{\mathrm{OX}}\left\{{\left(\frac{W}{L}\right)}_1+{\left(\frac{W}{L}\right)}_2\right\}}}-\sqrt{\frac{2I}{{\mu }_n{C_{\mathrm{OX}}\left(\frac{W}{L}\right)}_2}}& \left(31\right)\end{array}$

FIG. 15 illustrates another embodiment of the present invention. It comprises a source follower with a variable threshold transistor M1. The threshold voltage of M1 is controlled by changing the control voltage V_cont. In one example, the variable threshold transistor M1 is implemented in fully-depleted silicon-on-insulator (FDSOI) technology. The control voltage V_cont is applied to the back-gate voltage of the transistor. The FDSOI technology is superior to bulk CMOS technology for this embodiment because the threshold voltage can be varied by a large amount, by as much as a few hundred millivolts.

FIG. 16 illustrates another embodiment of the present invention. It comprises a FSF with a variable threshold transistor M1. The threshold voltage of M1 is controlled by changing the control voltage V_cont. In one example, the variable threshold transistor M1 is implemented in fully-depleted silicon-on-insulator (FDSOI) technology. The control voltage V_cont is applied to the back-gate voltage of the transistor.

FIG. 17 illustrates a feedback control loop to keep the level shift constant across PVT variations. A difference circuit 500 produces a difference between the level shift V_LS and a reference voltage V_REF. Th reference voltage V_REF is preferably independent of PVT variations. The reference voltage V_REF can be produced, for example, by a bandgap reference source. The difference V_LS−V_REF is integrated by an integrator 502. The output of the integrator drives the control voltage V_cont, in the various embodiments of the present invention. In one example, the difference circuit 500 and the integrator 502 are implemented by a differential switched-capacitor integrator shown in FIG. 18. Its operation is controlled by two non-overlapping clock phases, φ₁ and φ₂. When φ₁ is ‘high’ switches S1 and S2 are ‘ON’, and S3 and S4 are ‘OFF’. The input voltage and the output voltage of the buffer, the difference of which is equal to V_LS, are sampled across the sampling capacitor C_S during this phase. When φ₂ is ‘high’ switches S3 and S4 are ‘ON’, and S1 and S2 are ‘OFF’, applying the reference voltage V_REF on the lower plate of C_S. It can be shown that the resulting change of the integrator output voltage ΔV_o is given byΔV_o=V_IN−V_OUT=V_LS−V_REF  (32)If V_LS is larger than V_REF, the integrator output voltage keeps increasing after each clock cycle by ΔV_o, because ΔV_o is positive. This increases the control voltage V_cont. In the embodiments where the variable resistor is implemented by an NMOS transistor, this reduces R_var, and thus V_LS. Therefore, this negative feedback reduces V_LS such that V_LS=V_REF. In other embodiments, the control voltage needs to be reduced if V_LS is larger than V_REF. In these embodiments, a polarity inversion of the integrator output voltage is necessary. This can be accomplished for example, by using an inverting amplifier coupled to the output of the integrator, or by using a fully-differential integrator.

While various inventive embodiments have been described and illustrated herein, those of ordinary skill in the art will readily envision a variety of other means and/or structures for performing the function and/or obtaining the results and/or one or more of the advantages described herein, and each of such variations and/or modifications is deemed to be within the scope of the inventive embodiments described herein. More generally, those skilled in the art will readily appreciate that all parameters, dimensions, materials, and configurations described herein are meant to be exemplary and that the actual parameters, dimensions, materials, and/or configurations will depend upon the specific application or applications for which the inventive teachings is/are used. Those skilled in the art will recognize, or be able to ascertain using no more than routine experimentation, many equivalents to the specific inventive embodiments described herein. As a specific example, it may be desired to use PMOS input transistors in the amplifier circuits in FIGS. 3-16 instead of the NMOS input transistors as shown in the exemplary figures. Such “flipped” configurations will be appreciated by those who are skilled in the art. It is, therefore, to be understood that the foregoing embodiments are presented by way of example only and that, within the scope of the appended claims and equivalents thereto, inventive embodiments may be practiced otherwise than as specifically described. Inventive embodiments of the present disclosure are directed to each individual feature, system, article, material, kit, and/or method described herein. In addition, any sensible combination of two or more such features, systems, articles, materials, kits, and/or methods, if such features, systems, articles, materials, kits, and/or methods are not mutually inconsistent, is included within the inventive scope of the present disclosure.

Also, the invention described herein may be embodied as a method. The acts performed as part of the method may be ordered in any suitable way. Accordingly, embodiments may be constructed in which acts are performed in an order different than illustrated, which may include performing some acts simultaneously, even though shown as sequential acts in illustrative embodiments.

The invention should not be considered limited to the particular embodiments described above, but rather should be understood to cover all aspects of the invention as fairly set out in the attached claims. Various modifications, equivalent processes, as well as numerous structures to which the invention may be applicable, will be apparent to those skilled in the art to which the invention is directed upon review of this disclosure. The claims are intended to cover such modifications and equivalents.

Claims

1. A level-shifting buffer amplifier producing a level shift between input and output terminals, comprising: a current source;a first transistor electrically coupled to the input terminal and the output terminal;a second transistor electrically coupled to the input terminal and the current source; anda variable resistance electrically coupled to the first transistor and the output terminal, wherein a resistance of the variable resistance is a first function of a voltage at a control terminal,wherein the level shift is a second function of the voltage at the control terminal.

2. The buffer amplifier of claim 1, wherein a threshold voltage of the first transistor is lower than a threshold voltage of the second transistor.

3. The buffer amplifier of claim 2, further comprising a third transistor electrically coupled to the first and the second transistors, providing a negative feedback.

4. The buffer amplifier of claim 3, wherein the first transistor is an NMOS transistor.

5. The buffer amplifier of claim 3, wherein the first transistor is a PMOS transistor.

6. A level-shifting buffer amplifier, comprising: an input terminal and an output terminal a first transistor having a back-gate terminal; anda current source;a second transistor electrically coupled to the first transistor and the current source, providing a negative feedback,wherein: the first transistor is a fully-depleted silicon-on-insulator transistor, anda control voltage is applied to the back-gate terminal to provide a constant level shift between the input and the output terminals.

7. The buffer amplifier of claim 6, wherein the first transistor is an NMOS transistor.

8. The buffer amplifier of claim 6, wherein the first transistor is a PMOS transistor.