Patent Application
20100214210
The present invention relates to a current balancing device for balancing currents flowing through multiple loads connected in parallel, an LED lighting apparatus, an LCD backlight module, and an LCD display unit.
Conventional LED lighting devices in which multiple light emitting diodes (LEDs) are connected in series have been known from, for example, Japanese Patent Application Publications Nos. 2004-319583 (Patent Literature 1) and 2006-12659 (Patent Literature 2).
In the LED lighting device disclosed in Patent Literature 1, multiple LED units each including multiple LEDs connected in series are connected in parallel. If the multiple LED units each including the multiple LEDs connected in series are driven while being connected in parallel, the LED units have different voltage drops (forward voltage Vf of each LED), resulting in unbalanced currents flowing through the LED units connected in parallel. Accordingly, in Patent Literature 1, the currents flowing through the LED units are balanced by applying constant currents to the respective LED units by using a constant current circuit.
In an electric-discharge lamp lighting circuit disclosed in Patent Literature 2, currents flowing through multiple cold cathode fluorescent lamps (CCFLs) connected in parallel are balanced using transformers. Since the CCFLs are driven by an alternating current, sinusoidal currents flow through the balancing transformers. The currents are balanced by connecting the CCFLs and respective balancing transformers in series and by forming a closed circuit with secondary windings of the balancing transformers.
However, in Patent Literature 1, if the constant current circuit is connected, the differences in voltage drop of the LED units result in losses.
In Patent Literature 2, currents are balanced using the balancing transformers, and thus there is no loss due to the different voltages of the CCFLs. However, in the case of LEDs through which only direct currents flow, the direct currents cannot be balanced using transformers. In other words, the higher the frequency is, the smaller the balancing transformers can be made, but the lower the frequency is, the larger the balancing transformers are. In addition, the transformers are saturated with the direct current and cannot be used as the balancing transformers in the LED circuits.
An object of the present invention is to provide a current balancing device, an LED lighting apparatus, an LCD backlight module, and an LCD display unit, in which a loss in a circuit balancing currents flowing through multiple loads having different impedances is reduced to achieve a high efficiency.
A current balancing device according to an aspect of the present invention includes a power supply unit configured to output an alternating current; and a plurality of series circuits each connected to an output of the power supply unit, each series circuit including at least one winding, at least one rectifying element, and at least one load, which are connected in series. In the current balancing device, currents flowing respectively through the plurality of series circuits are balanced based on an electromagnetic force generated at the at least one winding.
An LED lighting apparatus according to an aspect of the present invention includes the current balancing device, and the one load is an LED load.
An LCD backlight module according to an aspect of the present invention includes the current balancing device, and the load is an LED load causing an LCD cell to emit light.
An LCD display according to an aspect of the present invention includes the current balancing device, and the load is an LED load causing an LCD cell to emit light.
Hereinafter, power supply apparatuses provided with current balancing devices according to embodiments of the present invention are described in detail with reference to the drawings.
At first, a transformer can balance an alternating current but cannot balance direct currents in a direct current-driving circuit such as an LED. The present invention includes multiple series circuits each of which is connected to an output of a power supply unit outputting an alternating current and includes at least one winding, at least one rectifying element, and at least one load which are connected in series and is characterized by balancing currents flowing through the multiple series circuits based on electromagnetic force generated at the at least one winding.
Each embodiment described below shows an example of the current balancing device where the loads having different impedances are LEDs.
In Embodiment 1 shown in
An end of the secondary winding Ns of the transformer T is connected to an end of a winding N1, and the other end of the winding N1 is connected to an anode of a diode D1 which half-wave rectifies the alternating current. Between a cathode of the diode D1 and the other end of the secondary winding Ns, a load LD1 (LEDs la to 1e) is connected. In Embodiment 1, a first series circuit is composed of the winding N1, diode D1, and load LD1.
The end of the secondary winding Ns of the transformer T is also connected to an end of a winding S1, and the other end of the winding S1 is connected to an anode of a diode D2 which half-wave rectifies the alternating current. Between a cathode of the diode D2 and the other end of the secondary winding Ns, a load LD2 (LEDs 2a to 2e) is connected. In Embodiment 1, a second series circuit is composed of the winding S1, diode D2, and load LD2. The windings N1 and S1 are electromagnetically coupled to each other, constituting a transformer T1. The impedance of the load LD1 is different from the impedance of the load LD2 in Embodiment 1.
First, at time t0, the switching element Q1 is on. The beginning of the winding Np of the transformer T has a negative potential, and the beginning of the winding Ns also has a negative potential. During a period ST1 starting from the time t0, because of the diodes D1 and D2 included in the series circuits, the alternating current supplied from the winding Ns does not flow through the first and second series circuits connected to the winding Ns. There is no current flowing through the transformer T and first and second series circuits. Accordingly, a magnetizing current of the transformer T flows through a path of Vin→Np→Q1→Vin.
When the switching element Q1 is turned off at time t1, the magnetizing current stored in the transformer T during the period ST1 generates counter-electromotive force with a positive potential at the beginning of the winding Np, and thus the beginning of the winding Ns also has a positive potential. Accordingly, during the period ST2 starting from the time t1, the diodes connected to the series circuits conduct the current. The current flows through the path of Ns→N1→D1→load LD1→Ns and the path of Ns→S1→D2→load LD2→Ns. In such a manner, the currents I(D1) and I(D2) whose magnitude change with time, that is, which have alternating components, flow through the individual series circuits.
The currents I(D1) and I(D2) flow through the windings N1 and S1, respectively, thus generating magnetic flux according to the currents. Since the windings N1 and S1 constitute the transformer T1, at this time, the magnetic fluxes generated at the individual windings interact with each other so that the magnitudes of the magnetic fluxes are equalized. Accordingly, even when the magnitudes of the currents I(D1) and I(D2) are originally different from each other, the currents I(D1) and I(D2) are balanced (equalized) in magnitude to a certain value and supplied to the loads LD1 and LD2. In such a manner, the loads LD1 and LD2 have different impedances, but the magnitudes of the currents I(D1) and I(D2) of the first and second series circuits are equal to each other.
In Embodiment 1, the currents are balanced in magnitude by the electromagnetic forces generated at the windings. Accordingly, there occurs a loss mainly due to the winding resistances. This loss is smaller than the loss in the constant current circuit of Patent Literature 1, and the loss in the balancing circuit can be thus reduced.
Embodiment 1 illustrates a lighting device including the loads LD1 and LD2 each including multiple LEDs connected in series. Accordingly, supplying the balanced currents to the loads LD1 and LD2 allows the multiple LEDs to uniformly emit light, thus illuminating a liquid crystal display (LCD) uniformly, for example.
Embodiments 2 to 5 shown in
The winding N1 (N2, N3, and N4) and the winding S1 (S2, S3, and S4) are magnetically coupled so that a current half-wave rectified by the diode D1 (D2, D3, and D4) is balanced, thus constituting the transformer T1 (T2, T3, and T4).
In other words, each series circuit includes two windings connected in series, and the two windings are electromagnetically coupled to each other in such a manner as to serve as the primary and secondary windings of the transformer, respectively.
In the connection of Embodiment 2, currents flowing through the winding N1 (N2, N3, and N4) and the winding S1 (S2, S3, and S4) of the transformer T1 (T2, T3, and T4) are equal to each other because of the characteristics thereof. The current supplied from the power supply unit 10 can be equally distributed to the loads LD1 to LD4. Accordingly, Embodiment 2 can provide a similar effect to that of the current balancing device according to Embodiment 1. Moreover, two windings are connected in each series circuit. Accordingly, the transformers can be reduced in size as a balancing transformer and the same transformer can be used for the two windings.
The windings S1 to S4 are connected to one another in a closed loop, and the winding N1 (N2, N3, and N4) and the winding S1 (S2, S3, and S4) are electromagnetically coupled to each other, constituting a transformer T1 (T2, T3, and T4). Specifically, the series circuits respectively include first windings, and second windings are electromagnetically coupled to the first windings, respectively. The second windings are connected to each other in series to form a closed loop. An equal current flows through the windings S1 to S4.
A current half-wave rectified by the diode D1 (D2, D3, and D4) flows through the winding N1 (N2, N3, and N4). The winding N1 (N2, N3, and N4) and the winding S1 (S2, S3, and S4) are magnetically coupled to each other so that the current flowing through the winding N1 (N2, N3, and N4) is balanced with the current flowing through the winding S1 (S2, S3, and S4), constituting the transformer T1 (T2, T3, and T4). Accordingly, in the connection of Embodiment 3, the currents flowing through the winding N1 (N2, N3, and N4) and the winding S1 (S2, S3, and S4) of the transformer T1 (T2, T3, and T4) are equal to each other because of the characteristics thereof. The current supplied from the power supply unit 10 can be equally distributed to the loads LD1 to LD4. Accordingly, the current balancing device according to Embodiment 3 can provide a similar effect to that of the current balancing device according to Embodiment 1. Moreover, the same transformer to serve as a balancing transformer can be used for the current.
The winding N1 (N2 and N3) and the winding S1 (S2 and S3) are magnetically coupled to each other so that currents to be half-wave rectified by the diodes can be balanced, constituting a transformer T1 (T2 and T3). Specifically, the current balancing device includes series circuits including a single winding and series circuits each including two windings electromagnetically coupled to each other as the primary and secondary windings of the transformers.
In the connection of Embodiment 4, currents flowing through the winding N1 (N2 and N3) and the winding S1 (S2 and S3) of the transformer T1 (T2 and T3) are equal to each other because of the characteristics thereof. The current supplied from the power supply unit 10 can be equally distributed to the loads LD1 to LD4. Accordingly, the current balancing device according to Embodiment 4 can provide a similar effect to that of the current balancing device according to Embodiment 1. Moreover, the transformer T4 composed of the windings N4 and S4, which is included in Embodiments 2 and 3, can be eliminated in Embodiment 4, and thus the current balancing device thereof can be configured at low cost.
The winding N1 (N2 and N3) and the winding S1 (S2 and S3) are magnetically coupled to each other so that currents to be half-wave rectified by the diodes can be balanced to constitute constituting a transformer T1 (T2 and T3). In the connection of Embodiment 5, currents flowing through the winding N1 (N2 and N3) and the winding S1 (S2 and S3) of the transformer T1 (T2 and T3) are equal to each other because of the characteristics thereof. The current supplied from the power supply unit 10 can be equally distributed to the loads LD1 to LD4. Accordingly, the current balancing device according to Embodiment 5 can provide a similar effect to that of the current balancing device according to Embodiment 1. Moreover, in Embodiment 5, the transformer T4 composed of the windings N4 and S4, which is included in Embodiments 2 and 3, can be eliminated, and thus the current balancing device of Embodiment 5 can be configured at low cost.
In Embodiment 6 shown in
The switching elements QL and QH are alternately turned on and off to supply the sinusoidal current from the winding Ns of the transformer T, the sinusoidal current generated by resonance of the leakage inductances Lr1 and Lr2 and the current resonant capacitor Cri.
First, at time t0, when the switching element QH is turned on while the switching element QL is off, the beginning of the winding Np of the transformer T has a negative voltage, and the beginning of the winding Ns also has a negative voltage. Accordingly, in a period ST1 starting from the time t0, the alternating current supplied from the winding Ns does not flow because of the diodes D1 and D2 included in the first and second series circuits connected to the winding Ns. Accordingly, no current flows through the first and second series circuits. The current I(QH) flowing through the switching element QH starts from the minus side to flow through a path of Vin (positive electrode)→QH (DH)→Lr1→Lp→Cri→Vin (negative electrode). Because of the resonance of the current resonant capacitor Cri, the magnetizing inductance Lp, and the leakage inductance Lr1, the magnitude of the current I(QH) increases with time. At this time, the current resonant capacitor Cri is charged.
Next, at time t1, when the switching element QH is turned off and the switching element QL is turned on, the current having flown to the magnetizing inductance Lp starts to flow through the path of Lp→Cri→DL(QL)→Lr1→Lp. The beginning of the winding Np has a positive voltage, and the beginning of the winding Ns also has a positive voltage.
Accordingly, in a period ST2 starting from the time t1, the diodes D1 and D2 connected to the first and second series circuits begin to conduct the current, and the current flows through the path of Ns→N1→D1→load LD1→Ns, which passes the winding N1, and the path of Ns→S1→D2→load LD2→Ns.
This current is supplied from the current resonant capacitance Cri through the transformer T in the path of Cri→Np→Lr2→Lr1→QL(DL)→Cri. This causes resonance of the current resonant capacitor Cri and the leakage inductances Lr1 and Lr2, thus supplying the sinusoidal half-wave current. In such a manner, the currents I(D1) and I(D2), whose magnitudes change with time, that is, which have alternating components, flow through the respective series circuits. The current balancing device according to Embodiment 6 can provide a similar effect to that of the current balancing device according to Embodiment 1. Furthermore, the sinusoidal current flows through the current balancing circuit. Accordingly, there is less noise generated in the current balancing device according to Embodiment 6 than in the current balancing device according to Embodiment 1.
The power supply unit 10a according to Embodiment 6 can be connected to the multiple series circuits shown in Embodiments 2 to 5.
In Embodiment 7 shown in
Both ends of the primary winding Np of the transformer Tare connected to a series circuit including a current resonant capacitor Cri and a switching element QH. Both ends of the switching element QH are connected to a diode DH. The transformer T includes leakage inductances Lr1 and Lr2. Reference letter Lp denotes a magnetizing inductance of the transformer T. The diodes DL and DH may be parasitic diodes Di of the switching elements QL and QH.
The power supply unit 10b according to Embodiment 7 is obtained by changing the configuration of the power supply unit 10a of Embodiment 6. The DC power supply Vin and the current resonant capacitor Cri are replaced with each other. The operation of Embodiment 7 provides the operation waveforms approximately the same as those of the operation of Embodiment 6, and the alternating current supplied from the power supply unit 10b according to Embodiment 7 is sinusoidal. The current balancing device according to Embodiment 7 therefore can provide a similar effect to that of the current balancing device according to Embodiment 6.
Note that the power supply unit 10b according to Embodiment 7 can be connected to the multiple series circuits shown in Embodiments 2 to 5.
In Embodiment 8 shown in
Note that the power supply unit 10 according to Embodiment 8 can be replaced with the power supply units 10a and 10b according to Embodiments 6 and 7. Moreover, the smoothing capacitors C1 and C2 according to Embodiments 8 can be applied to the multiple series circuits shown in Embodiments 2 to 5.
Note that the power supply unit 10a according to Embodiment 9 can be replaced with the power supply unit 10b according to Embodiment 7.
In Embodiment 10 shown in
First, at time t0, when the switching element QL is turned on while the switching element QH is off, the beginning of the winding Np then has a negative voltage, and the beginning of the winding Ns also has a negative voltage. Accordingly, during a period ST1 starting from the time t0, a reverse voltage is applied to the diodes D1 and D2, and there is no current flowing through the first and second series circuits.
However, a forward voltage is applied to the diode D10, and the current flows from the winding Ns in the path of Ns→C10→D10→Ns. This current is supplied from the winding Np through the transformer T, and the current I(QL) starts from the minus side to flow in the path of Cri→Np→QL(DL)→Cri and becomes a sinusoidal half-wave current because of the resonance of the current resonant capacitor Cri and the inductances Lr1 and Lr2. The magnitude of the current I(QL) increases with time to reach zero at time t1.
Next, at time t2, when the switching element QL is turned off and the switching element QH is turned on, the current having flown to the inductance Lp starts to flow in the path of Lp→Lr1→QH(DH)→Vin→Cri→Lp. The beginning of the winding Np of the transformer T then has a positive voltage, and the beginning of the winding Ns also has a positive voltage. Accordingly, in a period ST3 starting from the time t2, the diodes D1 and D2 connected to the series circuits conduct the current, and the current flows in the path of Ns→N1→D1→load LD1→Ns, which passes the winding N1, and in the path of Ns→S1→D2→load LD2→Ns.
This current then flows in the path of Vin→QH(DH)→Lr1→Lr2→Np→Cri→Vin and is supplied from Vin through the transformer T. The resonance of the current resonant capacitor Cri and leakage inductances Lr1 and Lr2 then supplies the sinusoidal half-wave current.
In such a manner, the currents I(D1) and I(D2), whose magnitudes change with time, that is, which have alternating components, flow through the respective series circuits. The current balancing device according to Embodiment 10 can therefore provide a similar effect to that of the current balancing device according to Embodiment 1. Moreover, in this embodiment, the full wave of the output current of the transformer T is used, thus increasing the utilization of the transformer T. The transformer T can be therefore miniaturized, and thus the current balancing device can be configured at low cost.
Note that the power supply unit 10a according to Embodiment 10 can be replaced with the power supply unit 10 or 10b according to Embodiments 1 or 7. The capacitors C10 and diode D10 according to Embodiment 10 can be applied to multiple series circuits shown in Embodiments 2 to 5.
The current balancing device of Embodiment 11 shown in
The power supply unit 10a according to Embodiment 11 can be replaced with one of the power supplies 10 or 10b according to Embodiment 1 or 7. Moreover, the capacitor C10 and the diode D10 according to Embodiment 11 can be applied to the multiple series circuits shown in Embodiments 2 to 5.
The current balancing device according to Embodiment 12 shown in
The resistor Rs detects currents flowing through the loads LD1, LD2, LD2, and LD3 collectively and outputs the current detection value to the PRC circuit 1 through the filter circuit. The PRC circuit 1 compares the current detection value with the reference voltage Vref and controls the ratio of on time of the switching element QH to on time of the switching element QL based on the error output thereof so that the currents flowing through the loads are held constant.
The waveform of each portion is basically the same as that shown in
According to the current balancing device of Embodiment 12, it is possible to obtain a similar effect to that of the current balancing device according to Embodiment 9 and to control and hold the current flowing through the load LD1 (LD2, LD3, and LD4) constant. Moreover, an end of each load is directly connected to the GND potential. Accordingly, noise generated in the current balancing device can be reduced at low cost.
Note that the current detector, the comparator, and the controller according to Embodiment 12 can be applied to the multiple series circuits shown in Embodiments 2 to 5. In addition, the filter circuit can be omitted.
The current balancing device according to Embodiment 13 shown in
The resistor Rs detects currents flowing through the loads LD1, LD2, LD3, and LD4 collectively and outputs a current detection value to the PFM circuit 1a through a filter circuit. The PFM circuit 1a compares the current detection value and the reference voltage Vref and controls the on-off frequency of the switching elements QH and QL based on an error output thereof so that the currents flowing through the loads are held constant.
Note that the waveform of each portion is basically the same as that shown in
According to the current balancing device of Embodiment 13, it is possible to provide the same operational effect as that of the current balancing device according to Embodiment 12. Embodiment 13 shown in
Note that the current detector, the comparator, and the controller according to Embodiment 13 can be applied to the multiple series circuits shown in Embodiments 2 to 5. In addition, the filter circuit can be omitted.
In
In the period ST1, the current from the secondary winding Ns flows through a first path of Ns→S2→N1→D1→C1→Ns, a second path of Ns→S3→N2→D2→C2→Ns, a third path of Ns→S4→N3→D3→C3→Ns, and a fourth path of Ns→S1→N4→D4→C4→Ns. Accordingly, the current flowing through the primary winding N1 is equal to the current flowing though the secondary winding S1, and the current flowing through the primary winding N2 is equal to the current flowing through the secondary winding S2. The currents flowing through the first to fourth paths are therefore equal.
The voltages of the above paths during the period ST1 are:
Vc1=Vns+Vs2−Vn1−Vf
Vc2=Vns+Vs3−Vn2−Vf
Vc3=Vns+Vs4−Vn3−Vf
Vc4=Vns+Vs1−Vn4−Vf,
where
Vcm is a voltage of the smoothing capacitor Cm (m is an integer of 1 to 4) (which is equal to the sum of forward voltage drops of LEDs ma to me),
Vns is a voltage of the winding Ns,
Vsm is a voltage of the winding Sm (m is an integer of 1 to 4),
Vnm is a voltage of the winding Nm (m is an integer of 1 to 4),
Vf is a forward voltage drop of the diode Dm (m is an integer of 1 to 4).
Vc is an average of Vc1, Vc2, Vc3, and Vc4 and is expressed as:
Vc=(Vc1+Vc2+Vc3+Vc4)/4
Vns=Vc+Vf
The voltage across the two windings connected in series in each path is:
Vs2−Vn1=Vc1−Vc
Vs3−Vn2=Vc2−Vc
Vs4−Vn3=Vc3−Vc
Vs1−Vn4=Vc4−Vc.
When the voltage Vc1, i.e., the sum of the forward voltage drops of LEDs 1a to 1e, is larger than the average value of the sums of the forward voltage drops of the LEDs ma to me, Vc1−Vc is positive, and the positive voltage is applied to the series circuit composed of the windings S2 and N1.
Meanwhile, when the voltage Vc1, i.e., the sum of the forward voltage drops of the LEDs 1a to 1e is smaller than the average value of the sums of the forward voltage drops of the LEDs ma to me, Vc1−Vc is negative, and the negative voltage is applied to the series circuit of the windings S2 and N1.
When Vcm (m is one of 1 to 4) is smaller than the average value Vc, a positive current flows through the corresponding magnetizing inductance Lm. When Vcm is larger than the average value Vc, a negative current flows through the corresponding magnetizing inductance Lm.
In the period ST2, the currents stored in the magnetizing inductances L1 to L4 of the balancing transformers T1a to T4a are reset. The negative currents stored in the magnetizing inductances L1 to L4 during the period ST1 generate a voltage opposite to the forward voltage of the diode Dm, and the diode Dm is subjected to a reverse voltage.
The conceivable condition for generating the largest reverse voltage during the reset period ST2 is that the deviation of Vc1, i.e., the sum of the forward voltage drops of the LEDs 1a to 1e, has a maximum value. In such a case, for example, it is considered that each of the deviations of the other Vc2, Vc3, and Vc4, i.e., the sums of the forward voltage drops of the LEDs xa to xe (x=2 to 4), has a minimum value. This is the case where the reverse voltage is applied to only the diode D1 during the reset period ST2.
The reverse voltage of the diode D1 in the aforementioned case is expressed as:
VD1=Vc1−Vns−Vn2+Vn1.
The forward voltages in the other second to fourth paths are expressed as:
Vc2=Vns+Vn3−Vn2−Vf
Vc3=Vns+Vn4−Vn3−Vf
Vc4=Vns+Vn1−Vn4−Vf
Accordingly, from the above three equations,
Vn1−Vn2=Vc2+Vc3+Vc4−3Vns+3Vf
The reverse voltage of the diode D1 is:
VD1=Vc1+Vc2+Vc3+Vc4−4Vns+3Vf
This reveals that the voltage of the diode to which the reverse voltage is applied during the reset period ST2 is small when the winding voltage Vns is positive.
In the operation waveforms shown in
VD1=Vc1+Vc2+Vc3+Vc4−4Vns+3Vf
VD1=4Vc−4(Vc−ΔV)+3Vf=4ΔV+3Vf.
Thus, the reverse voltage across the diode D1 can be suppressed to be low. Specifically, the switching element QL is turned off at time T4 after time T3 when the current flowing through the inductance L1 (L2, L3, and L4) becomes zero and the period to reset the balancing transformers T1a to T4a is terminated. Thereby, the reverse voltage of the diode D1 can be suppressed to be low.
If the switching element QL is turned off during the period ST2 to reset the balancing transformers T1a to T4a, the current having flown to the magnetizing inductance Lp starts to flow to the diode DH. The beginning of the primary winding of the transformer T then has a negative voltage, and the beginning of the secondary winding of the transformer T also has a negative voltage. Accordingly, Vns is expressed as:
Vns=(Vin−Vcri)/N,
where
N is turn ratio of the transformer T.
The reverse voltage of the diode D1 becomes very high value like:
VD1=Vc1+Vc2+Vc3+Vc4+4(Vin−Vcri)/N+3Vf.
As apparent from the above equation, since Vc1 is approximately equal to the total voltage of Vf of LED units (Vf of the LED units×the number of LED units connected in series), the reverse voltage across the diode D1 increases as the number of LED units connected in series increases.
Meanwhile, increasing the number of LED units connected in parallel requires high voltage diodes or restricts the breakdown voltage of the diodes. Accordingly, the number of LED units connected in series and the number of LED units connected in parallel cannot be increased. It is therefore very effective to control on and off of the switching elements QL and QH during the reset period ST2 so as to reverse the voltage of the transformer after the reset of the balancing transformers is finished.
In Embodiment 15 shown in
In
When the switching element Q1 is turned off at the time t0, the energy stored in the magnetizing inductance Lp generates counter-electromotive force, and the beginning of the winding Np then has a positive voltage. Accordingly, the beginning of the winding Ns has a positive voltage, and a current flows through the secondary winding Ns. The current on the primary side flows in the path of Lp→Np→Lp while currents on the secondary side flow in the paths of Ns→N1→D1→C1→Ns and Ns→S1→D2→C2→Ns. The currents are smoothed by the smoothing capacitors C1 and C2 and then flow to the loads LD1 and LD2.
As described in Embodiment 8, the balanced currents flow through the windings N1 and S1. At time t1, the energy stored in the magnetizing inductance Lp becomes zero, and the current I(NS) flowing through the winding Ns becomes zero. In the periods ST2 and ST3, during the period where the energy stored in the resonant capacitor Cv makes the voltage resonance with the magnetizing inductance Lp, the voltage of the winding Np gradually decreases because of the voltage resonance. Accordingly, the voltage of the winding Ns also gradually decreases, and the reverse voltage applied to the diodes D1 and D2 can be therefore reduced as shown in Embodiment 14. The switching element Q1 is turned on at time T3 to terminate the resonance period. The period ST2 is a period when the magnetizing inductance L1 of the transformer T′ is reset.
In contrast,
In the current balancing device according to Embodiment 15, the reverse voltages applied to the diodes D1 and D2 can be suppressed to be low. This makes it possible to use low voltage diodes or eliminate the diodes. The current balancing device can be therefore configured at low cost.
Moreover, it is possible to use a combination of some of the connection configuration of the balancing transformers shown in Embodiments 2 to 5. Furthermore, the current detector shown in Embodiments 12 and 13 may be configured to detect the current of the closed loop shown in Embodiment 3.
The current balancing device of the present invention can be applied to, for example, LED lighting apparatuses, LCD backlight (LCD B/L) modules, and LCD display units.
An LED lighting apparatus includes: a power converter configured to convert alternating current power from a commercial power supply into arbitrary alternating current power and to supply the alternating current; and a current balancing device in which currents each flowing through a corresponding one of multiple series circuits and at least one LED load are balanced based on an electromagnetic force generated at least one winding, the multiple series circuits being connected to an output of the power converter and each including the at least one winding, at least one rectifying element and the at least one LED load, which are connected in series.
An LCD B/L module includes an LCD cell and a current balancing device in which currents each flowing through a corresponding one of multiple series circuits and at least one LED load are balanced based on an electromagnetic force generated at least one winding, the multiple series circuits being connected to an output of a power converter and each including at least one winding, at least one rectifying element and at least one LED load for lighting the LCD cell, which are connected in series, the power converter converting alternating current power from a commercial alternating current power supply into arbitrary alternating power and then supplying the alternating current.
An LCD display unit includes: an LCD cell; a power converter configured to convert alternating current power from a commercial alternating current power supply into an arbitrary alternating current power and to supply the alternating current; a current balancing device in which currents each flowing through a corresponding one of multiple series circuits and at least one LED load are balanced based on an electromagnetic force generated at least one winding, the multiple series circuits being connected to an output of a power converter and each including at least one winding, at least one rectifying element and at least one LED load for lighting the LCD cell, which are connected in series. The LCD display unit is used in televisions, monitors, billboards, and the like.
Next, a description is given of a current balancing device of Embodiment 18. When a rectifying element is connected to a balancing transformer in order to rectify a current from the balancing transformer, in some cases, a counter-electromotive force is generated when the balancing transformer is reset, and the rectifying element is subjected to a large reverse voltage.
When the rectifying voltage (voltage of the rectifying element) is lower than the voltage of a secondary winding of a main transformer, the rectifying element connected to the balancing transformer is subjected to a current so as to be turned on at the reset of the balancing transformer. In contrast, when the rectifying voltage (voltage of the rectifying element) is higher than the voltage of the secondary winding of the main transformer, a counter-electromotive force is generated in a direction in which a reverse voltage is applied to the rectifying element at the reset of the balancing transformer. For suppressing the reverse voltage to be low, the circuit system and operation condition of the main circuit are restricted, thus resulting in a lower efficiency of the main circuit or an increase in size of the transformer of the main circuit.
In the current balancing device of Embodiment 18, a reverse voltage applied to a rectifying element connected to a balancing transformer in series is reduced.
The multiple series circuits are connected in parallel, and each series circuit includes the windings N1 and S1 (N2, S2 to N4, S4) of the balancing transformer T1 (T2 to T4), the diode D1 (D2 to D4), and the capacitor C1 (C2 to C4). The capacitor C1 (C2 to C4) is connected to the load LD1 (LD2 to LD4) through the resistor Rs.
The cathode of the diode D6 is connected to the balancing transformer T1 (T2 to T4), and the anode of the diode D6 is connected to the capacitor C1 (C2 to C4). The anode of the diode D5 is connected to an end of the secondary winding Ns of the transformer T, and the cathode of the diode D5 is connected to the balancing transformer T1 (T2 to T4).
The current balancing device of Embodiment 18 is characterized as follows. The diode D6 is added thereto, and when the secondary winding Ns of the positive winding has a negative voltage, a reset current is applied to the diode D6 in order to maintain the reset voltage at a certain voltage even when the secondary winding Ns has a negative voltage. The reverse voltage of the diode D1 (D2 to D4) connected to the balancing transformer T1 (T2 to T4) is suppressed to be low, thus achieving a high efficiency of the entire circuit and miniaturization thereof.
Next, a description is given of an operation of the current balancing device of Embodiment 18 configured as described above. First, as for the reverse voltage at the reset, the direction of the generated counter-electromotive force varies depending on the direction of a magnetizing current to be stored of the balancing transformer T1 (T2 to T4). In a steady state, the voltage of the secondary winding Ns of the transformer T of the main circuit is an average of voltage drops of the diodes D1 to D4, i.e., rectified voltages of the diodes D1 to D4 connected to the balancing transformers T1 to T4.
Accordingly, in some cases, the magnetizing current is stored in the direction that the diode D1 (D2 to D4) is charged when the balancing transformer T1 (T2 to T4) is reset (forward bias). In the other cases, the magnetizing current is stored so that the diode D1 (D2 to D4) is subjected to the reverse voltage when the balancing transformer T1 (T2 to t4) is reset (reverse bias).
The reverse voltage of the diode connected to the balancing transformer in series during the reset period has a maximum value Vr regardless of the connection configuration of the balancing transformers when one of the rectified voltages is higher than the average rectified voltage VC and the other rectified voltages are lower than the average rectified voltage VC:
VC1>VC>VC2=VC3 . . . =VCN
Herein, the number of balancing transformers and rectifying circuits connected in parallel is N, and
VC=(VC1+VC2+ . . . VCN)/N
At this time, the reverse voltage Vr1 of the diode D1 connected to the capacitor C1 in series is expressed as:
Vr1=VC1+VC2+ . . . +VCN−N·VNS+N·Vf (1)
VNS is a voltage of the secondary winding NS of the transformer T, and Vf is a forward voltage of the rectifying element.
The reverse voltage Vr1 thus varies depending on the voltage of the secondary winding Ns of the main circuit. In particular, when the voltage (VNS) of the secondary winding Ns of the main circuit becomes negative, the reverse voltage Vr1 is maximized. In other words, when the voltage of the secondary winding Ns of the transformer T is reversed during the period when the balancing transformer T1 (T2 to T4) is reset, a large reverse voltage Vr1 is generated.
In Embodiment 18, when the switching element Q1 is off, a current flows from the secondary winding Ns of the transformer T to the balancing transformer T1 (T2 to T4) through the diode D5.
Next, when the switching element Q1 is turned on and the voltage of the secondary winding Ns of the transformer T is reversed from the positive voltage to the negative voltage, a reset current flows to the balancing transformer T1 (T2 to T4) through the diode D6. In other words, the diode D6 is turned on and thereby the negative voltage of the secondary winding Ns is clamped to the forward voltage Vf.
At this time, the diode D5 is reverse biased, and there is no current flowing from the diode D6 to the diode D5. In other words, the provision of the diode D5 prevents the secondary winding Ns from being short circuited when the switching element Q1 is turned on.
The reverse voltage at the reset in the case where the number of balancing transformers and rectifying elements connected in parallel is N has the maximum value Vr when one rectified voltage is higher than the average rectified voltage VC and the other rectified voltages are lower than the average rectified voltage VC.
This is the case where VC1>VC>VC2=VC3 . . . =VCN,
where VC=(VC1+VC2+ . . . VCN)/N.
At this time, the reverse voltage Vr1 of the diode D1 connected to the capacitor C1 in series is expressed as:
Vr1=VC1+VC2+ . . . +VCN+N·Vf
In the circuit including the diodes D5 and D6, the reverse voltage Vr is −N·VNS (VNS is negative) smaller than that in the circuit not including the diodes 5 and 6. Accordingly, the diode D1 (D2 to D4) connected to the balancing transformer T1 (T2 to T4) can be configured to have a low breakdown voltage. Moreover, the aforementioned effect is not limited by the circuit configuration of the main circuit, the operation conditions thereof, or the configuration of the transformers of the main circuit, and the power supply unit can be therefore reduced in size and cost.
Note that the power supply unit 10 according to Embodiment 18 can be replaced with the power supply unit 10b shown in
According to the current balancing device of Embodiment 19, a reverse voltage Vr1 varies depending on the voltage of the secondary winding Ns of the main circuit and is maximized especially when the voltage (VNS) of the secondary winding Ns of the main circuit becomes negative.
In Embodiment 19, when the switching element Q1 is turned on and the voltage of the second winding Ns of the transformer T is reversed from a positive voltage to a negative voltage, a reset current flows from the secondary winding Ns through the voltage source VRS and the diode D6 to the balancing transformer T1 (T2 to T4).
At this time, the reverse voltage Vr1 of the diode D1 connected to the capacitor C1 in series is expressed as:
Vr1=VC1+VC2+ . . . +VCN−N·VRS+N·Vf (2)
Specifically, in the circuit not including the diodes D5 and D6 and the DC power supply VRS, as shown in Equation (1), Vr1 includes −N·VNS. Herein, −N·VNS is positive since VNS is negative, and the reverse voltage Vr becomes high.
In contrast, in the circuit including the diodes D5 and D6 and the DC power supply VRS in Embodiment 19, Vr1 includes −N·VRS as shown in Equation (2). Herein, since VRS is positive, and the reverse voltage Vr1 is therefore low. In other words, the reverse voltage can be reduced by the voltage of the DC power source VRS. Accordingly, the diode D1 (D2 to D4) connected to the balancing transformer T1 (T2 to T4) can be configured to have a low breakdown voltage.
Moreover, the voltage of the DC power source VRS is set to a value smaller than the average of voltages VLD1 to VLDN of the loads LD1 to LD4, and thereby the reverse voltage to be applied to the diodes connected to the balancing transformers in series can be made extremely low.
Accordingly, the number of LEDs connected in series in the LED units can be increased, and the number of balancing transformers can be reduced. The number of LED units connected in parallel can be therefore increased, thus reducing the number of transformers (the number of main circuits). It is therefore possible to considerably reduce the costs in a whole circuit and to configure a low-cost LED driver.
Note that the power supply unit 10 according to Embodiment 19 can be replaced with the power supply unit 10b shown in
In Embodiment 20 shown in
An end of the secondary winding Ns1 and an end of the secondary winding NS2 are connected to an anode of the diode D7, and a cathode of the diode D7 is connected to the other end of the secondary winding Ns2 and the capacitor C1 (C2 to C4) through the capacitor C7. The cathode of the diode D7 and an end of the capacitor C7 are connected to the anode of the diode D6, and the cathode of the diode D6 is connected to the balancing transformer T1 (T2 to T4). The anode of the diode D5 is connected to the other end of the secondary winding Ns1, and the cathode of the diode D5 is connected to the balancing transformer T1 (T2 to T4).
The other end of the secondary winding Ns1 and the anode of the diode D5 are connected to the cathode of the diode D10, and the anode of the diode D10 is connected to an end of the resistor Rs and an end of the capacitor C10, and the other end of the capacitor C10 is connected to the other end of the secondary winding Ns and capacitor C1 (C2 to C4).
According to Embodiment 20 thus configured, when the switching element QL is turned off from on, the voltage of the secondary winding Ns of the transformer T is reversed from a positive voltage to a negative voltage, and a reset current flows to the balancing transformer T1 (T2 to T4) through the capacitor C7 and the diode D6.
In Embodiment 20, the DC power supply VRS is generated by the diode D7 and the capacitor C7, and the reverse voltage Vr1 is low as similar to Embodiment 19. In other words, the reverse voltage can be suppressed to be low. Accordingly, the diode D1 (D2 to D4) connected to the balancing transformer T1 can be configured to have a low breakdown voltage.
The power supply unit 10c according to Embodiment 20 can be replaced with the power supply unit 10b shown in
In Embodiment 21, the secondary winding Ns1 is connected to multiple series circuits including the balancing transformer T1 (T2 to T4) and the diode D1 (D2 to D4) which are connected in series. The secondary winding Ns2 is connected to a power source in series which is composed of the diode D10 and capacitor C10. This allows reduction of the numbers of turns of the secondary windings Ns1 and Ns2 of the transformer Ta connected to the balancing transformer T1 (T2 to T4). In other words, the value of VNS of −N·VNS in the equation (1) above is reduced, and thereby the reverse voltage of the diode D1 (D2 to D4) connected to the balancing transformer T1 (T2 to T4) can be reduced.
By contrast, in the current balancing device shown in
The rectified negate voltage of the secondary winding Ns of the transformer T constitutes a power source composed of the diode D10 and the capacitor C10. Each of the loads LD1 to LD4 is connected to the series circuit of the capacitors C1 to C4 and the capacitor C10.
Like the current balancing device shown in
As described above, according to the embodiments of the present invention, currents supplied from an output of a power supply unit to multiple loads can be balanced based on an electromagnetic force generated at least one winding connected to at least one load in series. Moreover, since the currents are balanced by the electromagnetic force generated at the at least one winding, the loss due to variations of multiple load impedances can be reduced. It is therefore possible to reduce the loss in the circuit balancing the currents flowing through multiple loads having different impedances and to achieve a high efficiency.
The embodiments of the present invention can be applied to LED illumination and LED lighting apparatuses for lighting LEDs, for example, used in backlights of liquid crystal displays.
| Number | Date | Country | Kind |
|---|---|---|---|
| 2009-044620 | Feb 2009 | JP | national |
| 2009-106849 | Apr 2009 | JP | national |
