The invention will now be described, by way of non-limiting example, with reference to the figures of the annexed drawings, in which:
In the solution described herein, the signal processing occurs in current rather than in voltage.
For this purpose, the periodic voltage signals v1(t), v2(t), . . . , vN(t) to be processed and the reference voltage VREF are converted into as many respective currents i1(t), i2(t), . . . , iN(t) via appropriate transconductor elements Gm1, Gm2, . . . , GmN. Likewise, the reference voltage VREF is converted into a corresponding reference current IREF via a further transconductor element Gm0. Thus we have:
All the currents in question converge towards a single current-comparison node 100 where the comparison effectively takes place.
The current IREF is generated in such a way that its contribution to the node present with negative sign, and hence, on the direct branch downstream, a comparison current iI(t) is obtained, given by:
Said comparison current is sent to a current rectifier 110, the function of which is to reproduce at output only the positive portion of the input current waveform.
Consequently, the output current iO(t) at output from the current rectifier can be expressed symbolically as
i
o(t)=pos{iI(t)} (5)
where the function pos{·} is defined as follows:
As long as the input current iI(t) is maintained always negative, the output current iO(t) is zero and the output voltage vO(t) on the hold capacitor CO remains low as a result of the discharge-current generator IDIS.
Instead, if the input current presents positive during part of the period or for the entire duration thereof, also the output current is accordingly positive and tends to charge the capacitor CO, increasing the output voltage vO(t).
Assuming that the capacitor CO is of a value that is sufficiently high as to enable the ripple of the output voltage to be neglected, just the d.c. component VO of the voltage vO(t) can be considered.
Said d.c. component VO can be calculated by performing the balancing of the currents at the output node and evaluating the average thereof over a period. By so doing, the following expression is obtained:
where RO is the total impedance at the output node (resulting from the parallel between the output impedance of the current rectifier and the impedance that in the real circuit is inevitably associated to the generator IDIS), and the sign
When approximating Eq. 7, the quantity
Furthermore, assuming that the discharge current IDIS will determine in steady-state conditions a negligible contribution on VO, Eq. 8 can be approximated as follows:
which is formally equivalent to Eq. 1 that was to be implemented.
Eq. 9 confirms that a current comparison followed by a current rectifier closed, with high impedance, on a capacitor is equivalent to a voltage peak detector followed by a voltage comparison. In particular, it is to be noted that the diagram proposed in
From Eq. 9 it may incidentally be noted that the need for a high comparison gain results, in a circuit embodiment, in the need for a high impedance at the output node in
As regards the discharge-current generator IDIS, it is to be noted that, in spite of the error term ROIDIS caused thereby in Eq. 8, its presence proves indispensable to prevent the discharge times of CO from becoming too long with respect to the requirements of the application in question.
A possible circuit embodiment of
In the detail, the amplifier stage comprises four bipolar transistors (BJTs) T1, T2, T3, and T4.
Of these, the transistors T1 and T3, both with their base (i.e., their control terminal, represented by the gate in the case of a FET implementation) short-circuited on their collector (i.e., their current-inflow terminal or current-collection terminal, represented by the drain in the case of a FET implementation), are cascaded (i.e., set in series) between the supply voltage VCC and ground so as to have their respective collector-emitter paths traversed by one and the same current I1. The voltage VI undergoing comparison is supplied via a resistor R1 at the point of connection between the collector of the “bottom” transistor T1 and the emitter of the “top” transistor T3.
The transistors T2 and T4, the bases of which are not short-circuited on the collectors, are also cascaded (i.e., set in series) between the supply voltage VCC and ground so as to have their respective collector-emitter paths traversed by one and the same current I2. The reference voltage VREF is supplied via a resistor R2 at the point of connection between the collector of the “bottom” transistor T2 and the emitter of the “top” transistor T4, coming under which is the hold capacitor C0, across which the output voltage V0 is taken. The hold capacitor C0 is thus driven via the second pair of transistors T2, T4.
The embodiment of
Operation of said circuit can be explained by analogy with the schematic circuit diagram described previously: the capacitor CO and the current generator I2 in
The current rectifier 110 of
The current-comparison node 100 in
Finally, the voltage-to-current conversion provided by the scheme in
For an understanding of the function of the transistor T3 and of the generator I1 in
where iCo(t) is the current that flows through the capacitor CO, whilst vC1(t) and vC2(t) are the voltages at the collectors of the transistors T1 and T2, respectively. If the following condition is satisfied:
where AE1 and AE2 are the emitter areas of the transistors T1 and T2, then
Thanks to the presence of the transistor T3, the voltages vC1(t) and vC2(t) are approximately the same as one another and cancel out (since vC1−vC2=vBE4−vBE3 and vBB4≅vBE3), to obtain
Hence, it may be stated that the use of the transistor T3 (and of the generator I1 necessary for biasing it) is functional for eliminating the offset terms that arise in the voltage-to-current conversion performed by the resistances R1 and R2.
We note incidentally that, on account of the natural direction of the current rectifier T4 and of the generator I2, the response of the circuit in
In an implementation with MOSFETs, a similar improvement can be obtained by replacing the simple mirror T1-T2 in
This modification achieves the effect of increasing significantly the output impedance of the rectifier and therewith the comparison gain of the circuit (see Eq. 9).
It should be noted that the improvements that can be obtained via the versions illustrated in
In fact, the function implemented by the circuit in
which is true, provided that the following rule is respected:
The reason why the condition of Eq. 15 must be satisfied can be reduced to what has already been noted with reference Eq. 12 as regards the possibility that the offset terms (vC1(t) and vC2(t)) cancel out. In fact, if we write the balancing of the currents at the current-comparison node 100, we realize that said simplification occurs only if Eq. 15 is satisfied.
In fact, via the capacitor CM, it is possible to carry out a compensation by the Miller effect. As is known, the pole splitting associated to said compensation displaces the high-frequency pole at the output node, and hence the single low-frequency node that survives is the one at the collector of T4. As has already been noted, limiting the number of low-frequency poles of the loop gain proves advantageous in terms of closed-loop bandwidth and/or phase margin.
All the circuit schemes described can function also by replacing the current generators I1 and I2 with as many resistors, thus obtaining even simpler topologies, even though characterized by a smaller comparison gain.
With reference to what has been described up to now, we can state that the method and the circuits proposed present various advantages as compared to the known art.
In the first place, the devices proposed enable implementation of Eq. 1 with a number of poles that is lower as compared to the known art. In fact, if the classic solution shown in
Observation of the circuit diagrams provided confirms, then, that the solution described herein is characterized by an extreme topological simplicity. In an integrated embodiment, the topological simplicity conveniently results in a reduced area occupation.
In addition, in the circuit diagrams proposed, the input voltage-to-current conversion is implemented via resistors. This approach enables input voltages to be handled that are even higher than the supply level (Vcc) or lower than the ground reference. Consequently, the circuits proposed are characterized by an excellent input voltage dynamics.
Finally, it is worthwhile noting that the solution described enables an improved accuracy in peak detection as compared to the classic open-loop solutions (see
It should noted that, although the present invention has been described with reference to circuit diagrams using BJTs, it must be understood as being independent of the reference technology, it being possible for the aforesaid circuits to be implemented either by bipolar transistors or by field-effect transistors.
In addition, the present invention is not limited to the circuit implementations shown. Different topological modifications can, in effect, be made (such as, for example, replacement of the npn transistors with pnp transistors, and vice versa) without departing from the idea underlying the invention.
Even though the present invention has been described in relation to applications for telecommunications, it can find use in all the fields where closed-loop control is required of the peak of a periodic signal (or of a combination of periodic signals), irrespective of the nature, frequency and level of the signals involved.
Consequently, without prejudice to the principle of the invention, the details of construction and the embodiments may vary, even significantly, with respect to what is described and illustrated herein, purely by way of non-limiting example, without thereby departing from the scope of the invention, as defined in the annexed claims.
Number | Date | Country | Kind |
---|---|---|---|
EP 06425526.8 | Jul 2006 | EP | regional |