Patent Application
20240109430
The present disclosure relates to the technical field of motor modulation, and in particular, to a differential steering control method for a vehicle, which is applied to a motor system including a three-phase bridge type voltage inverter.
With the continuous development and popularization of electric vehicles, it is more and more important to optimize and precisely control a motor system. Over-modulation control of a motor is an important control strategy that can allow the motor to maintain stability beyond its normal operating range and provide an additional output capacity. However, existing over-modulation control methods have some problems in terms of control precision and stability. Therefore, a new technical solution is needed to improve the precision and effect of over-modulation control of a vehicle motor.
In the related technology, for example, Chinese patent document CN109672381B provides an over-modulation control method for a motor, which relates to a motor modulation technology. The method includes: acquiring an error signal and a current feedback signal between a feedback rotating speed of the motor and a given speed, and obtaining instruction voltage signals Uα* and Uβ* of the motor through coordinate transformation from a two-phase rotating coordinate system to a two-phase stationary coordinate system, obtaining a two-variable function F (θh, θr) of an angle pair (θh, θr) through space vector pulse width modulation (SVPWM), obtaining a modulation ratio MI, and calculating the modulation ratio MI; obtaining a value of the angle pair (θh, θr) according to a value of the modulation ratio MI because the modulation ratio MI is in one-to-one correspondence to the angle pair (θh, θr); and calculating a voltage vector required by the motor and a corresponding duty ratio according to the value of the angle pair (θh, θr), and obtaining six driving signals of an inverter. However, this scheme has the problems of trajectory limitations of an angle function and lack of proportion integration differentiation (PID) control. These problems may lead to low precision of over-modulation control of the motor.
For the problem of low precision of differential steering control of a vehicle in the prior art, the present disclosure provides a differential steering control method for a vehicle. The precision of differential steering control of a vehicle is improved by parallel application of speed closed-loop control, current closed-loop control, an over-modulation mode, and the like.
The objective of the present disclosure is achieved by the following technical solutions:
Embodiments of this specification provide a differential steering control method for a vehicle, including: acquiring rotating speeds of left and right motors, and obtaining feedback speeds; setting speeds of left and right wheels of the vehicle, and obtaining target speeds; comparing the feedback speeds with the target speeds, and obtaining given current values of currents of the left and right motors through a proportion integration differentiation (PID) controller; acquiring the currents of the left and right motor, and obtaining feedback current values; comparing the given current values with the feedback current values, and obtaining given voltage values of voltages of the left and right motor through the PID controller; calculating a modulation ratio M of voltage vectors of the left and right motor, and indicating, when M is greater than M0, that the motors have entered an over-modulation mode; in the over-modulation mode, calculating and obtaining, according to the given voltage values, voltage space vectors of the motors and corresponding duty ratios by using space vector pulse width modulation (PWM); outputting control signals to an inverter by using a digital signal processor (DSP) according to the voltage space vectors of the motors and the corresponding duty ratios, and performing over-modulation control on the motors; and enabling the motors to control differential steering of the vehicle through PWM signals.
Further, the step of comparing the given current values with the feedback current values, and obtaining given voltage values of voltages of the left and right motors through the PID controller includes: calculating difference values between the given current values and the feedback current values to obtain current errors e(k); and outputting the given voltage values u(k) according to the current errors e(k) by using the PID controller.
Further, the given voltage values u(k) are calculated through the following formula:
u(k)=Kp*e(k)+Ki*Σe(k)+Kd*(e(k)−e(k−1))
wherein Kp is a proportional gain of the PID controller; Ki is an integral gain of the PID controller; Kd is a differential gain of the PID controller; e(k) is an error between the given current values and the feedback current values at a current sampling moment; and e(k−1) is an error between the given current values and the feedback current values at a previous sampling moment.
Further, the step of calculating a modulation ratio M of voltage vectors of the left and right motor, and indicating, when M is greater than the threshold M0, that the motors have entered an over-modulation mode includes: a three-phase bridge type voltage inverter includes six non-zero voltage space vectors and two zero voltage space vectors, and a direct current bus voltage of the inverter is Udc; a radius Umax of an inscribed circle of the hexagon is determined if the six non-zero voltage space vectors form a hexagon; and when a given voltage vector exceeds a range of the hexagon, that is, when the modulation ratio M is greater than the threshold M0, the motors enter the over-modulation mode; the modulation ratio M is calculated through the following formula:
M=U
max
/U
dc
wherein Umax is the radius of the inscribed circle of the hexagon, which represents a maximum output voltage vector amplitude of the inverter.
Further, the step of: in the over-modulation mode, calculating and obtaining, according to the given voltage values, voltage space vectors of the motors and corresponding duty ratios by using PWM includes: determining whether the modulation ratio M is greater than the threshold M0; when the modulation ratio M is greater than the threshold M0, indicating that the given voltage vector Uref exceeds the range of the hexagon, and selecting a substitute voltage vector Ualt of the given voltage vector Uref; calculating action time T0 when the substitute voltage vector is a zero voltage vector U0; and calculating, according to the action time T0 of the zero voltage vector, the voltage space vector Uout and the corresponding duty ratio Dout by using the PWM.
Further, the action time T0 of the zero voltage vector U0 is used as a start point of determining over-modulation; when T0 is less than zero, it is determined that the over-modulation mode is activated; action time Ti of a basic voltage vector of an ith moment is calculated, wherein
T
i
=T
s*sin θ1*Umax/Udc
action time Ti+1 of a basic voltage vector of an (i+1)th moment is calculated, wherein
T
i+1
=T
s*sin θ2*Umax/Udc
action time Ts of the given voltage vector Uref within an original PWM cycle before over-modulation occurs is calculated;
T0 is calculated through the following formula:
T
0
=T
s
−T
i
−T
i+1
wherein θ1 and θ2 are electric angles of the basic voltage vectors at the ith moment and the (i+1)th moment; an angle sum of θ1 and θ2 is π/3; Uref is the given voltage vector; Ualt is the substitute voltage vector; U0 is the zero voltage vector; T0 is the action time of the zero voltage vector; Ti and Ti+1 are respectively the action times of the basic voltage vector at the ith moment and the (i+1)th moment; Ts is action time of the given voltage vector Uref within a current PWM cycle; Umax is the direct current bus voltage; and Udc is a direct current voltage.
Further, the voltage space vector Uout and the corresponding duty ratio Dout are calculated by using the PWM according to the action time T0 of the zero voltage vector; a position of the voltage space vector is determined by using a space vector PWM calculation method according to the action time T0 of the zero voltage vector U0, the action times Ti and Ti+1 of the basic voltage vectors, and time Ts of a PWM cycle; and a projection of the voltage space vector under a three-phase coordinate and a time proportion corresponding to the voltage space vector are calculated as a duty ratio of each bridge arm of the inverter according to the position of the voltage space vector, a maximum amplitude Umax of the direct current bus voltage, and the direct current voltage Udc.
Further, the step of outputting control signals to an inverter by using a DSP according to the voltage space vectors of the motors and the corresponding duty ratios, and performing over-modulation control on the motor includes: adding an output signal of a speed closed-loop PID controller to an output signal of a current closed-loop PID to obtain the given voltage value; inputting the given voltage value to a reverse Park transformation module, and transforming the given voltage value into a given voltage under an αβ coordinate; inputting the given voltage under the αβ coordinate to a space vector PWM module for over-modulation to obtain the voltage space vector and the duty ratio corresponding to each motor; outputting the voltage space vector and the duty ratio to six metal oxide semiconductor (MOS) transistors configured to control upper and lower bridge arms of the inverter to perform over-modulation control on the motor; acquiring feedback current signals from two phases of an output end of the motor; inputting the acquired two phases of currents to a current sampling module to obtain sampled current values; calculating and obtaining three phases of currents according to the sampled current values and the Kirchhoff's Current Law; inputting the three phases of currents to a Clarke transformation module, and transforming the three phases of currents to be under the αβ coordinate; inputting the currents under the αβ coordinate to the Park transformation module, and transforming the currents to be under a dq coordinate; and taking the currents under the dq coordinate as feedback signals, and inputting the feedback signals to the current closed-loop PID controller.
Further, the step of enabling the motors to control differential steering of the vehicle through PWM signals includes: outputting, by an upper computer, target rotating speed control signals ωl1 and ωr1 of the left and right wheels of the vehicle; inputting ωl1 and ωr1 to the speed closed-loop PID controller; outputting, by the speed closed-loop PID controller, PWM control signals of the motors of the left and right wheels; controlling, by the motors of the left and right wheels, rotating speeds ωl2 and ωr2 of the left and right wheels of the vehicle according to the PWM signals; and acquiring the rotating speeds ωl2 and ωr2 of the left and right wheels of the vehicle, and feeding back the rotating speeds to the speed closed-loop PID controller, wherein when the rotating speeds ωl2 and ωr2 of the left and right wheels of the vehicle are equal, the vehicle is driven along a straight line; and when the rotating speeds ωl2 and ωr2 of the left and right wheels of the vehicle are unequal, a driving trajectory of the vehicle is a concentric arc.
Compared with the prior art, the present disclosure has the advantages below:
A method and system provided by the embodiments of this specification will be described in detail below in conjunction with accompanying drawings.
S110, speeds of left and right wheels of the vehicle are set, and target speeds are obtained; rotating speeds of left and right motors of the vehicle are acquired through a sensor, an encoder, or another device, so as to obtain feedback signals of actual rotating speeds of the motors. The target speeds of the left and right wheels of the vehicle are set according to a control requirement of the vehicle or a user input, and the target speeds are used as target values of differential steering control. S120, given current values of currents of the left and right motor are obtained through a PID controller: The given current values of the currents of the left and right motors are calculated using the PID controller by comparing the feedback speeds with the target speeds. The values are configured to control rotations of the motors. S130, the given current values are compared with feedback current values, and given voltage values of voltages of the left and right motors are obtained through the PID controller; and actual values of the currents of the left and right motor are acquired by a current sensor or another device, so as to obtain feedback signals of the currents. The given current values are compared with the feedback current values using the PID controller, and the given voltage values of the voltages of the left and right motors are calculated. The values are configured to control voltage outputs of the motors. S140, a modulation ratio M of voltage vectors of the left and right motors are calculated, and it is indicated, when M is greater than M0, that the motors have entered an over-modulation mode: The modulation ratio M of the vectors of the left and right voltage motors is obtained by calculation. S150, in the over-modulation mode, voltage space vectors of the motors and corresponding duty ratios are calculated and obtained according to the given voltage values by using space vector PWM. These values are configured to control a voltage signal output by an inverter. S160, control signals are output to the inverter by using a DSP according to the voltage space vectors of the motors and the corresponding duty ratios, and over-modulation control is performed on the motors. S170, the motors control differential steering of the vehicle through PWM signals, and a modulation signal output by the inverter is converted into a motor control signal by using the PWM signals. The steering control of the vehicle is achieved by controlling the rotating speeds and running directions of the left and right wheels.
Difference values between the feedback speeds and the target speeds of the motors are calculated, and speed errors ν(k) of the motors are obtained. The given current values i(k) are output using the PID controller according to the speed errors ν(k). The speed errors are obtained by comparing the feedback speeds with the target speeds, and the PID controller can adjust the given current values in real time according to the errors. The real-time adjustment can make a quick response to a change of a steering requirement and precisely control output currents of the motors, so as to provide precise steering control for the vehicle. Integral and differential parts of the PID controller can consider historical changes and change rates of the errors, thus suppressing oscillations of a system. Due to the integral and differential effects of the PID controller, the system can better adapt to the change of the steering requirement, and the stability of the system is improved.
Difference values between the given current values and the feedback current values are calculated, and current errors e(k) are obtained. The current errors are obtained by comparing the given current values with the feedback current values, and the PID controller can adjust the given voltage values in real time according to the errors. By the precise current control, the vehicle can achieve more precise differential adjustment in a steering process, so that the stability and adjustability of steering are improved. The given voltage values u(k) are output according to the current errors e(k) by using the PID controller. The given voltage values u(k) are calculated through the following formula:
u(k)=Kp*e(k)+Ki*Σe(k)+Kd*(e(k)−e(k−1))
where Kp is a proportional gain of the PID controller; Ki is an integral gain; Kd is a differential gain; e(k) is an error between the given current values and the feedback current values at a current sampling moment; and e(k−1) is an error between the given current values and the feedback current values at a previous sampling moment. The proportional item Kp can enable the currents to make a quick response to changes in the given values, which improves the instantaneity of steering. The integral item Ki can eliminate a steady state error, which improves the stability. The differential item Kd improves the sensitivity to current changes and is conductive to suppressing motor disturbance. Use of the current error e(k) and the previous error e(k−1) can further improve the stability. The PID closed-loop control causes the currents to precisely follow the given values, so as to achieve precise torque control. The precise torque control is then converted into wheel rotating speed control, so as to finally achieve high-precision differential steering. Two closed-loop controls are equivalent to double insurances, which greatly improves the stability and precision of steering control.
In this embodiment, for the PID controller, Kp=0.8, Ki=0.5, and Kd=0.2. The given current value of the current sampling moment is set to be I_setpoint; the feedback current value of the current sampling moment is set to be I_feedback; the given current value of the previous sampling moment is set to be I_setpoint_prev; and the feedback current value of the previous sampling moment is set to be I_feedback_prev. Firstly, the current error (e) is calculated: e=I_setpoint−I_feedback; and a control signal is calculated: PID_output=P*e+I*Σe+D*(e−e_prev).
The PID controller can quickly adjust and output the given voltage values according to the current errors to make the current errors tend to be zero. During steering, the vehicle can quickly adjust differential speeds, which reduces a steering error and delay and achieves more precise steering control. The integral part can eliminate the steady state error to ensure that the current errors tend to be zero for a long time. The differential part can suppress the oscillations of the system to cause the control system to be more stable. Stable steering control means that the vehicle can steer at a more stable speed and differential speed and maintain steadiness and controllability of driving.
In this embodiment, an encoder of the left wheel acquires a feedback speed ω1=10 rad/s, and a target speed of the right wheel is set to be ωr=15 rad/s. ωl is compared with ωr, and the PID controller outputs a given current value I*=20 A. A current feedback Il=18 A is acquired, and the PID controller outputs a given voltage value *=300V. A modulation ratio M=U*/Udc=1.2 is calculated, and an over-modulation mode is activated. An over-modulation voltage vector Uover is equal to 340 V, and a duty ratio D=85% is calculated by using SVPWM. A DSP outputs a PWM signal to drive an inverter to achieve over-modulation control. Due to the left wheel ωl=10 rad/s and the right wheel ωr=15 rad/s, the vehicle achieves precise differential steering. The over-modulation control is achieved by calculating the modulation ratio, so that a larger motor torque can be obtained, and the precision and flexibility of differential steering are improved.
M=U
max
/U
dc
where Umax is the radius of the inscribed circle of the hexagon, which represents a maximum output voltage vector amplitude of the inverter.
Specifically, whether the motors are in the over-modulation mode can be determined by calculating the modulation ratio M and comparing the modulation ratio M with the preset threshold M0. The over-modulation mode means that the given voltage vector exceeds the normal range of the hexagon. The over-modulation control can improve the precision and responsiveness of motor outputs can be improved. A range of output voltages of the motors can be limited by determining the radius of the inscribed circle of the hexagon composed of the six non-zero voltage space vectors. This can ensure that the output voltages of the motors can be within a controllable range, avoid an extremely large or small output, and improve the precision of differential steering control. The over-modulation control can control the differential steering of the vehicle more precisely, thus improving the steering precision and response speed.
More specifically, whether the voltages exceed the range of the hexagon is determined in the over-modulation mode by comparing the modulation ratio with the preset threshold M0. When the modulation ratio M is greater than the threshold M0, it is indicated that the given voltage vector Uref exceeds the range of the hexagon, and a substitute voltage vector Ualt of the given voltage vector Uref is selected, where Uref is usually a neighboring effective voltage vector, which is a voltage vector close to a boundary of the hexagon. Through Uref, the voltage output by each motor can be still kept in the controllable range in the over-modulation mode. Action time T0 when Uref is a zero voltage vector U0 is calculated. The substitute voltage vector is selected, so that a zero voltage vector may be generated. Action time of the zero voltage vector is calculated, that is, a length of the action time of the substitute voltage vector is determined. The action time will be configured for subsequent PWM calculation. It is general that M0 can be set to be 1.05 to 1.2 times a ratio of a maximum output voltage of an inverter to a direct current bus voltage. In this embodiment, the threshold M0 is preferably 1.
The voltage space vector Uout and the corresponding duty ratio Dout are calculated by using the PWM according to the action time T0 of the zero voltage vector. The action time of a pure zero voltage vector will allocate a zero level to a corresponding PWM waveform, and the action time of the substitute voltage vector will be configured to allocate a non-zero level for a corresponding PWM waveform.
In this embodiment, the three-phase bridge type inverter outputs the six non-zero basic voltage vectors to form the hexagon. The radius of the inscribed circle is Umax=400V. The bus voltage of the inverter is Udc=380V. It is defined that the threshold is K=1.05. A give voltage vector amplitude is calculated to be U=420V. U, Umax, and Udc are substituted into a formula to calculate the modulation ratio: M=U/Udc=420/380=1.11>K. It is determined that the motor has entered the over-modulation mode. In the over-modulation mode, a space vector and a PWM duty ratio are recalculated. The over-modulation mode can be flexibly used by determining the modulation ratio threshold. During over-modulation, a larger torque can be generated, thus improving the precision and flexibility of differential steering.
The action time T0 of the zero voltage vector U0 is used as a start point of determining over-modulation; when T0 is less than zero, it is determined that the over-modulation mode is activated. More precise steering control can be achieved by calculating the action time of the basic voltage vector and adjusting the action time according to the position of the given voltage vector. This can improve the accuracy and stability of steering of the vehicle.
Action time Ti of the basic voltage vector of an ith moment is calculated, wherein Ti=Ts*sin θ1*Umax/Udc; action time Ti+1 of the basic voltage vector of an (i+l)th moment is calculated, wherein Ti+1=Ts*sin θ2*Umax/Udc; action time Ts of the given voltage vector Uref within an original PWM cycle before over-modulation occurs is calculated; T0 is calculated through the following formula: T0=Ts−Ti−Ti+1; and Ts is the action time of the given voltage vector within the original PWM cycle, which reflects a basic feature of the voltage vector. sin θ1 and sin θ2 are respectively proportions for calculating the action times of two adjacent vectors, thus achieving π/3 phase distribution and improving the PWM control performance. Umax is the maximum output voltage amplitude of the inverter. By comparison with Udc, an over-modulation condition is calculated, and a modulation range of the inverter is expanded. Ti is calculated according to a ratio of Umax to Udc, so that reasonable allocation of the voltage amplitude after over-modulation is achieved, and the over-modulation control effect is optimized. The time proportions of the adjacent vectors are distributed reasonably, which is conductive to reducing impact and improving the stability, thus improving the precision of steering control of the vehicle.
Specifically, θ1 and θ2 are electric angles of the basic voltage vectors at the ith moment and the (i+1)th moment, and an angle sum of θ1 and θ2 is π/3, so that voltage vectors of adjacent moments are distributed alternately. Positions of the voltage vectors change smoothly, which is conductive to reducing fluctuations of magnetic energy of an iron core of the moving motor and reducing vibrations and noises. The adjacent vectors are distributed at 60 degrees, so that the calculation of the SVPWM can be simplified. A simple vector superposition rule is used to reduce the calculation amount of the control system and lower the implementation difficulty. The adjacent vectors distributed at 60 degrees are matched with a physical characteristic of a three-phase winding, which is conductive to balancing a three-phase current and reducing current fluctuations. The adjacent vectors are distributed at 60 degrees, which can avoid a high current impact caused by large changes in the voltage vectors. The SVPWM is a PWM method in motor vector control. In this method, by use of the concept of the voltage space vector, eight on-off states of the inverter are expressed by six non-zero voltage vectors and two zero voltage vectors. The six non-zero voltage vectors form a hexagonal voltage space vector range. A toggle state and a corresponding duty ratio of each bridge arm of the inverter can be determined by calculating a projection of the given voltage vector in the hexagonal region, so that a size and position of the voltage space vector of the motor can be flexibly controlled.
Ts is action time of the given voltage vector Uref within a current PWM cycle; Umax is the direct current bus voltage; and Udc is a direct current voltage. Uref is the given voltage vector; Ualt is the substitute voltage vector; U0 is the zero voltage vector; T0 is the action time of the zero voltage vector; and Ti and Ti+1 are respectively the action times of the basic voltage vector at the ith moment and the (i+1)th moment. During SVPWM control, to achieve smooth transitioning, the zero voltage vector needs to be added, that is, the action time of the zero voltage vector is added. This over-modulation control can effectively make use of the direct current bus voltage. The voltage modulation range is expanded without increasing a current and a voltage, thus increasing the utilization rates of both the inverter and the motor and improving the capacity and efficiency of the system.
Specifically, Ts is the action time of the given voltage vector within the original PWM cycle before over-modulation occurs. A calculation formula of the action time is: Ts=Tpwm*Ti/Tpwm, wherein Tpwm is the time of the original PWM cycle, and Ti is the action time of the given voltage vector within the original PWM cycle. That is, before the over-modulation, within a PWM cycle, a proportion of the action time of the given voltage vector in the PWM cycle is Ts.
In this embodiment, the position of the given voltage vector is θ=30°, the maximum voltage is Umax=300 V, and the bus voltage is Udc=220 V. The action times of the adjacent vectors are calculated to be T1=0.5 ms, and T2=0.3 ms. The action time within the original PWM cycle is Ts=T1+T2=0.8 ms. If the action time of the zero vector is calculated to be T0=Ts−T1−T2=−0.1 ms<0, it is determined that over-modulation is activated. Umax is compared with Udc to determine that an over-modulation condition is satisfied. It is calculated according to T1/T2=0.5/0.3=1.67 that T1′=0.4 ms, and T2′=0.24 ms. Action time within a new PWM cycle is Ts′=T1′+T2′=0.64 ms. The position of a voltage vector angle after adjustment is kept at 30°, thus achieving precise steering control.
The space vector PWM calculation method is used to determine the position of the voltage space vector. The differential steering control system can more precisely obtain a desired motor output by precisely calculating the position of the voltage space vector and apply the motor output to a steering operation. The projection of the voltage space vector under the three-phase coordinate and the time proportion corresponding to the voltage space vector can be calculated as the duty ratio of each bridge arm of the inverter on the basis of the position of the voltage space vector, the maximum amplitude of the direct current bus voltage, and the direct current voltage. By precisely calculating duty ratio, the system can precisely adjust an on-off state of the inverter to control a voltage output of the motor. It is conductive to achieving precise steering control and providing an accurate steering torque, thus improving the precision and controllability of differential steering.
In this embodiment, it is set that the action time of the zero voltage vector is T0=0.2 ms, the action time of an over-modulation voltage vector is Ta=0.5 ms, and the PWM cycle is Ts=1 ms. The SVPWM method is used to calculate the position of the space vector to be 60° according to T0, Ta, and Ts. The direct current bus voltage is Udc=300 V, and the given voltage vector amplitude is Uref=340 V, which is greater than Udc, so that it is in an over-modulation state. Three phases of voltages VA=210 V, VB=344 V, and VC=188 V are calculated according to the position 60° of the space vector, Udc=300 V, and Uref=340 V. The duty ratios of the three phases of voltages are respectively VA/Udc=70%, VB/Udc=98%, and VC/Udc=63%.
In this embodiment, the feedback signals ia and ib of the two phases of currents are acquired from the output end of the motor. A third phase of current ic can be deduced according to the Kirchhoff's Current Law: ic=−ia−ib, so that feedback signals of the three phases of currents can be obtained: ia, ib, and ic. The three phases of currents are input to the Clarke transformation module for αβ transformation: The purpose of Clarke transformation is to transform the three phases of currents to be under the αβ rotating coordinate system for subsequent Park transformation to the dq coordinate system. In the αβ coordinate system, current components consistent with directions of the magnetic axis and a potential axis can be respectively obtained.
Two driving motors can be separately and precisely controlled using vector control and the Clarke and Park transformations to harmoniously control a steering motion of the vehicle. The feedback currents achieve a current closed-loop, so as to suppress the influence of the instability factors of the motors on steering. The over-modulation control increases the torque, and the closed-loop control improves the precision. Combining the over-modulation control and the closed-loop control greatly improve the precision of steering control of the vehicle.
Specifically, the corresponding voltage space vector and duty ratio are calculated through the space vector PWM module for over-modulation according to the given voltage under the coordinate, and are then output to the six MOS transistors configured to control the upper and lower bridge arms of the inverter. The feedback signals of the two phases of currents are acquired from the output end of the motor and are sampled through the current sampling module for calculation to obtain the three-phases of currents. The currents are transformed to be under the coordinate through Clarke transformation and the Park transformation and are used as the feedback signals input to the current closed-loop PID controller. This current feedback control can monitor changes of currents of the motors in real time and adjust the currents to keep the stability and accuracy of outputs of the motors.
In this embodiment, a target angular speed ω*=10 deg/s is input, and a given current value I*=15 A is output via the speed PID controller. I* is input to an over-modulation determining module. If a given voltage exceeds the hexagonal range, the over-modulation is confirmed. An over-modulation vector Uover=380 V is selected, which is transformed to Uα=340 V, Uβ=180 V under the αβ coordinate via reverse Park transformation. Uαβ is input to the SVPWM module, and a PWM signal is output. The duty ratio is 75%. The PWM signal drives the inverter to output an over-modulation voltage Uover, so that the motor achieves a rotating speed of 10 deg/s. The feedback current I=13 A is input to the current PID controller. In this way, the over-modulation control can provide a large torque and enable the vehicle to smoothly complete a steering radius of d/ω*=0.2 m.
The upper computer outputs the target speeds of the left and right wheels. The PWM control signals are calculated and output through the speed closed-loop PID. The rotating speeds of the left and right wheels are adjusted in real time to achieve follow-up control of the rotating speeds. When the rotating speeds of the left and right wheels are consistent, the vehicle is driven along the straight line; and when the rotating speeds are inconsistent, the vehicle has a differential speed and precisely achieves an arc steering motion. Compared with open-loop control, the closed-loop control can eliminate the influence of load disturbance and parameter errors. Compared with single-closed-loop control, double-closed-loop control enhances the control effect, and the steering control is precise and flexible.
Specifically, through the closed-loop PID control and the rotating speed feedbacks, the system can precisely adjust the output of the motor, so as to achieve accurate steering control. A response to differential steering is more precise. The vehicle can be driven according to an expected steering direction. By the adoption of the PID control and rotating speed feedback mechanisms, the system can monitor and adjust the rotating speeds of the wheels in real time, so as to make a quick response to a steering requirement of the vehicle.
In this embodiment, the upper computer respectively outputs the target rotating speed control signals ωl1 and ωr1 of the left and right wheels, for example, ωl1=10 rad/s and ωr1=15 rad/s; ωl1 and ωr1 are input to the speed PID controller, and the PWM control signals Duty_l and Duty_r are output; the motors of the left and right wheels control the rotating speeds ωl2=10 rad/s and ωr2=15 rad/s of the left and right wheels according to Duty_l and Duty_r; and the rotating speed feedbacks ωl2 and ωr2 of the left and right wheels are acquired and input to the speed PID controller. In this embodiment, vehicle parameters are as follows: If a wheel track is =1 m, a steering radius of the vehicle is R=d/(ωl2/ωr2−ωr2/ωl2)=5 m.
In summary, according to the flow of the over-modulation algorithm of the motor shown in the figure, the PWM signals are used to control differential steering of the vehicle, so that the precision of differential steering control of the vehicle can be improved. These technical features achieve precise steering control, improve the instantaneity and responsiveness, and improve the stability, thus improving the precision and controllability of differential steering control.
| Number | Date | Country | Kind |
|---|---|---|---|
| 202311268835.0 | Sep 2023 | CN | national |
