Display device

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a National Phase filing under 35 U.S.C. §371 of International Application No. PCT/JP2011/072920 filed on Oct. 5, 2011, and which claims priority to Japanese Patent Application No. 2010-262534 filed on Nov. 25, 2010.

TECHNICAL FIELD

The present invention relates to an active matrix type display device.

BACKGROUND ART

A mobile terminal such as a mobile telephone or a mobile game machine uses a liquid crystal display device as its displaying means, in general. In addition, since the mobile telephone is driven by a battery, it is strongly required to reduce power consumption. Therefore, information such as a time or remaining battery level which needs to be constantly displayed is displayed on a reflective subpanel in some mobile telephones. In addition, recently, both normal display by way of a full-color display and reflective constant display are required to be realized on the same main panel.

FIG. 38 shows an equivalent circuit of a pixel circuit of a general active matrix type liquid crystal display device. In addition, FIG. 39 shows a circuit arrangement example of the active matrix type liquid crystal display device having m×n pixels. In addition, each of the numbers m and n is two or more integer.

As shown in FIG. 39, a switch element composed of a thin film transistor (TFT) is provided at each intersecting point of m source lines SL1, SL2, . . . , SLm and n scanning lines GL1, GL2, . . . , GLn. In FIG. 38, the source lines SL1, SL2, . . . , SLm are represented by a source line SL, and similarly, the scanning lines GL1, GL2, . . . , GLn are represented by a scanning line GL.

As shown in FIG. 38, a liquid crystal capacitive element Clc and an auxiliary capacitive element Cs are connected in parallel through the TFT. The liquid crystal capacitive element Clc has a laminated structure in which a liquid crystal layer is provided between a pixel electrode 20 and an opposite electrode 80. The opposite electrode is also referred to as a common electrode.

In addition, in FIG. 39, as for the pixel circuit, the TFT and the pixel electrode (black rectangular part) are simply shown.

The auxiliary capacity Cs has one end (one electrode) connected to the pixel electrode 20, and the other end (the other electrode) connected to an auxiliary capacity line CSL, and is provided to stabilize a voltage of the pixel data held in the pixel electrode 20. The auxiliary capacity Cs has an effect of preventing the voltage of the pixel data held in the pixel electrode from fluctuating due to a leak current of the TFT, a fluctuation of electric capacity of the liquid crystal capacitive element Clc between a black display and a white display due to dielectric constant anisotropy of liquid crystal molecules, and a voltage fluctuation generated through parasitic capacitance between the pixel electrode and a surrounding wiring. By sequentially controlling a voltage of the scanning line, the TFT connected to the scanning line is turned on, and a voltage of pixel data supplied to the source line is written in the corresponding pixel electrode with respect to each scanning line.

As for the normal display by way of the full-color display, even when display contents are still images, the same display contents are repeatedly written in the same pixel with respect to each frame. Thus, the voltage of the pixel data held in the pixel electrode is updated, so that the voltage fluctuation of the pixel data is minimized, and a high-quality display of the still image can be maintained.

Power consumption to drive the liquid crystal display device is mainly dominated by power consumption to drive a source line by a source driver, and roughly expressed by a relational expression shown in the following formula 1, wherein P represents power consumption, f represents a refreshing rate (the number of times to perform a refreshing action for one frame per unit time), C represents load capacity driven by the source driver, V represents a drive voltage of the source driver, n represents the number of scanning lines, and m represents the number of source lines. Here, the refreshing action means an action to apply the voltage to the pixel electrode through the source line while maintaining the display contents.

P∝f·C·V²·n·m (Formula 1)

Meanwhile, in the case of the constant display, since the display contents are still images, it is not always necessary to update the voltage of the pixel data with respect to each frame. Therefore, in order to further reduce the power consumption, a refreshing frequency is lowered at the time of this constant display. However, when the refreshing frequency is lowered, the pixel data voltage held in the pixel electrode fluctuates due to a leak current of the TFT. The voltage fluctuation leads to a fluctuation of display brightness (transmittance of liquid crystal) of each pixel, and this is recognized as a flicker. In addition, since an average potential is lowered in each frame period, a display quality could be lowered such that a sufficient contrast cannot be provided.

Here, as a method to solve the problem that the display quality is lowered due to the lowering of the refreshing frequency and to cut the power consumption at the same time in the constant display of the still image of the remaining battery level or the time display, a configuration is disclosed in the following patent document 1. According to the configuration disclosed in the patent document 1, a liquid crystal display can be implemented by both transmissive and reflective functions, and moreover, a memory part is provided in a pixel circuit in a pixel region in which the reflective liquid crystal display can be provided. This memory part holds information to be displayed in the reflective liquid crystal display part as a voltage signal. At the time of the reflective liquid crystal display, information corresponding to this voltage is displayed when the pixel circuit reads the voltage held in the memory part.

According to the patent document 1, since the memory part is composed of a SRAM, and the voltage signal is statically held, the refreshing action is not needed, so that the display quality can be maintained, and the power consumption is reduced at the same time.

PRIOR ART DOCUMENT
Patent Document

Patent document 1: Japanese Unexamined Patent Publication No. 2007-334224

SUMMARY OF THE INVENTION
Problems to be Solved by the Invention

However, when the above-described configuration is employed in the liquid crystal display device such as the mobile telephone, it is necessary to provide a memory part to store pixel data with respect to each pixel or each pixel group, in addition to an auxiliary capacitive element to hold the voltage of the pixel data serving as analog information at the time of a normal action. This causes an increase in the number of the element and the number of signal lines to be formed on an array substrate (active matrix substrate) in the display part of the liquid crystal display device, so that an aperture ratio is lowered in a transmissive mode. In addition, when a polarity inversion drive circuit to perform AC driving for the liquid crystal is provided together with the above memory part, the aperture ratio is further lowered. Thus, when the aperture ratio is lowered due to the increase in the number of the elements and the signal lines, brightness of the display image is lowered in the normal display mode.

In addition, at least two gradations are assumed in the above constant display mode, but a multicolored display is required to be implemented in the constant display mode. However, when such display mode is implemented in the conventional configuration, the number of the required memory parts increases as a matter of course, and accordingly the number of the elements and the number of the signal lines further increase.

The present invention was made in view of the above problems, and it is an object of the present invention to provide a pixel circuit and a display device in which a liquid crystal display is prevented from deteriorating and a display quality is prevented from being lowered at low power consumption without lowering an aperture ratio, and especially to enable a refreshing action to be performed in a multi-colored display mode while preventing an increase in the number of the elements and signal lines.

Means for Solving the Problem

In order to attain the above object, a display device according to the present invention includes

a pixel circuit array comprising a plurality of pixel circuits arranged in a row direction and a column direction, respectively, wherein

each of the pixel circuits has: a display element part including a unit display element; an internal node composing a part of the display element part, for holding a pixel data voltage applied to the display element part; a first switch circuit; a second switch circuit; and a control circuit including a first capacitive element,

the second switch circuit has one end connected to the internal node and has a series circuit of a first transistor element and a diode element,

the control circuit has a series circuit of the first capacitive element and a second transistor element, a first terminal of the second transistor element is connected to the internal node, and a second terminal of the second transistor element is connected to a control terminal of the first transistor and one end of the first capacitive element to form an output node,

the first switch circuit has one end connected to the internal node, and includes a third transistor element,

a common electrode is connected to a terminal opposite to a terminal connected to the internal node, among terminals of the unit display element,

the other end of the first switch circuit and the other end of the second switch circuit in each of the pixel circuits arranged in the same column are connected to one of data signal lines in common,

a control terminal of the third transistor element in each of the pixel circuits arranged in the same row is connected to one of scan signal lines in common,

a control terminal of the second transistor element in each of the pixel circuits arranged in the same row or the same column is connected to one of first control lines in common,

the other end of the first capacitive element in each of the pixel circuits arranged in the same row or the same column is connected to one of second control lines in common,

a data signal line drive circuit for driving the data signal lines individually, a control line drive circuit for driving the first and second control lines individually, and a scan line drive circuit for driving the scan signal lines individually are provided,

the internal node of each of the pixel circuits in the pixel circuit array holds one voltage state among a plurality of discrete voltage states, in which multi-gradation is implemented by the different voltage states,

at a time of a self-refreshing action for compensating voltage fluctuations of the internal nodes at the same time by activating the second switch circuits and the control circuits in the plurality of the pixel circuits while sequentially changing a target gradation to be subjected to the self-refreshing action,

the scan signal line drive circuit applies a predetermined voltage to the scan signal lines connected to all of the pixel circuits in the pixel circuit array to turn off the third transistor elements,

the data signal line drive circuit applies a refreshing input voltage to the data signal lines, the refreshing input voltage being provided by adding a predetermined first adjusting voltage corresponding to a voltage drop in the second switch circuit, to a refreshing desired voltage corresponding to the voltage state of the target gradation to be subjected to a refreshing action,

the control line drive circuit applies a refreshing reference voltage to the first control lines, the refreshing reference voltage being provided by adding a predetermined second adjusting voltage corresponding to a voltage drop in the first control lines and the internal node, to a refreshing isolation voltage defined by a middle voltage between a voltage state of a gradation one step lower than the target gradation and the voltage state of the target gradation, and applies a boost voltage having a predetermined amplitude to the second control lines so as to apply a voltage change due to capacitive coupling through the first capacitive element to the output node, so that, when the voltage state of the internal node is higher than the refreshing desired voltage, the diode element is reversely biased from each of the data signal lines to the internal node, and each of the data signal lines and the internal node are not connected, when the voltage state of the internal node is lower than the refreshing isolation voltage, a potential fluctuation of the output node due to application of the boost voltage is suppressed, the first transistor element is turned off, and each of the data signal lines and the internal node are not connected, and when the voltage state of the internal node is not less than the refreshing isolation voltage and not more than the refreshing desired voltage, the diode element is forwardly biased from each of the data signal lines to the internal node, the potential fluctuation of the output node is not suppressed, the first transistor element is turned on, and the refreshing desired voltage is applied to the internal node, so that the refreshing action is executed for the pixel circuit having the internal node showing the voltage state of the target gradation,

with the boost voltage continuously applied, the target gradation is set to a one step higher gradation, the refreshing reference voltage applied to the first control lines is changed, and thereafter the refreshing input voltage applied to the data signal lines is changed, so that the refreshing action is sequentially executed for the pixel circuits having the internal nodes showing voltage states of different gradations, and

after the refreshing action is performed for all of the gradations except for a lowest gradation, the control line drive circuit reduces the voltage applied to the first control lines to turn off the second transistor elements in all of the gradations, the application of the boost voltage to the second control lines is stopped, and then the voltage applied to the first control lines is increased to turn on the second transistor elements in all of the gradations.

Here, it is preferable that the refreshing input voltage be set to a voltage value provided by further adding a predetermined extra voltage provided based on the potential fluctuations of the internal node and the output node caused when the voltages applied to the first control lines and the second control lines are fluctuated, due to parasitic capacitance of the second transistor element.

The display device according to the present invention has another characteristic that

the other end of the second switch circuit in each of the pixel circuits arranged in the same column is connected to one of voltage supply lines in common instead of being connected to one of the data signal lines in common,

each of the voltage supply lines is individually driven by the control line drive circuit, and

at the time of the self-refreshing action, the refreshing input voltage is applied from the control line drive circuit to the voltage supply lines instead of being applied from the data signal line drive circuit to the data signal lines.

The second switch circuit of each of the pixel circuits may have a series circuit of the first transistor element, the diode element, and a fourth transistor element having a control terminal connected to one of the second control lines.

The second switch circuit of each of the pixel circuits may have a series circuit of the first transistor element, the diode element, and a fourth transistor element,

a control terminal of the fourth transistor element in each of the pixel circuits arranged in the same row or the same column may be connected to one of third control lines in common, and the third control lines may be individually driven by the control line drive circuit, and

at the time of the self-refreshing action, the control line drive circuit may apply the boost voltage to the second control lines, while applying a predetermined voltage to turn on the fourth transistor element, to the third control lines.

The second switch circuit of each of the pixel circuits may have a series circuit of the first transistor element, the diode element, and a fourth transistor element,

at the time of the self-refreshing action, the control line drive circuit may apply a predetermined voltage to turn on the fourth transistor element, to the third control lines, while applying the boost voltage to the second control lines.

In addition, in the above respective configurations, the diode element may include a MOS transistor in which a gate and a source are connected to each other.

Effect of the Invention

According to the configuration of the present invention, in addition to the normal writing action, the action (self-refreshing action) to restore the absolute value of the voltage between both ends of the display element part, to a value at the time of the last writing action can be performed without depending on the writing action. Especially, according to the present invention, only the pixel circuit having the internal node to be recovered to the voltage state of the target gradation can be automatically refreshed among the pixel circuits, by applying the pulse voltage one time, so that the self-refreshing action can be performed in the circumstance where the multivalued voltage state is held in the internal node.

In the case where the plurality of pixel circuits are arranged, the normal writing action is executed with respect to each row in general. Thus, it is necessary to drive the driver circuit up to the number of times corresponding to the number of rows of the arranged pixel circuits.

According to the pixel circuit of the present invention, the refreshing action can be executed for the plurality of arranged pixels collectively with respect to each voltage state held therein by performing the self-refreshing action. Therefore, the number of times required to drive the driver circuit from the start to the end of the refreshing action can be considerably reduced, and power consumption can be cut.

Thus, since it is not necessary to separately provide a memory part such as a SRAM in the pixel circuit, the aperture rate is not largely lowered unlike the conventional technique.

Especially, according to the configuration of the present invention, at the time of self-refreshing action, with the second transistor element turned off once, the application of the boost voltage to the second control line is stopped on the assumption that the potential fluctuation of the internal node is generated due to the parasitic capacitance of the transistor when the voltages applied to the first control line and the second control line are fluctuated. In this case, the potentials of the internal node and the output node in the pixel circuit in each gradation are previously reduced a little, and then the voltage applied to the first control line is increased, so that the potentials of both of the nodes become equal to each other. Thus, even when the self-refreshing action is repeatedly executed, it is possible to prevent the internal node from being set at the voltage higher than the refreshing desired voltage due to the parasitic capacitance after the refreshing action.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram showing one example of a schematic configuration of a display device of the present invention.

FIG. 2 is a partial cross-sectional schematic structure diagram of a liquid crystal display device.

FIG. 3 is a block diagram showing one example of a schematic configuration of a display device of the present invention.

FIG. 4 is a circuit diagram showing a basic circuit configuration of a pixel circuit of the present invention.

FIG. 5 is a circuit diagram showing another basic circuit configuration of a pixel circuit of the present invention.

FIG. 6 is a circuit diagram showing another basic circuit configuration of a pixel circuit of the present invention.

FIG. 7 is a circuit diagram showing a first type circuit configuration example, among the pixel circuits of the present invention.

FIG. 8 is a circuit diagram showing another first type circuit configuration example, among the pixel circuits of the present invention.

FIG. 9 is a circuit diagram showing a second type circuit configuration example, among the pixel circuits of the present invention.

FIG. 10 is a circuit diagram showing a second type circuit configuration example, among the pixel circuits of the present invention.

FIG. 11 is a circuit diagram showing a second type circuit configuration example, among the pixel circuits of the present invention.

FIG. 12 is a circuit diagram showing a second type circuit configuration example, among the pixel circuits of the present invention.

FIG. 13 is a circuit diagram showing a second type circuit configuration example, among the pixel circuits of the present invention.

FIG. 14 is a circuit diagram showing a second type circuit configuration example, among the pixel circuits of the present invention.

FIG. 15 is a circuit diagram showing a second type circuit configuration example, among the pixel circuits of the present invention.

FIG. 16 is a circuit diagram showing a third type circuit configuration example, among the pixel circuits of the present invention.

FIG. 17 is a circuit diagram showing a third type circuit configuration example, among the pixel circuits of the present invention.

FIG. 18 is a timing chart of a self-refreshing action according to a second embodiment in the first and third type pixel circuits.

FIG. 19 is another timing chart of a self-refreshing action according to the second embodiment in the first and third type pixel circuits.

FIG. 20 is another timing chart of a self-refreshing action according to the second embodiment in the first and third type pixel circuits.

FIG. 21 is a timing chart of a self-refreshing action according to the second embodiment in the second type pixel circuit.

FIG. 22 is another timing chart of a self-refreshing action according to the second embodiment in the second type pixel circuit.

FIG. 23 is a timing chart of a self-refreshing action according to a third embodiment in the first type pixel circuit.

FIG. 24 is a timing chart of a self-refreshing action according to the third embodiment in the second type pixel circuit.

FIG. 25 is another timing chart of a self-refreshing action according to the third embodiment in the second type pixel circuit.

FIG. 26 is another timing chart of a self-refreshing action according to the third embodiment in the first type pixel circuit.

FIG. 27 is a timing chart of a self-refreshing action according to the fourth embodiment in the first type pixel circuit.

FIG. 28 is a timing chart of a writing action in the constant display mode in the first type pixel circuit.

FIG. 29 is a timing chart of a writing action in the constant display mode in the second type pixel circuit.

FIG. 30 is a timing chart of a writing action in the constant display mode in the second type pixel circuit.

FIG. 31 is a timing chart of a writing action in the constant display mode in the third type pixel circuit.

FIG. 32 is a flowchart showing an execution procedure of the writing action and the self-refreshing action in the constant display mode.

FIG. 33 is one example of a timing chart of a writing action in a normal display mode in the first type pixel circuit.

FIG. 34 is one example of a timing chart of a writing action in a normal display mode in the second type pixel circuit.

FIG. 35 is a circuit diagram showing still another basic circuit configuration of a pixel circuit in the present invention.

FIG. 36 is a circuit diagram showing still another basic circuit configuration of a pixel circuit in the present invention.

FIG. 37 is a circuit diagram showing still another configuration of a pixel circuit in the present invention.

FIG. 38 is an equivalent circuit diagram of a pixel circuit of a general active matrix type liquid crystal display device.

FIG. 39 is a block diagram showing a circuit arrangement example of an active matrix type liquid crystal display device having m×n pixels.

MODE FOR CARRYING OUT THE INVENTION

Hereinafter, a description will be given of each embodiment of a pixel circuit and a display device of the present invention with reference to the drawings. In addition, the same components as those in FIGS. 38 and 39 are marked with the same references.

First Embodiment

In a first embodiment, a description will be given of configurations of the display device of the present invention (hereinafter, simply referred to as the “display device”) and the pixel circuit constituting the display device.

<<Display Device>>

FIG. 1 shows a schematic configuration of a display device 1. The display device 1 includes an active matrix substrate 10, an opposite electrode 80, a display control circuit 11, an opposite electrode drive circuit 12, a source driver 13, a gate driver 14, and various signal lines which will be described below. On the active matrix substrate 10, a plurality of pixel circuits 2 are arranged in raw and column directions, respectively, and a pixel circuit array is formed.

In addition, the pixel circuit 2 is shown as a block in FIG. 1 so as to prevent the drawing from becoming complicated. Moreover, for descriptive purposes, the active matrix substrate 10 is shown above the opposite electrode 80 so as to make it clear that the various signal lines are formed on the active matrix substrate 10.

According to this embodiment, the display device 1 can make a screen display in two display modes such as a normal display mode and a constant display mode with the same pixel circuit 2. In the normal display mode, a moving image or a still image is displayed in full color and a transmissive liquid crystal display is made with a backlight. Meanwhile, in the constant display mode in this embodiment, three or more gradations are displayed by a pixel circuit unit, and the three adjacent pixel circuits 2 are allocated to three primary colors (R, G, B), respectively. For example, in a case where the number of the gradations is 3, 27 colors are displayed, and in a case where the number of the gradations is 4, 64 colors are displayed. However, the assumed number of the gradations is smaller than that of the normal display mode.

In addition, in the constant display mode, the number of display colors can be increased by an area coverage modulation by further combining a plurality of sets of the three adjacent pixel circuits. Moreover, the constant display mode in this embodiment can be used in the transmissive liquid crystal display and a reflective liquid crystal display.

In the following description, for descriptive purposes, a minimum display unit corresponding to the one pixel circuit 2 is referred to as the “pixel”, and “pixel data” to be written in each pixel circuit is gradation data of each color, in a case of a color display with the three primary colors (R, B, G). In a case of a color display which includes brightness data of the plurality of gradations, in addition to the primary colors, the brightness data is also included in the pixel date.

FIG. 2 is a schematic cross-sectional structure view showing a relationship between the active matrix substrate 10 and the opposite electrode 80, and shows a structure of a display element part 21 (refer to FIG. 4) serving as a component of the pixel circuit 2. The active matrix substrate 10 is a light transmissive transparent substrate composed of glass or plastic.

As shown in FIG. 1, the pixel circuit 2 each including the signal lines are formed on the active matrix substrate 10. In FIG. 2, a pixel electrode 20 is shown as a representative of the component of the pixel circuit 2. The pixel electrode 20 is composed of a light transmissive transparent conductive material such as ITO (indium tin oxide).

A light transmissive opposite substrate 81 is arranged so as to be opposed to the active matrix substrate 10, and a liquid crystal layer 75 is held in a gap between the substrates. A polarization plate (not shown) is attached to an outer surface of each substrate.

The liquid crystal layer 75 is sealed with a sealing material 74, in a surrounding area of both substrates. On the opposite substrate 81, the opposite electrode 80 composed of the light transmissive transparent conductive material such as ITO is formed so as to be opposed to the pixel electrode 20. This opposite electrode 80 is formed as a single film so as to spread nearly all over the opposite substrate 81. Here, a unit liquid crystal display element Clc (refer to FIG. 4) is composed of the one pixel electrode 20, the opposite electrode 80, and the liquid crystal layer 75 held therebetween.

Furthermore, a backlight device (not shown) is arranged on a back surface side of the active matrix substrate 10, and can emit light in a direction from the active matrix substrate 10 toward the opposite substrate 81.

As shown in FIG. 1, the signal lines are formed on the active matrix substrate 10 in horizontal and vertical directions. Thus, the pixel circuits 2 are formed, in the shape of a matrix, at intersecting points of m source lines (SL1, SL2, . . . , SLm) extending in the vertical direction (column direction), and n gate lines (GL1, GL2, . . . , GLn) extending in the horizontal direction (row direction). Each of the numbers m and n is two or more natural number. In addition, the source lines are represented by the “source line SL”, and the gate lines are represented by the “gate line GL”.

Here, the source line SL corresponds to a “data signal line”, and the gate line GL corresponds to a “scanning signal line”. In addition, the source driver 13 corresponds to a “data signal line drive circuit”, the gate driver 14 corresponds to a “scanning signal line drive circuit”, the opposite electrode drive circuit 12 corresponds to an “opposite electrode voltage supply circuit”, and the display control circuit 11 partially corresponds to a “control line drive circuit”.

In addition, in FIG. 1, each of the display control circuit 11 and the opposite electrode drive circuit 12 is illustrated so as to exist independently from the source driver 13 and the gate driver 14, but the display control circuit 11 and the opposite electrode drive circuit 12 may be included in these drivers.

According to this embodiment, a reference line REF, an auxiliary capacity line CSL, and a boost line BST are provided as the signal lines to drive the pixel circuit 2, as well as the source line SL and the gate line GL described above. In addition, as another configuration example, a selection line SEL can be further provided. FIG. 3 shows a configuration of the display device in this case.

The reference line REF, the boost line BST, and the selection line SEL correspond to a “first control line”, a “second control line”, and a “third control line”, respectively, and are driven by the display control circuit 11. The auxiliary capacity line CSL corresponds to a “fourth control line” or a “fixed voltage line” and is driven by the display control circuit 11, as one example.

Referring to FIGS. 1 to 3, each of the reference line REF, the boost line BST, and the auxiliary capacity line CSL is provided in each row so as to extend in a row direction, and wirings of each row are mutually connected and unified in a periphery part of the pixel circuit array, but the wiring in each row may be individually driven and a common voltage may be applied thereto according to an operation mode, or each line may be provided in each column so as to extend in a column direction. Basically, each of the reference line REF, the boost line BST, and the auxiliary capacity line CSL is configured to be shared by the plurality of pixel circuits 2. In addition, in the case where the selection line SEL is further provided, it may be provided in the same manner as that of the boost line BST.

The display control circuit 11 controls a writing action in the normal display mode and the constant display mode, and a self-refreshing action in the constant display mode as will be described below.

At the time of the writing action, the display control circuit 11 receives a data signal Dv and a timing signal Ct representing an image to be displayed, from an external signal source, and based on the signals Dv and Ct, generates a digital image signal DA and a data side timing control signal Stc to be applied to the source driver 13, a scan side timing control signal Gtc to be applied to the gate driver 14, and an opposite voltage control signal Sec to be applied to the opposite electrode drive circuit 12 as signals to display the image on the display element part 21 (refer to FIG. 4) of the pixel circuit array, and signal voltages to be applied to the reference line REF, the boost line BST, the auxiliary capacity line CSL, and the selection SEL (in the case it is provided).

The source driver 13 is controlled by the display control circuit 11 so as to apply a source signal having a predetermined voltage amplitude to each source line SL at predetermined timing at the time of the writing action and the self-refreshing action.

At the time of the writing action, the source driver 13 generates a voltage appropriate for a voltage level of an opposite voltage Vcom which corresponds to a pixel value for one display line represented by the digital signal DA, as source signals Sc1, Sc2, . . . , Scm with respect to each horizontal period (also referred to as the “H period”), based on the digital image signal DA and the data side timing control signal Stc. As this voltage, a multi-gradation voltage is assumed in both of the normal display mode and the constant display mode, but the gradation number in the constant display mode is smaller than that in the normal display mode in this embodiment, and the voltage is a three-gradation (three-valued) voltage. Thus, these source signals are applied to the corresponding source lines SL1, SL2, . . . , SLm, respectively.

In addition, at the time of the self-refreshing action, the source driver 13 is controlled by the display control circuit 11 so as to apply the same voltage to all the source lines SL connected to the target pixel circuits 2, at the same timing (detail will be described below).

The gate driver 14 is controlled by the display control circuit 11 so as to apply a gate signal having a predetermined voltage amplitude to each gate line GL at predetermined timing at the time of the writing action and the self-refreshing action. In addition, the gate driver 14 may be formed on the active matrix substrate 10 like the pixel circuit 2.

At the time of the writing action, the gate driver 14 sequentially selects the gate lines GL1, GL2, . . . , GLn for roughly each horizontal period, in each frame period of the digital image signal DA, in order to write the source signals Sc1, Sc2, . . . , Scm in each pixel circuit 2, based on the scan side timing control signal Gtc.

In addition, at the time of the self-refreshing action, the gate driver 14 is controlled by the display control circuit 11 so as to apply the same voltage to all the gate lines GL connected to the target pixel circuits 2, at the same timing (detail will be described below).

The opposite electrode drive circuit 12 applies the opposite voltage Vcom to the opposite electrode 80 through an opposite electrode wiring CML. According to this embodiment, the opposite electrode drive circuit 12 outputs the opposite voltage Vcom so that it is alternately switched between a predetermined high level (5 V) and a predetermined low level (0 V) in the normal display mode and the constant display mode. Thus, the action to drive the opposite electrode 80 while switching the voltage between the high level and the low level is referred to as the “opposite AC driving”.

According to the “opposite AC driving” in the normal display mode, the opposite voltage Vcom is switched between the high level and the low level with respect to each horizontal period and each frame period. That is, in a certain frame period, a voltage polarity between the opposite electrode 80 and the pixel electrode 20 is changed between the two adjacent horizontal periods. In addition, in the same horizontal period, the voltage polarity between the opposite electrode 80 and the pixel electrode 20 is changed between the two adjacent frame periods.

Meanwhile, in the constant display mode, the same voltage level is maintained in the one frame period, and the voltage polarity between the opposite electrode 80 and the pixel electrode 20 is changed between the two adjacent writing actions.

When the voltage having the same polarity is continuously applied to between the opposite electrode 80 and the pixel circuit 20, burn-in of the display screen (surface burn-in) is caused, so that a polarity inverting action is needed, and when the “opposite AC driving” is employed, a voltage amplitude to be applied to the pixel electrode 20 can be reduced in the polarity inverting action.

<<Pixel Circuit>>

Next, a configuration of the pixel circuit 2 will be described with reference to FIGS. 4 to 17. FIGS. 4 to 6 show basic circuit configurations of the pixel circuits 2 of the present invention. The pixel circuit 2 includes the display element part 21 including the unit liquid crystal display element Clc, a first switch circuit 22, a second switch circuit 23, a control circuit 24, and an auxiliary capacitive element Cs, in common with all circuit configurations. The auxiliary capacitive element Cs corresponds to a “second capacitive element”.

In addition, the basic circuit configurations shown in FIGS. 4, 5, and 6 show common circuit configurations including basic circuit configurations belonging to first to third types which will be described below. Since the unit liquid crystal display element Clc has been already described with reference to FIG. 2, its description is omitted.

The pixel electrode 20 is connected to one ends of the first switch circuit 22, the second switch circuit 23, and the control circuit 24, whereby an internal node N1 is formed. The internal node N1 holds a voltage of the pixel data supplied from the source line SL at the time of the writing action.

The auxiliary capacitive element Cs has one end connected to the internal node N1, and the other end connected to the auxiliary capacity line CSL. This auxiliary capacitive element Cs is additionally provided so that the internal node N1 can stably hold the voltage of the pixel data.

One end of the first switch circuit 22, which does not compose the internal node N1, is connected to the source line SL. The first switch circuit 22 has a transistor T3 functioning as a switch element. The transistor T3 is a transistor whose control terminal is connected to the gate line, and corresponds to a “third transistor element”. The first switch circuit 22 is turned off and the source line SL and the internal node N1 are not connected when at least the transistor T3 is off.

One end of the second switch circuit 23, which does not compose the internal node N1, is connected to the source line SL. The second switch circuit 23 is a series circuit composed of a transistor T1 and a diode D1. In addition, the transistor T1 is a transistor whose control terminal is connected to an output node N2 of the control circuit 24, and corresponds to a “first transistor element”. In addition, the diode D1 performs a rectifying action in a direction from the source line SL to the internal node N1, and corresponds to a “diode element”. The diode D1 is formed with a PN junction in this embodiment, but it may be formed with a schottky junction or diode connection of a MOSFET (MOSFET in which a drain or source is connected to a gate).

Hereinafter, as shown in FIG. 4, a configuration in which the second switch circuit 23 is the series circuit composed of the transistor T1 and the diode D1, and a transistor T4 is not included is referred to as a first type.

Unlike the first type, as shown in FIGS. 5 and 6, the second switch circuit 23 may be a series circuit including the transistor T4 in addition to the transistor T1 and the diode D1. At this time, two types are provided in FIGS. 5 and 6 respectively, depending on the signal line to which the control terminal of the transistor T4 is connected. According to the type (second type) of the pixel circuit shown in FIG. 5, the selection line SEL is additionally provided in addition to the boost line BST, and a control terminal of the transistor T4 is connected to the selection line SEL. Meanwhile, according to the type (third type) of the pixel circuit shown in FIG. 6, the control terminal of the transistor T4 is connected to the boost line BST. In addition, the selection line SEL does not exist in the first type as a matter of course. The transistor T4 corresponds to a “fourth transistor element”.

In the case of the first type, when the transistor T1 is on, and a potential difference more than a turn-on voltage is generated between both ends of the diode D1, the second switch circuit 23 is turned on in a direction from the source line SL to the internal node N1. Meanwhile, in the case of the second and third types, when both of the transistors T1 and T4 are on, and the potential difference more than the turn-on voltage is generated between both ends of the diode D1, the second switch circuit 23 is turned on in the direction from the source line SL to the internal node N1.

The control circuit 24 is a series circuit composed of a transistor T2 and a boost capacitive element Cbst. A first terminal of the transistor T2 is connected to the internal node N1, and a control terminal thereof is connected to the reference line REF. In addition, a second terminal of the transistor T2 is connected to a first terminal of the boost capacitive element Cbst and the control terminal of the transistor T1, whereby the output node N2 is formed. A second terminal of the boost capacitive element Cbst is connected to the boost line BST. The transistor T2 corresponds to a “second transistor element”.

Meanwhile, one end of the auxiliary capacitive element Cs, and one end of the liquid crystal capacitive element Clc are connected to the internal node N1. In order to prevent the references from becoming complicated, electrostatic capacity of the auxiliary capacitive element (referred to as the “auxiliary capacity”) is represented by Cs, and electrostatic capacity of the liquid crystal capacitive element (referred to as the “liquid crystal capacity”) is represented by Cls. At this time, total capacity which is parasitic in the internal node N1, that is, pixel capacity Cp in which the pixel data is written and held is roughly expressed by a sum of the liquid crystal capacity Clc and the auxiliary capacity Cs (Cp≅Clc+Cs).

At this time, the boost capacitive element Cbst is set such that Cbst<<Cp is established wherein Cbst represents electrostatic capacity of this element (referred to as the “boost capacity”).

When the transistor T2 is on, the output node N2 holds the voltage according to the voltage level of the internal node N1, but when the transistor T2 is off, it maintains an original holding voltage even when the voltage level of the internal node N1 changes. This holding voltage of the output node N2 controls on/off of the transistor T1 of the second switch circuit 23.

Each of the four kinds of transistors T1 to T4 is a thin film transistor such as a polycrystalline silicon TFT or an amorphous silicon TFT which is formed on the active matrix substrate 10, and one of the first and second terminals corresponds to a drain electrode, and the other thereof corresponds to a source electrode, and the control terminal corresponds to a gate electrode. In addition, each of the transistors T1 to T4 may be composed of a single transistor element, but in a case where a leak current is highly required to be suppressed, it may be configured such that the several transistors are connected in series and their control terminals are connected to one another. In the following description about the operation of the pixel circuit 2, it is assumed that the each of the transistors T1 to T4 is an N-channel type polycrystalline silicon TFT, and its threshold voltage is about 2 V.

In addition, similar to the transistors T1 to T4, the diode D1 is also formed on the active matrix substrate 10. According to this embodiment, the diode D1 is provided as the PN junction composed of polycrystalline silicon.

First, a description will be given of the pixel circuit belonging to the first type, in which the second switch circuit 23 is the series circuit composed of the transistor T1 and the diode D1.

At this time, as described above, pixel circuits 2A shown in FIGS. 7 and 8 are assumed, depending on the configuration of the first switch circuit 22.

The first type pixel circuit 2A shown in FIG. 7 has the first switch circuit 22 only composed of the transistor T3.

Here, FIG. 7 shows a configuration example in which the second switch circuit 23 is the series circuit composed of the diode D1 and the transistor T1, the first terminal of the transistor T1 is connected to the internal node N1, the second terminal of the transistor T1 is connected to a cathode terminal of the diode D1, and an anode terminal of the diode D1 is connected to the source line SL, as one example. However, as shown in FIG. 8, the positions of the transistor T1 and the diode D1 may be exchanged in the series circuit. In addition, as another circuit configuration, the transistor T1 may be sandwiched between the two diodes D1.

Next, a description will be given of the pixel circuit belonging to the second type, in which the second switch circuit 23 is the series circuit composed of the transistor T1, the diode D1, and the transistor T4, and the control terminal of the transistor T4 is connected to the selection line SEL.

In the second type, pixel circuits 2B shown in FIGS. 9 to 11 and pixel circuits 2C shown in FIGS. 12 to 15 are assumed, depending on the configuration of the first switch circuit 22.

According to the pixel circuit 2B shown in FIG. 9, the first switch circuit 22 is only composed of the transistor T3. In addition, similar to the first type, as for the configuration of the second switch circuit 23, variation circuits can be implemented depending on the arrangement of the diode D1 (refer to FIGS. 10 and 11). In addition, the positions of the transistors T1 and T4 may be exchanged in the circuits.

The pixel circuit 2C shown in FIG. 12 has the first switch circuit 22 which is the series circuit composed of the transistor T3 and the transistor T4. A variation circuit is implemented as shown in FIG. 13 by changing the arranged position of the transistor T4. In addition, a variation circuit can be implemented as shown in FIG. 14 by providing the plurality of transistors T4.

Furthermore, as shown in FIG. 15, instead of the transistor T4 in the first switch circuit 22, a variation circuit can be implemented such that a transistor T5 is connected to the transistor T4 through their control terminals.

Next, a description will be given of the pixel circuit belonging to the third type in which the second switch circuit 23 is the series circuit composed of the transistor T1, the diode D1, and the transistor T4, and the control terminal of the transistor T4 is connected to the boost line BST.

The third type pixel circuit has a configuration in which the control terminal of the transistor T4 is connected to the boost line BST, and the selection SEL is not provided, compared to the second type pixel circuit. Therefore, the pixel circuits corresponding to the pixel circuits 2B shown in FIGS. 9 to 11, and the pixel circuits 2C shown in FIGS. 12 to 15 can be realized. As one example, a pixel circuit 2D corresponding to the pixel circuit 2B shown in FIG. 9 is shown in FIG. 16, and a pixel circuit 2E corresponding to the pixel circuit 2C shown in FIG. 12 is shown in FIG. 17.

In addition, in the above type of pixel circuits, the same transistor elements or diode elements may be connected in series, respectively.

Second Embodiment

In a second embodiment, a description will be given of a self-refreshing action in each of the first to third type pixel circuits with reference to the drawings.

The self-refreshing action means an action in the constant display mode performed for the plurality of the pixel circuits 2 such that the first switch circuits 22, the second switch circuits 23, and the control circuits 24 are activated in a predetermined sequence, and the potentials of the pixel electrodes 20 (this is also the potentials of the internal nodes N1) are restored to a potential of the gradation written in the last writing action, and for the pixels of all gradations, the pixel circuits are collectively recovered at the same time with respect to each gradation. The self-refreshing action is a specific action by the pixel circuits 2A to 2E in the present invention, and power consumption can be considerably reduced, compared to the conventional “external refreshing action” in which the potential of the pixel electrode 20 is restored by performing the normal writing action. In addition, the above term “at the same time” in “collectively at the same time” means “the same time” having a time width of a series of actions in the self-refreshing action.

Meanwhile, in the conventional case, an action to invert the polarity only of a liquid crystal voltage Vcl applied to between the pixel electrode 20 and the opposite electrode 80 was executed while maintaining its absolute value (external polarity inverting action) by performing the writing action. When this external polarity inverting action is performed, the polarity is inverted and the absolute value of the liquid crystal voltage Vcl is updated to a state at the time of the last writing. That is, the polarity inverting and the refreshing action are performed at the same time. Therefore, it is not normally performed to execute the refreshing action with a view to only updating the absolute value of the liquid crystal voltage Vcl without inverting the polarity, but hereinafter, such refreshing action is referred to as the “external refreshing action” with a view to comparing it with the self-refreshing action, for convenience in description.

In addition, in the case where the refreshing action is executed by the external polarity inverting action, the writing action is still performed. That is, also when compared to this conventional method, the power consumption is considerably reduced by the self-refreshing action in this embodiment.

As will be described below, according to the self-refreshing action in this embodiment, all of the pixel circuits are set to the same voltage state, but actually, under this voltage state, the pixel circuit in which the internal node N1 shows the voltage state of specific one gradation is only automatically selected, and the potential of the internal node N1 is restored (refreshed). That is, although the voltage is applied to all the pixel circuits, the potential of the internal node N1 is refreshed in some pixel circuits, and it is not refreshed in the other pixel circuits, at the time of the voltage application, in practice.

Therefore, in order to avoid confusion in description, the term “self-refreshing (action)” and the term “refreshing (action)” are to be intentionally distinguished in the following description. The former is used in a wide concept referring to a series of actions to restore the potential of the internal node N1 of each pixel circuit. Meanwhile, the latter is used in a narrow concept referring to an action to actually restore the potential (potential of the internal node) of the pixel electrode. That is, according to the “self-refreshing action” in this embodiment, only the internal node showing the voltage state of the specific one gradation is automatically and selectively “refreshed” by setting the same voltage state for all the pixel circuits. Thus, the value of the voltage is changed so as to change the gradation as the “refreshing” target, and the voltage is similarly applied, so that “refreshing” is performed for all gradations. Thus, according to the “self-refreshing action” in this embodiment, the “refreshing action” is performed with respect to each gradation.

The voltage is applied to all the gate lines GL, the source lines SL, the reference lines REF, the auxiliary capacity lines CSL, and the boost lines BST which are connected to the pixel circuit 2 serving as the target of the self-refreshing action, and to the opposite electrode 80 at the same timing. In the case of the second type pixel circuit having the selection line SEL, the voltage is similarly applied to the selection line SEL.

Thus, under the same timing, the same voltage is applied to all the gate lines GL, the same voltage is applied to all the reference lines REF, the same voltage is applied to all the auxiliary capacity lines CSL, and the same voltage is applied to all the boost lines BST. The timing control of the voltage application is performed by the display control circuit 11 shown in FIG. 1, and individual voltage application is performed by the display control circuit 11, the opposite electrode drive circuit 12, the source driver 13, and the gate driver 14.

In the constant display mode in this embodiment, as described in the first embodiment, it is also assumed that the three-gradation (three-valued) pixel data is held in the pixel circuit unit. At this time, the potential VN1 (this is also the potential of the pixel electrode 20) held in the internal node N1 shows three voltage states such as first to third voltage states. According to this embodiment, as one example, the first voltage state (high voltage state) is set to 5 V, the second voltage state (middle voltage state) is set to 3 V, and the third voltage state (low voltage state) is set to 0 V.

In the state just before the execution of the self-refreshing action, it is assumed that there are the pixel in which the pixel electrode 20 is written in the first voltage state, the pixel in which it is written in the second voltage state, and the pixel in which it is written in the third voltage state. However, according to the self-refreshing action in this embodiment, the voltage is applied based on the same sequence regardless of the voltage state of the pixel electrode 20, so that the refreshing action can be executed for all the pixel circuits. These contents will be described with reference to a timing chart and a circuit diagram.

In addition, hereinafter, a case where the voltage is written in the first voltage state (high level voltage) in the last writing action, and the high level voltage is to be restored is referred to as the “case H”, a case where the voltage is written in the second voltage state (middle level voltage) in the last writing action, and the middle level voltage is to be restored is referred to as the “case M”, and a case where the voltage is written in the third voltage state (low level voltage) in the last writing action, and the low level voltage is to be restored is referred to as the “case L”.

In addition, as described in the first embodiment, it is assumed that the threshold voltage of each transistor is 2 V. In addition, it is assumed that the turn-on voltage of the diode D1 is 0.6 V.

First, a description will be given of the self-refreshing action for the first type pixel circuit 2A in which the second switch circuit 23 is the series circuit composed of the transistor T1 and the diode D1 only. Here, the pixel circuit 2A shown in FIG. 7 is assumed.

FIG. 18 shows a timing chart of the first type self-refreshing action. As shown in FIG. 18, the self-refreshing action is divided into two steps S1 and S2, and the step S1 is provided with two phases P1 and P2. FIG. 18 illustrates voltage waveforms of all the gate lines GL, the source lines SL, the boost lines BST, the reference lines REF, the auxiliary capacity lines CSL, and the boost lines BST which are connected to the pixel circuits 2A serving as the target of the self-refreshing action, and a voltage waveform of the opposite voltage Vcom. In addition, according to this embodiment, it is assumed that all the pixel circuits of the pixel circuit array are the target of the self-refreshing action.

In addition, FIG. 18 shows waveforms showing changes of the potential (pixel voltage) VN1 of the internal node N1, and the potential VN2 of the output node N2 in the each of the cases, H, M, and L, and on/off states of the transistors T1 to T3 in each step and each phase. Furthermore, FIG. 18 shows the case in the parentheses. For example, VN1 (H) is a waveform showing the change of the potential VN1 in the case H.

In addition, it is assumed that the high level has been written in the case H, the middle level has been written in the case M, and the low level has been written in the case L at a point before a time (t1) to start the self-refreshing action.

After the writing action has been executed and the time has elapsed, the potential VN1 of the internal node N1 changes due to generation of a leak current of each transistor in the pixel circuit. In the case H, the VN1 is 5 V just after the writing action, but this value becomes lower than the original value after the time has elapsed. Similarly, in the case M, the VN1 is 3 V just after the writing action, but this value becomes lower than the original value after the time has elapsed. In each of these cases H and M, the potential of the internal node N1 gradually decreases with time mainly because a leak current flows toward a lower potential (such as the ground line) through the off-state transistor.

In addition, in the case L, the potential VN1 is 0 V just after the writing action, it could rise a little with time. This is because when the writing voltage is applied to the source line SL at the time of the writing action in another pixel circuit, a leak current flows from the source line SL to the internal node N1 through the off-state transistor even in the unselected pixel circuit.

FIG. 18 shows that the VN1(H) is a little lower than 5 V, the VN1(M) is a little lower than 3 V, and the VN1(L) is a little higher than 0 V, at the time t1. This is because the above potential fluctuation is considered.

The self-refreshing action in this embodiment is mainly divided into the two steps S1 and S2. The step S1 corresponds to a “refreshing step”, and the step S2 corresponds to a “stand-by step”.

In the step S1, the refreshing action is directly executed for the case H and the case M by applying pulse voltages. Meanwhile, in the step S2, the refreshing action is indirectly executed for the case L by applying a constant voltage for a time longer than that of the step S1 (such as ten times or more). In addition, the term “directly executed” means that the internal node N1 and the source line SL are connected through the second switch circuit 23, so that the voltage applied to the source line SL is applied to the internal node N1, and the potential VN1 of the internal node is set to a desired value. In addition, the term “indirectly executed” means that the internal node N1 and the source line SL are not connected through the second switch circuit 23, but the potential VN1 of the internal node N1 is brought closer to a desired value by using a leak current slightly flowing between the internal node N1 and the source line SL through the off-state first switch circuit 22.

In addition, in the step S1, the phase P1 and P2 differ depending on whether the case H or M is refreshed. In FIG. 18, in the phase P1, only the internal node N1 of the case H (high voltage writing) is refreshed, and in the phase P2, only the internal node N1 of the case M (middle voltage writing) is refreshed. Hereinafter, this operation will be described in detail.

<<Step S1/Phase P1>>

In the phase P1 to be started at the time t1, a voltage which can completely turn off the transistor T3 is applied to the gate line GL. Here, the voltage is −5 V. In addition, during the execution of the self-refreshing action, the transistor T3 is constantly off, so that the voltage applied to the gate line GL may remain unchanged during the self-refreshing action.

The opposite voltage Vcom applied to the opposite electrode 80, and a voltage applied to the auxiliary capacity line CSL are set to 0 V. Here, the voltage is not limited to 0 V, and a voltage value before the time t1 may be maintained as it is. In addition, these voltages also may remain unchanged during the self-refreshing action.

At the time t1, a voltage provided by adding a turn-on voltage Vdn of the diode D1 to the desired voltage of the internal node N1 to be restored by the refreshing action is applied to the source line SL. In the phase P1, the refreshing target is the case H, so that the desired voltage of the internal node N1 is 5 V. Therefore, when the turn-on voltage Vdn of the diode D1 is 0.6 V, 5.6 V is applied to the source line SL.

In addition, the desired voltage of the internal node N1 corresponds to a “refreshing desired voltage”, the turn-on voltage Vdn of the diode D1 corresponds to a “first adjusting voltage”, and the voltage actually applied to the source line SL in the refreshing step S1 corresponds to a “refreshing input voltage”. Thus, with the above terms, it is defined that <refreshing input voltage=refreshing desired voltage+first adjusting voltage>. In the phase P1, the refreshing input voltage is 5.6 V.

At the time t1, in a case where the internal node N1 shows the voltage state (gradation) as the refreshing target or higher (high gradation), a voltage that turns off the transistor T2 is applied to the reference line REF, while in a case where it shows the voltage state (low gradation) lower than the voltage state (gradation) as the refreshing target, a voltage that turn on the transistor T2 is applied thereto. In the phase P1, the refreshing target is the case H (first voltage state), and there is no voltage state higher than this, so that in the case where the internal node N1 is in the first voltage state (case H), the voltage that turns off the transistor T2 is applied to the reference line REF, while in the case where it is in the second voltage state (case M) and the third voltage state (case L), the voltage that turns on the transistor T2 is applied thereto.

More specifically, since a threshold voltage Vt2 of the transistor T2 is 2 V, the transistor T2 in the case M can be turned on by applying the voltage higher than 5 V (=3+2) to the reference line REF. However, when the voltage higher than 7 V (=5+2) is applied to the reference line REF, the transistor T2 in the case H as the target in the phase P1 comes to be also turned on. Therefore, the voltage between 5 V and 7 V is to be applied to the reference line REF.

In addition, it is assumed that the potential of the internal node N1 falls from the voltage state written by the last writing action by a certain level just before the execution of the self-refreshing action due to the above-described leak current. That is, the potential VN1 of the internal node N1 corresponding to the case M could fall to about 2.5 V just before the execution of the self-refreshing action. In this case, when the voltage of about 5.1 V is supposedly applied to the reference line REF, the transistor T2 could be turned off in the case M also, depending on the degree of the potential fall of the internal node N1, so that the voltage is set to 6.5 V with a view to staying on the safe side.

When 6.5 V is applied to the reference line REF, the transistor T2 is turned off in the pixel circuit in which the potential VN1 of the internal node N1 is 4.5 V or more. Meanwhile, the transistor T2 is turned on in the pixel circuit in which the VN1 is lower than 4.5 V. The self-refreshing action is to be executed for the internal node N1 in the case H written to 5 V in the last writing action before it falls by 0.5 V or more due to the generation of the leak current so that the VN1 can be at 4.5 V or more, and as a result, the transistor T2 is turned off. Meanwhile, the internal node N1 in the case M written to 3 V and the internal node N1 of the case L written to 0 V by the last writing action do not become 4.5 V or more even after the time has elapsed, so that the transistor T2 is turned on in these cases.

Based on the above description, a value provided by subtracting the threshold voltage Vt2 of the transistor T2 from a voltage Vref applied to the reference line REF needs to exist between the internal node potential VN1 in the case H serving as the refreshing target in this phase, and the internal node potential VN1 in the case M in the voltage state one step lower than the above. In other words, in this phase P1, the voltage Vref applied to the reference line REF needs to satisfy the condition that 3 V<(Vref−Vt2)<5 V. The voltage of Vref−Vt2 corresponds to a “refreshing isolation voltage”, and the Vt2 corresponds to a “second adjusting voltage”, and the Vref corresponds to a “refreshing reference voltage”. When the above condition is described with these terms, the “refreshing reference voltage” to be applied to the reference line REF in the phase P1 corresponds to the voltage value provided by adding the “second adjusting voltage” corresponding to the threshold voltage of the transistor T2, to the “refreshing isolation voltage” defined as the middle voltage between the voltage state (gradation) serving as the refreshing action target, and the voltage state (gradation) one step lower than the above.

As for the boost line BST, a voltage that turns on the transistor T1 in the case H in which the transistor T2 is off as described above, and turns off the transistor T1 in the cases M and L in which the transistor T2 is on is applied thereto.

The boost line BST is connected to the one end of the boost capacitive element Cbst. Therefore, when the high level voltage is applied to the boost line BST, the potential of the other end of the boost capacitive element Cbst, that is, a potential VN2 of the output node N2 is thrust upward. Thus, hereinafter, an action to thrust the potential of the output node N2 upward by increasing the voltage to be applied to the boost line BST is referred to as the “boost upthrust”.

As described above, in the case H, the transistor T2 is off in the phase P1. Therefore, a potential fluctuation amount of the node N2 due to the boost upthrust is determined by a ratio between the boost capacity Cbst and the total capacity which is parasitic in the node N2. For example, in a case where the ratio is 0.7, when the potential of one electrode of the boost capacitive element increases by ΔVbst, the potential of the other electrode, that is, the node N2 increases by roughly 0.7 ΔVbst.

In the case H, the potential VN1 (H) of the internal node N1 shows roughly 5 V at the time t1. When a potential higher than the VN1 (H) by the threshold voltage 2 V or more is applied to the gate of the transistor T1, that is, the output node N2, the transistor T1 is turned on. According to this embodiment, it is assumed that the voltage applied to the boost line BST at the time t1 is 10 V. In this case, the potential of the output node N2 rises by 7 V. As will be described below in a fifth embodiment, since the transistor T2 is on in the writing action, the node N2 shows roughly the same potential (5 V) as that of the node N1 at the point just before the time t1. Thus, the potential of the node N2 shows about 12 V due to the boost upthrust. Therefore, the potential difference more than the threshold voltage is generated between the gate of the transistor T1 and the node N1, so that the transistor T1 is turned on.

On the other hand, in the case M and the case L in which the transistor T2 is off in the phase P1, unlike the case H, the output node N2 and the internal node N1 are electrically connected. In this case, the potential fluctuation amount of the output node N2 due to the boost upthrust is affected by the total parasitic capacitance of the internal node N1, in addition to the boost capacity Cbst and the total parasitic capacitance of the node N2.

Since the internal node N1 is connected to the one end of the auxiliary capacitive element Cs, and the one end of the liquid crystal capacitive element Clc, the total capacity Cp which is parasitic in the internal node N1 is expressed by the sum of the liquid crystal capacity Clc and the auxiliary capacity Cs as described above. Thus, the boost capacity Cbst is considerably smaller than the liquid crystal capacity Cp. Therefore, a ratio of the boost capacity to the total capacity is extremely small such as about 0.01 or less. In this case, when the potential of one electrode of the boost capacitive element increases by ΔVbst, the other electrode, that is, the potential of the output node N2 only increases by up to 0.01 ΔVbst. That is, in the case M and the case L, even when ΔVbst=10 V, the potentials VN2 (M) and VN2 (L) of the output nodes N2 hardly increase.

In the case M, the potential VN2 (M) shows almost 3 V at the point just before the time t1. In addition, in the case L, the VN2 (L) show almost 0 V at the point just before the time t1. Therefore, in both cases, even when the boost upthrust is performed at the time t1, a potential sufficient to turn on the transistor is not applied to the gate of the transistor T1. That is, unlike the case H, the transistor T1 is still off.

In addition, in the cases M and L, the potential of the output node N2 just before the time t1 is not necessary to be 3 V and 0 V, respectively, and the potential only has to be a potential which does not turn on the transistor T1 even when a fine potential fluctuation due to the pulse voltage application to the boost line BST is considered. Similarly, in the case H, the potential of the node N1 just before the time t1 is not necessarily 5 V, and the potential only has to be a potential which turns on the transistor T1 after due consideration on the potential fluctuation due to the boost upthrust under the condition that the transistor T2 is in off state.

In the case H, the transistor T1 is turned on due to the boost upthrust. In addition, 5.6 V is applied to the source line SL, so that when the potential VN1 (H) of the internal node N1 falls a little from 5 V, a potential difference more than the turn-on voltage Vdn of the diode D1 is generated between the source line SL and the internal node N1. Therefore, the diode D1 is turned on from the source line SL toward the internal node N1, and a current flows from the source line SL toward the internal node N1. Thus, the potential VN1 (H) of the internal node N1 rises. In addition, the potential continues to rise until the potential difference between the source line SL and the internal node N1 becomes equal to the turn-on voltage Vdn of the diode D1, and stops when the potential difference becomes equal to the Vdn. Here, the voltage applied to the source line SL is 5.6 V, and the turn-on voltage Vdn of the diode D1 is 0.6 V, so that the rise of the potential VN1 (H) of the internal node N1 stops at 5 V. That is, the refreshing action is executed in the case H.

Thus, as described above, in the cases M and L, since the transistor T1 is off, the source line SL and the internal node N1 are not connected. Thus, the voltage applied to the source line SL does not affect the potentials VN1 (M) and VN1 (L) of the internal node N1.

To summarize the above, the refreshing action is executed for the pixel circuit in which the potential of the internal node N1 is the refreshing isolation voltage or more and the refreshing desired voltage or less. In the phase P1, the refreshing isolation voltage is 4.5 V (=6.5−2 V), and the refreshing desired voltage is 5 V, so that the refreshing action to refresh the potential VN1 to 5 V is executed only for the pixel circuit in which the potential VN1 of the internal node N1 is 4.5 to 5 V, that is, for the case H.

In addition, after the phase P1, the voltage application to each of the source line SL, the boost line BST, and the reference line REF is once stopped. Then, the next phase P2 starts at a time t2.

<<Step S1/Phase 2>>

In the phase P2 to be started at the time t2, the case M (middle voltage writing node) is the refreshing target.

More specifically, 3.6 V is applied to the source line SL as the refreshing input voltage. This voltage 3.6 V is a value provided by adding the turn-on voltage Vdn of the diode D1 to the refreshing desired voltage (3 V) of the internal node N1 in the phase P2.

Thus, in a case where the internal node N1 shows the voltage state (case M) serving as the refreshing target or the higher voltage state (case H), a voltage that turns off the transistor T2 is applied to the reference line REF, while in a case where it shows the voltage state (case L) lower than the voltage state (case M) serving as the refreshing target, a voltage that turns on the transistor T2 is applied thereto. Considering similarly to the phase P1, when the voltage higher than 2 V is applied to the reference line REF, the transistor T2 can be turned on in the case L. However, when the voltage higher than 5 V is applied to the reference line REF, the transistor T2 in the case M comes to be also turned on. Therefore, formally, the voltage between 2 V and 5 V is to be applied to the reference line REF. However, since the voltage has to be applied with a view to staying on the safe side similar to the phase P1, 4.5 V is applied as one example here. This voltage 4.5 V corresponds to the refreshing reference voltage in the phase P2, and the voltage 2.5 V which is provided by subtracting the threshold voltage of the transistor T2 therefrom corresponds to the refreshing isolation voltage.

At this time, when the potential VN1 of the internal node N1 is the refreshing isolation voltage of 2.5 V or more, the transistor T2 is turned off. Meanwhile, the transistor T2 is turned on in the pixel circuit in which the VN1 is lower than 2.5 V. That is, in the case H written to 5 V, and the case M written to 3 V in the last writing action, the VN1 is 2.5 V or more, so that the transistor T2 is turned off. Meanwhile, in the case L written to 0 V in the last writing action, the VN1 is lower than 2.5 V, so that the transistor T2 is turned on.

As for the boost line BST, a voltage that turns on the transistor T1 in the cases H and M in which the transistor T2 is off, and a voltage that turns off the transistor T1 in the case L in which the transistor T2 is on is applied thereto. Here, the voltage is 10 V similar to the phase P1. While the transistor T1 is turned on because the potential of the output node N2 is thrust upward due to the boost upthrust in the cases H and M, the transistor T1 is not turned on in the case L because the potential VN2 (L) of the output node N2 hardly changes even when the boost upthrust is performed. This principle is similar to the phase P1, so that detailed description is omitted.

In the case H, the transistor T1 is turned on due to the boost upthrust. However, 3.6 V is applied to the source line SL. Even when the potential VN1 (H) of the internal node N1 falls a little from 5 V, the fall amount is less than 1 V. Thus, a reversely-biased state is provided from the source line SL toward the internal node N1, so that the source line SL and the internal node N1 are not connected due to a rectifying action of the diode D1. That is, the potential VN1 (H) of the internal node N1 is not affected by the voltage applied to the source lines SL.

Also in the case M, the transistor T1 is turned on due to the boost upthrust. Since the voltage 3.6 V is applied to the source line SL, in the case where the potential VN1 (M) of the internal node N1 falls a little from 3 V, a potential difference more than the turn-on voltage Vdn of the diode D1 is generated between the source line SL and the internal node N1. Therefore, the diode D1 is turned on from the source line SL toward the internal node N1, and a current flows from the source line SL toward the internal node N1. Thus, the potential VN1 (M) of the internal node N1 continues to rise until the potential difference between the source line SL and the internal node N1 becomes equal to the turn-on voltage Vdn (=0.6 V). That is, the VN1 (M) reaches 3 V, and maintains the potential. Thus, the refreshing action is executed for the case M.

Thus, as described above, since the transistor T1 is off in the case L, the source line SL and the internal node N1 are not connected. Thus, the voltage applied to the source line SL does not affect the potential of the VN1 (L) of the internal node N1.

To summarize the above, in the phase P2, the refreshing isolation voltage is 2.5 V (=4.5−2 V), and the refreshing desired voltage is 3 V, so that the refreshing action to refresh the potential VN1 to 3 V is executed only for the pixel circuit in which the potential VN1 of the internal node N1 is 2.5 to 3 V, that is, for the case M.

In addition, after the phase P2, the voltage application to each of the source line SL, the boost line BST, and the reference line REF is stopped. Then, the process moves to the stand-by step S2.

<<Step S2>>

In the step S2 to be started at a time t3, a voltage that surely turns on the transistor T2 regardless of the potential VN1 of the internal node N1 is applied to the reference line REF. Here, 10 V is applied. The other signal lines maintain the same voltage states as those at the end of the phase P2.

In these voltage states, the transistor T2 is turned on, and the transistor T1 is turned off in all the cases H, M, and L. In addition, since the low level voltage is still applied to the gate line GL, the transistor T3 remains off. Thus, the potential VN1 of the internal node N1 remains the state just after the end of the refreshing step S1. In addition, the output node N2 is connected to the internal node N1, so that the VN2 is equal to the VN1.

Then, at a time t4, the voltage applied to the reference line REF is shifted to low level (0 V). Thus, the transistor T2 is turned off.

In this step S2, the same voltage states are maintained over a time which is sufficiently longer than the step S1. Since 0 V is applied to the source line SL in this period, a leak current is generated from the internal node N1 to the source line SL through the off-state transistor T3. As described above, even when the VN1 (L) is a little higher than 0 V at the time t1, the VN1 (L) is gradually brought closer to 0 V over the period of the stand-by step S2. Thus, the refreshing action is executed “indirectly” for the case L.

However, the generation of this leak current is not limited to the case L, and it is generated in the case H and the case M. Therefore, in the case H and the case M also, the VN1 is refreshed to 5 V and 3 V, respectively at the point just after the step S1, but in the step S2, the VN1 gradually falls. Therefore, it is preferable to execute the refreshing action for the cases H and M again by executing the refreshing step S1 again after the voltage state of the stand-by step S2 has lasted for a certain period of time.

As described above, the potential VN1 of the internal node N1 can be returned to the last written state in each of the cases H, M, and L by repeating the refreshing step S1 and the stand-by step S2.

Like the conventional case, in the case where the refreshing action is performed for each pixel circuit by the “writing action” through the source line SL, it is necessary to scan the gate line GL in a vertical direction one by one. Therefore, it is necessary to apply a high level voltage to the gate line GL by the number (n) of the gate lines. In addition, it is necessary to apply the same potential level as the potential level written in the last writing action to each source line SL, so that charge/discharge actions are needed for the source lines SL up to n times.

Meanwhile, according to this embodiment, the potential of the internal node N1, that is, the voltage of the pixel electrode 20 can be returned to the potential state at the time of the writing action for all the pixel circuits, regardless of the voltage state of the internal node N1, by only applying the pulse voltage in twice in the refreshing step S1, and then maintaining the constant voltage state in the subsequent stand-by step. That is, the number of times to change the voltage applied to each line to return the potential of the pixel electrode 20 of each pixel can be considerably reduced in the one frame period, and furthermore, its control contents can be simplified. Therefore, power consumption for the gate driver 14 and the source driver 13 can be considerably cut.

In addition, the self-refreshing action described with reference to FIG. 18 assumes the pixel circuit 2A in FIG. 7, but it is clear that the self-refreshing action can be executed by the same method for the variation type pixel circuit shown in FIG. 8.

In addition, in the case where the two or more diodes D1 are provided in the second switch circuit 23, the source line SL and the internal node N1 are not connected unless a potential difference provided by multiplying the turn-on voltage Vdn by the number of the diodes D1 or more exists in a direction from the source line SL to the internal node N1, in the second switch circuit 23. Therefore, in the case where the two diodes D1 are provided in the second switch circuit 23, for example, it is necessary to apply a voltage which is provided by adding a twofold value of the turn-on voltage Vdn, as the first adjusting voltage, to the refreshing desired voltage in each case, as the refreshing input voltage to the source line SL. As for the other points, the self-refreshing action can be executed by the same method as that in FIG. 18.

In addition, instead of the voltage application method shown in FIG. 18, the following method can be used.

1) In FIG. 18, the refreshing action is executed for the case H in the phase P1, and then the refreshing action is executed for the case M. This order may be reversed.

In addition, as for the order of the step S1 and the step S2, since the steps S1 and S2 are repeated, discussion about it is not meaningful.

2) The voltage 10 V is applied to the boost line BST in both the phases P1 and P2. However, the transistor T1 in the case H only has to be turned on in the phase P1, and the transistor T1 in the case M only has to be turned on in the phase P2. In the phase P2, the voltage applied to the source line SL is 3.6 V, and the threshold voltage of the transistor T3 is 2 V, so that a voltage of at least 5.6 V may be applied when the turn-on voltage Vdn of the diode D1 is not considered. That is, in the phase P2, the voltage applied to the boost line BST can be lower than that of the phase P1, to the extent that the transistor T1 in the case M is turned on.

3) In the stand-by step S2, the high level voltage (10 V) is applied to the reference line REF from the time t3 to t4. This voltage is applied to allow the potential VN2 of the output node N2 to become equal to the potential VN1 of the internal node N1. Thus, the high level voltage may be applied to the reference line REF in any timing in the period of the step S2.

4) In FIG. 18, in the refreshing step S1, after the refreshing action in the phase P1, the voltages applied to the source line SL and the reference line REF are lowered to the low level (0 V) once, and then the refreshing action is executed in the phase P2. However, the voltage applied to each line is not necessarily lowered to low level. For example, as shown in FIG. 19, in a period between the phases P1 and P2, that is, in a period while the level of the boost line BST is at low level (0 V), the voltages applied to the source line SL and the reference line REF may be set to values to be applied in the phase P2. In this case, compared to FIG. 18, a fluctuation width of the voltage applied to each of the source line SL and the reference line REF can be small.

5) In the above embodiment, as the series of self-refreshing actions, it is assumed that the refreshing action is performed for the case H and the case M in the refreshing step S1, the stand-by step S2 is performed, and then the steps are repeated. Meanwhile, as another configuration, after the refreshing action has been performed for a predetermined gradation in the refreshing step S1 in a certain term, the stand-by step S2 is performed, and then the refreshing action is performed for another gradation in the refreshing step S1 in the next term (refer to FIG. 20). In FIG. 20, the refreshing action is performed for the node N1 of the case H, in the refreshing step S1 in a term T1 (P1), the stand-by step S2 is performed, and the refreshing action is performed for the node N1 of the case M in the refreshing step S1 in the next term T2 (P2). Thus, the gradation as the target of the refreshing action may be changed with respect to each term.

Next, a description will be given of the pixel circuit belonging to the second type in which the second switch circuit 23 is the series circuit composed of the transistor T1, the diode D1, and the transistor T4, and the control terminal of the transistor T4 is connected to the selection line SEL.

First, a description will be given of a case where the self-refreshing action is executed for the second type pixel circuit 2B shown in FIG. 9. It differs from the pixel circuit 2A shown in FIG. 7 in that the conduction state of the second switch circuit 23 is controlled by the transistor T4 in addition to the transistor T1 and the diode D1.

Here, as described in the above first type, the source line SL and the internal node N1 are connected through the second switch circuit 23 only in the refreshing step S1. Thus, in the refreshing step S1, only the case serving as the target of the refreshing action is turned on by the diode D1 or the transistor T1, and in the other cases, the second switch circuit 23 is off because the diode D1 is reversely biased, or the transistor T1 is turned off. These are the same in the second type also.

In the second type, the transistor T4 is provided, and the selection line SEL to control the on/off of the transistor T4 is provided separately from the boost line BST. Therefore, totally the same voltage state as in the first type can be implemented by applying the voltage that keep the transistor T4 on, to the selection line SEL in the refreshing step S1. FIG. 21 shows a timing chart in this case. In addition, the voltage applied to the selection line SEL is 10 V here.

As a matter of course, the pulse-shaped voltage may be applied to the selection line SEL at the same timing as the timing when a boost voltage is applied to the boost line BST. FIG. 22 shows a timing chart in this case.

The above description is applied to the pixel circuits 2B shown in FIGS. 10 and 11, and the pixel circuits 2C shown in FIGS. 12 to 15 as a matter of course, and its description is omitted.

A description will be given of the pixel circuit belonging to the third type in which the second switch circuit 23 is the series circuit composed of the transistor T1, the diode D1, and the transistor T4, and the control terminal of the transistor T4 is connected to the boost line BST.

According to the pixel circuit belonging to the third type, compared to the pixel circuit belonging to the second type, the control terminal of the transistor T4 is connected to the boost line BST, and the selection line SEL is not provided. Therefore, unlike the second type pixel circuit, the boost line BST controls the on/off of the transistor T4.

However, as shown in FIG. 22, in the second type, when the pulse voltage is applied to the selection line SEL at the same timing as the boost line BST, totally the same voltage states as that of the first type pixel circuit can be implemented. Thus, this means that totally the same voltage state can be implemented even when the control terminal of the transistor T4 is connected to the boost line BST.

Therefore, the self-refreshing action can be executed for the pixel circuit 2D shown in FIG. 16 by providing the same voltage state as that in FIG. 18. Thus, this is the same with the pixel circuit 2E shown in FIG. 17. Detailed description is omitted.

Third Embodiment

In the third embodiment, a description will be given of a case where the self-refreshing action is executed by a voltage application method different from that of the second embodiment, with reference to the drawings. In addition, the self-refreshing action of this embodiment is divided into the refreshing step S1 and the stand-by step S2, similar to the second embodiment.

According to the second embodiment, only the internal node N1 of the case H (high voltage writing) is refreshed in the phase P1, and only the internal node N1 of the case M (middle voltage writing) is refreshed in the phase P2. Thus, in the step S1, the pulse voltage needs to be applied to the boost line BST in each of the phase P1 and phase P2.

Meanwhile, according to this embodiment, only the internal node N1 of the case M (middle voltage writing) is refreshed in the phase P1, and only the internal node N1 of the case H (high voltage writing) is refreshed in the phase P2 as will be described below. Thus, in the step S1, the high level voltage is applied to the boost line BST from the phase P1 to the phase P2. Thus, the number of times to change the voltage applied to the boost line BST is reduced in the step S1, so that power consumption at the time of the self-refreshing action can be cut. Hereinafter, this operation will be described in detail.

A description will be given of a case where the self-refreshing action in this embodiment is performed for the first type pixel circuit 2A, with reference to a timing chart shown in FIG. 23. The pixel circuit 2A assumes the pixel circuit 2A shown in FIG. 7 similar to the second embodiment.

<<Step S1/Phase P1>>

It is assumed that in the phase P1, the writing node N1 (M) of the case M (middle voltage state) is the refreshing target.

In the step S1 to be started at the time t1, a voltage which can completely turn off the transistor T3 is applied to the gate line GL. Here, the voltage is −5 V. In addition, during the execution of the self-refreshing action, the transistor T3 is constantly off, so that the voltage applied to the gate line GL may remain unchanged during the self-refreshing action.

The opposite voltage Vcom applied to the opposite electrode 80, and the voltage applied to the auxiliary capacity line CSL are set to 0 V. The voltage is not limited to 0 V, and a voltage value before the time t1 may be maintained as it is. In addition, these voltages also may remain unchanged during the self-refreshing action.

In a case where the internal node N1 shows the voltage state (gradation) as refreshing target or higher voltage state (high gradation), a voltage that turns off the transistor T2 is applied to the reference line REF, while in a case where it shows the voltage state (low gradation) lower than the voltage state (gradation) as the refreshing target, a voltage that turns on the transistor T2 is applied thereto. In the phase P1, the refreshing target is the second voltage state (case M), so that the voltage that turns off the transistor T2 is applied to the reference line REF in the case where the internal node N1 is in the second voltage state (case M) and in the case where it is in the first voltage state (case H), while the voltage that turns on the transistor T2 is applied thereto in the case where it is in the third voltage state (case L).

More specifically, since the threshold voltage Vt2 of the transistor T2 is 2 V, the transistor T2 in the case L can be turned on by applying the voltage higher than 2 V to the reference line REF. However, when the voltage higher than 5 V is applied to the reference line REF, the transistor T2 in the target case M in the phase P1 comes to be also turned on. Therefore, the voltage between 2 V and 5 V is to be applied to the reference line REF. In FIG. 23, 4.5 V is applied to the reference line REF.

When 4.5 V is applied to the reference line REF, the transistor T2 is turned off in the pixel circuit in which the potential VN1 of the internal node N1 is 2.5 V or more. Meanwhile, the transistor T2 is turned on in the pixel circuit in which the VN1 is lower than 2.5 V.

As for the internal node N1 of the case M written to 3 V in the last writing action, by executing the self-refreshing action before it falls by more than 0.5 V due to the leak current, the VN1 can be 2.5 V or more, so that the transistor T2 is turned off. In addition, as for the internal node N1 of the case H written to 5 V in the last writing action, the VN1 can be 2.5 V or more for the same reason, so that the transistor T2 is turned off. Meanwhile, as for the internal node N1 of the case L written to 0 V in the last writing action, it does not reach 2.5 V or more even when a time has elapsed, so that the transistor T2 is turned on.

A voltage provided by adding the turn-on voltage Vdn of the diode D1 to the desired voltage of the internal node N1 to be restored by the refreshing action is applied to the source line SL (time t2). Here, since the refreshing target is the case M, in the phase P1 in this embodiment, the desired voltage of the internal node N1 is 3 V. Therefore, when the turn-on voltage Vdn of the diode D1 is 0.6 V, 3.6 V is applied to the source line SL. In addition, the time t1 to apply 4.5 V to the reference line REF, and the time t2 to apply 3.6 V to the source line SL may be the same time.

In addition, the desired voltage of the internal node N1 corresponds to a “refreshing desired voltage”, the turn-on voltage Vdn of the diode D1 corresponds to a “first adjusting voltage”, and the voltage actually applied to the source line SL in the refreshing step S1 corresponds to a “refreshing input voltage”. Thus, the refreshing input voltage is 3.6 V in the phase P1.

As for the boost line BST, a voltage that turns on the transistor T1 in the case M and the case H in which the transistor T2 is off as described above is applied thereto, while a voltage that turns off the transistor T1 in the case L in which the transistor T2 is in on is applied thereto (time t3). The boost line BST is connected to the one end of the boost capacitive element Cbst. Therefore, when the high level voltage is applied to the boost line BST, the potential of the other end of the boost capacitive element Cbst, that is, the potential of the output node N2 is thrust upward.

As described above, in the case M and the case H, the transistor T2 is off in the phase P1. Therefore, a potential fluctuation amount of the node N2 due to the boost upthrust is determined by a ratio between the boost capacity Cbst and the total capacity which is parasitic in the node N2. For example, when the ratio is 0.7, and one electrode of the boost capacitive element increases by ΔVbst, the other electrode, that is, the node N2 increases by about 0.7 ΔVbst.

In the case M, the potential VN1 (M) of the internal node N1 shows roughly 3 V at the time t1. When a potential higher than the VN1 (M) by the threshold voltage of 2 V or more is applied to the gate of the transistor T1, that is, the output node N2, the transistor T1 is turned on. According to this embodiment, it is assumed that the voltage applied to the boost line BST at the time t1 is 10 V. In this case, the potential of the output node N2 rises by 7 V. Since the transistor T2 is on in the writing action, the node N2 shows roughly the same potential (about 3 V) as that of the node N1 at the point just before the time t1. Thus, the node N2 shows about 10 V due to the boost upthrust. Therefore, the potential difference more than the threshold voltage is generated between the gate of the transistor T1 and the node N1, so that the transistor T1 is turned on.

In the case H also, since the node N2 shows about 12 V due to the boost upthrust, the transistor T1 is turned on.

On the other hand, in the case L in which the transistor T2 is on in the phase P1, unlike the case M and the case H, the output node N2 and the internal node N1 are electrically connected. In this case, the potential fluctuation amount of the output node N2 due to the boost upthrust is affected by the total parasitic capacitance of the internal node N1, in addition to the boost capacity Cbst and the total parasitic capacitance of the node N2.

The internal node N1 is connected to the one end of the auxiliary capacitive element Cs, and the one end of the liquid crystal capacitive element Clc, and the total capacity Cp which is parasitic in the internal node N1 is expressed by the sum of the liquid crystal capacity Clc and the auxiliary capacity Cs. Thus, the boost capacity Cbst is considerably smaller than the liquid crystal capacity Cp. Therefore, a ratio of the boost capacity to the total capacity is extremely small such as about 0.01 or less. In this case, when one electrode of the boost capacitive element increases by ΔVbst, the other electrode, that is, the output node N2 only increases by up to 0.01 ΔVbst. That is, in the case L, even when ΔVbst=10 V, the potential VN2 (L) of the output node N2 hardly increases.

In the case L, the potential VN2 (L) shows roughly 0 V at the point just before the time t1. Therefore, even when the boost upthrust is performed at the time t1, a potential sufficient to turn on the transistor is not applied to the gate of the transistor T1. That is, unlike the case M, the transistor T1 is still off.

In the case M, the transistor T1 is turned on due to the boost upthrust. In addition, 3.6 V is applied to the source line SL, so that when the potential VN1 (M) of the internal node N1 falls a little from 3 V, a potential difference more than the turn-on voltage Vdn of the diode D1 is generated between the source line SL and the internal node N1. Therefore, the diode D1 is turned on from the source line SL toward the internal node N1, and a current flows from the source line SL toward the internal node N1. Thus, the potential VN1 (M) of the internal node N1 rises. In addition, the potential continues to rise until the potential difference between the source line SL and the internal node N1 becomes equal to the turn-on voltage Vdn of the diode D1, and stops when it becomes equal to the Vdn. Here, the voltage applied to the source line SL is 3.6 V, and the turn-on voltage Vdn of the diode D1 is 0.6 V, so that the rise of the potential VN1 (M) of the internal node N1 stops at 3 V. That is, the refreshing action is executed for the case M.

In the case H also, the transistor T1 is turned on due to the boost upthrust. However, 3.6 V is applied to the source line SL. Even when the potential VN1 (H) of the internal node N1 falls a little from 5 V, its fall amount is less than 1 V. Thus, reversely-biased state is provided from the source line SL toward the internal node N1, so that the source line SL and the internal node N1 are not connected due to the rectifying action of the diode D1. That is, the potential VN1 (H) of the internal node N1 is not affected by the voltage applied to the source line SL.

Thus, in the case L, since the transistor T1 is off, the source line SL and the internal node N1 are not connected. Thus, the voltage applied to the source line SL does not affect the potential VN1 (L) of the internal node N1.

To summarize the above, the refreshing action is executed for the pixel circuit in which the potential of the internal node N1 is the refreshing isolation voltage or more and the refreshing desired voltage or less, in the phase P1. In the phase P1, the refreshing isolation voltage is 2.5 V (=4.5−2 V), and the refreshing desired voltage is 3 V, so that the refreshing action to refresh the potential VN1 to 3 V is executed only for the pixel circuit in which the potential VN1 of the internal node N1 is 2.5 V to 3 V, that is, for the case M.

<<Step S1/Phase 2>>

In the phase P2, the writing node N1 (H) of the case H (high voltage state) is the refreshing target.

The voltage applied to the boost line BST is constantly 10 V from the phase P1.

Thus, when the internal node N1 shows the voltage state (case H) serving as the refreshing target, a voltage that keeps the transistor T2 turned off is applied to the reference line REF, while when it shows the voltage state (cases M and L) lower than the voltage state (case H) serving as the refreshing target, a voltage that turns on the transistor T2 is applied thereto at the time t4.

More specifically, the threshold voltage Vt2 of the transistor T2 is 2 V, and the voltage VN1 (M) of the internal node N1 of the case M is 3 V, so that when a voltage higher than 5 V (=2+3) is applied to the reference line REF, the transistor T2 can be turned on in the case M. At this time, the transistor T2 in the case L is turned on as a matter of course.

However, when a voltage higher than 7 V is applied to the reference line REF, the transistor T2 in the case H comes to be also turned on. Therefore, formally, the voltage between 5 V and 7 V is to be applied to the reference line REF. However, since the voltage has to be applied with a view to staying on the safe side similar to the phase P1, 6.5 V is applied as one example here. This voltage 6.5 V corresponds to the refreshing reference voltage, and the voltage 4.5 V which is provided by subtracting the threshold voltage of the transistor T2 therefrom corresponds to the refreshing isolation voltage in the phase P2.

At this time, when the potential VN1 of the internal node N1 is the refreshing isolation voltage of 4.5 V or more, the transistor T2 is turned off. Meanwhile, the transistor T2 is turned on in the pixel circuit in which the VN1 is lower than 4.5 V. That is, in the case H written to 5 V in the last writing action, the VN1 is 4.5 V or more, so that the transistor T2 is turned off. Meanwhile, in the case L written to 0 V and in the case M written to 3 V in the last writing action, the VN1 is lower than 4.5 V, so that the transistor T2 is turned on.

A voltage provided by adding the turn-on voltage Vdn of the diode D1 to the desired voltage of the internal node N1 to be restored by the refreshing action is applied to the source line SL (time t5). Here, since the refreshing target is the case H in the phase P2 in this embodiment, the desired voltage of the internal node N1 is 5 V. Therefore, when the turn-on voltage Vdn of the diode D1 is 0.6 V, 5.6 V is applied to the source line SL. In addition, as will be described below, the time t5 at which 5.6 V is applied to the source line SL needs to be later than the time t4 at which 6.5 V is applied to the reference line REF in this phase P2.

In the case H, the transistor T2 still remains off state from the phase P1, and the potential of the internal node N2 holds the state of the phase P1, so that the transistor T1 is turned on. In this state, when the voltage of 5.6 V is applied to the source line SL, in the case where the potential VN1 (H) of the internal node N1 falls a little from 5 V, a potential difference more than the turn-on voltage Vdn of the diode D1 is generated between the source line SL and the internal node N1. Therefore, the diode D1 is turned on in a direction from the source line SL toward the internal node N1, and a current flows from the source lines SL toward the internal node N1. Thus, the potential VN1 (H) of the internal node N1 continues to rise until the potential difference between the source line SL and the internal node N1 becomes equal to the turn-on voltage Vdn (=0.6 V). That is, the VN1 (H) reaches 5 V, and maintains the potential. Thus, the refreshing action is executed for the case H.

The case M will be described in detail. At a stage just before the time t4 at which 6.5 V is applied to the reference line REF, the potential VN2 (M) of the node N2 is about 12 V, and the VN1 (M) is 3 V. In this state, when 6.5 V is applied to the reference line REF at the time t4, the transistor T2 is turned on in the direction from the node N2 to the node N1, and a current is generated in this direction. However, as described above, the parasitic capacitance of the node N1 is extremely larger than the parasitic capacitance of the node N2, the potential of the node N2 falls due to the current generation, but the potential of the node N1 remains unchanged. The node N2 falls until it becomes the same potential (that is, 3 V) as that of the node N1 and then potential fall stops. In addition, at this point, since the refreshing action has been already executed for the case M in the phase P1, the potential VN2 (M) of the node N2 also becomes the same potential as the VN1 (M) after the refreshing action.

When the potential of the node N2 falls below the voltage (that is, 5 V) provided by adding the threshold voltage (2 V) of the transistor T1 to the potential of the node N1, the transistor T1 is turned off. Thus, as described above, the node N2 becomes the same potential as that of the node N1, and the potential change stops, so that the transistor T1 is still off. Therefore, in this state, even when 5.6 V is applied to the source line SL, this voltage is not supplied to the node N1 (M) through the transistor T1. That is, the voltage (5.6 V) applied to the source line SL in the phase P2 does not affect the potential of the potential VN1 (M) of the internal node N1.

In other words, in the case where 5.6 V is applied to the source line SL at the time t5, in order to prevent this voltage from being supplied to the internal node N1 of the case M, the transistor T1 has to be off at the time t5. At the stage just before 6.5 V is applied to the reference line REF, the transistor T1 of the case M is on, so that in order to turn off it, after 6.5 V has been applied to the reference line REF, the potential VN2 of the node N2 has to be lower than 5 V. Therefore, after 6.5 V has been applied to the reference line REF at the time t4 and a time has passed so that the potential VN2 of the node N2 falls below 5 V, the voltage applied to the source line SL has to be changed to 5.6 V. Therefore, the time t5 at which 5.6 V is applied to the source line SL is required to be later than the time t4 at which 6.5 V is applied to the reference line REF. In FIG. 23, this is expressed by delaying the timing a little when the transistor T1 (M) shifts from on to off than the time t4.

In the case L, since the transistor T1 remains off continuously from the phase P1, the source line SL and the internal node N1 are not connected. Thus, the voltage applied to the source line SL does not affect the potential of the potential VN1 (L) of the internal node N1.

To summarize the above, in the phase P2, the refreshing action is executed for the pixel circuit in which the potential of the internal node N1 is the refreshing isolation voltage or more and the refreshing desired voltage or less. Here, since the refreshing isolation voltage is 4.5 V (=6.5-2 V), and the refreshing desired voltage is 5 V, the refreshing action to refresh the potential VN1 to 5 V is performed only for the pixel circuit in which the potential VN1 of the internal node N1 is 4.5 V to 5 V, that is, for the case H.

After the refreshing action of the case H, the voltage application to the boost line BST is stopped (time t6), and the high voltage (here, 10 V) is applied to the reference line REF to turn on the transistor T2 in each of the cases H, M, and L (time t7). Thus, the voltage application to the source line SL is stopped (time t8). In addition, the order of the times t6 to t8 is not limited to this order, and they may be executed at the same time.

<<Step S2>>

After the time t8, the process is moved to the stand-by step S2 with the voltage state unchanged (times t8 to t9). At this time, since the high voltage is applied to the reference line REF, the nodes N1 and N2 show the same potential in each of the cases H, M, and L. A time sufficiently longer than that of the reference step S1 is ensured in the stand-by step S2, which is similar to the second embodiment.

As described above, according to the self-refreshing action in this embodiment shown in FIG. 23, the number of times to fluctuate the voltage to the boost line BST can be suppressed, compared to the second embodiment shown in FIG. 18, and the power consumption can be further cut. In addition, the above description is also applied to the variation pixel circuit shown in FIG. 8 other than the pixel circuit 2A shown in FIG. 7, as a matter of course.

In addition, the order of refreshing actions of the case H and the case M can be exchanged in the second embodiment, but in this embodiment in which the number of times to fluctuate the voltage to the boost line BST is one, the refreshing action needs to be performed for the case H after the refreshing action for the case M, so that the order cannot be reversed. This is because when 10 V is applied to the boost line BST to execute the refreshing action for the case H first, the potential of the node N2 of the case M does not thrust upward, so that it is necessary to generate the voltage fluctuation in the boost line BST again to execute the refreshing action for the case M.

In addition, in this embodiment, 10 V (that can turn on the transistor T2 regardless of the cases H, M, and L) is applied to the reference line REF just before the time t1, and in the stand-by step S2, but like the second embodiment, 0 V may be applied to the reference line REF to turn off the transistor T2. However, it is to be noted that the fluctuation of the voltage applied to the reference line REF can be suppressed when the voltage application in this embodiment is performed.

In the second type pixel circuit 2B shown in FIG. 9, the transistor T4 is provided and the selection line SEL to control the on/off of the transistor T4 is provided separately from the boost line BST. Therefore, the totally the same voltage state as the first type can be implemented by applying the voltage that surely turns on the transistor T4 to the selection line SE, during the refreshing step S1. FIG. 24 shows a timing chart in this case. In addition, here, the voltage applied to the selection line SEL is 10 V.

In addition, the pulse-shaped voltage may be applied to the selection line SEL at the same timing as that when the boost voltage is applied to the boost line BST. FIG. 25 shows a timing chart in this case.

The above description can be applied to the pixel circuits 2B shown in FIGS. 10 and 11, and the pixel circuits 2C shown in FIGS. 12 to 15, as well as the pixel circuit 2B shown in FIG. 9, as a matter of course. Detailed description is omitted.

According to the pixel circuits 2D and 2E belonging to the third type, compared to the pixel circuit belonging to the second type, the control terminal of the transistor T4 is connected to the boost line BST, and the selection line SEL is not provided. Therefore, unlike the second type pixel circuit, the boost line BST controls the on/off of the transistor T4.

However, as shown in FIG. 25, when the pulse voltage is applied to the selection line SEL at the same timing as the boost line BST, in the second type, totally the same voltage states as that of the first type pixel circuit can be implemented. Thus, this means that totally the same voltage state can be implemented when the control terminal of the transistor T4 is connected to the boost line BST.

Therefore, the self-refreshing action can be executed for the pixel circuit 2D shown in FIG. 16 by providing the same voltage state as that in FIG. 25. Thus, this is applied to the pixel circuit 2E shown in FIG. 17. Detailed description is omitted.

Fourth Embodiment

According to a fourth embodiment, a description will be given of a case where a voltage application method is partially changed and a self-refreshing action is executed, based on the self-refreshing method of the third embodiment, with reference to the drawings.

As described above, the self-refreshing action can be performed by the method of the third embodiment, but when this method is repeatedly executed, a following problem could be caused. According to a self-refreshing method in this embodiment, it is possible to solve the problem which could be caused when the self-refreshing action is repeatedly executed by the method of the third embodiment.

First, the problem which could be caused by the self-refreshing method of the third embodiment will be described. Here, the description will be given of the case where the self-refreshing action shown in FIG. 23 is performed for the pixel circuit 2A shown in FIG. 7, but the same discussion can be applied to other pixel circuits.

FIG. 26 is a timing chart exaggeratingly showing a problem which could be caused when totally the same self-refreshing action as that in FIG. 23 is performed.

As described above, when the refreshing action is performed, the voltages applied to the reference line REF and the boost line BST are raised or lowered. When the voltage applied to the reference line REF is abruptly increased/reduced, the potential fluctuations of the nodes N1 and N2 could be generated due to the parasitic capacitance of the transistor (T2 especially) in the pixel circuit. After the refreshing action has been repeatedly executed, this potential fluctuation reaches a level which cannot be ignored, and as a result, the refreshing action cannot be correctly performed. Hereinafter, this point will be described.

When a voltage applied to the reference line REF is reduced from 10 V to 4.5 V at a time t1, potentials of the nodes N1 and N2 are also thrust downward to a certain level due to the reduction of the voltage applied to the REF. This reduction in potential is shown in the timing chart in FIG. 26 (refer to FIG. 23 and FIG. 26).

At a time t2, a voltage applied to the source line SL is set to 3.6 V, and then at a time t3, a voltage applied to the boost line BST is increased to 10 V. At this time, as described above in the third embodiment, the potential of the node N2 is largely thrust upward in the case H and the case M in which the transistor T2 is in off state.

In the case M, when the transistor T1 is turned on due to the potential rise of the node N2, the voltage applied to the source line SL is supplied to the internal node N1. Since 3.6 V is applied to the source line SL, the potential VN1 (M) of the internal node N1 is increased to 3 V which is provided by subtracting the turn-on voltage Vdn (=0.6 V) of the diode D1 from the voltage applied to the source line SL.

In the case H, since the voltage applied to the source line SL is lower than the potential of the internal node N1, the source line SL and the internal node N1 are not electrically connected due to a rectifying action of the diode D1. As a result, the potential of the internal node N1 is not affected by the voltage applied to the source line SL. This point is the same as that of the third embodiment.

However, also in the case H, due to the existence of the parasitic capacitance of the node N1, the potential of the node N1 is slightly increased when the potential of the BST line is thrust upward. This is the same in the case L. These potential increases are shown in the timing chart in FIG. 26 (refer to FIG. 23 also).

In addition, in the case M, since the VN1 (M) is affected by the voltage applied to the source line SL, and its increase stops when it reaches 3 V like the third embodiment.

Then, at a time t4, the voltage applied to the reference line REF is increased to 6.5 V. For a reverse reason from that when the potentials of the nodes N1 and N2 are reduced at the time t1, the potential values of the nodes N1 and N2 are slightly increased in each case.

In addition, in the case M, since the transistor T2 is turned on due to the increase of the voltage applied to the REF, each of the nodes N1 and N2 reaches a middle potential between the VN1 (M) and the VN2 (M) provided at a point just before the time t4. However, as described above in the third embodiment, since the parasitic capacitance of the node N1 is sufficiently large, compared with the node N2, the middle potential is drawn to the potential VN1 (M) of the node N1 in practice, but it is slightly increased from a value of the VN1 (M) provided at the point just before the time t4. That is, after the time t4, each of the VN1 (M) and VN2 (M) shows a value which is slightly increased from 3 V.

Then, when the voltage applied to the source line SL is set to 5.6 V at a time t5, the voltage applied to the source line SL is supplied to the internal node N1 only in the case H because the transistor T1 is on state only in the case H. As a result, the potential of the internal node N1 (H) is refreshed to 5 V. This is the same as that of the third embodiment.

Then, at a time t6, the voltage applied to the boost line BST is lowered to 0 V. At this time, as described above in the third embodiment, in the case H in which the transistor T2 is in off state, the potential of the node N2 is largely thrust downward. Thus, like the time t3, the transistor T2 in off state functions as the capacitive element, so that the potential of the node N1 (H) is also slightly thrust downward.

In addition, also in the cases M and L, for a reverse reason from that when the potentials of the nodes N1 and N2 are increased at the time t3, the potential values of the nodes N1 and N2 are slightly reduced in each case.

Then, at a time t7, the voltage applied to the reference line REF is increased to 10 V. At this time, the potential of the node N1 is slightly increased due to the increase of the voltage applied to the REF line. In addition, when the 10 V is applied to the REF line, the transistor T2 is turned on, so that the potential of the node N2 reaches the same value as the potential of the node N1.

At this time, as for the case M especially, although the VN1 (M) has been refreshed to 3V at the time t3, the potential VN1 (M) is slightly increased at the time t4. After that, the VN1 (M) is lowered due to the reduction of the voltage applied to the BST line at the time t6, but the VN1 (M) is slightly increased again due to the increase of the voltage applied to the REF line at the time t7. As a result, the potential of the VN1 (M) is slightly higher than 3 V at the end of the refreshing action (refer to an arrow E1 in FIG. 26).

In order to prevent such a thing from occurring, according to the self-refreshing action in this embodiment, the voltage is applied in a sequence partially different from that in the third embodiment.

FIG. 27 is a timing chart showing the self-refreshing action in this embodiment. Similar to FIG. 26, a description will be given of the case where the self-refreshing action is performed for the pixel circuit 2A in FIG. 7. In addition, in the timing chart shown in FIG. 27, similar to the case shown in FIG. 26, the voltage applied to the REF line takes into account the fluctuations of the potentials of the nodes N1 and N2 due to the parasitic capacitance when the voltage applied to the BST line is changed.

Actions from times t1 to t4 are the same as those in FIG. 26, so that their description is omitted.

At a time t5, the voltage applied to the source line SL is slightly increased in this embodiment, compared with the case in FIG. 26. Here, the voltage is 5.7 V which is higher by 0.1 V.

Thus, the VN1 (H) shows a value provided by decreasing the turn-on voltage (0.6 V here) of the diode D1 from 5.7 V, that is, 5.1 V. That is, the potential is slightly increased from 5 V which is the refreshing desired voltage. In addition, as for the VN2 (H) and the potentials of the nodes N1 and N2 in the other cases, they are the same as those in FIG. 26.

Then, at a time t6, the voltage applied to the REF line is reduced from 6.5 V to 0 V. Thus, the potentials of the nodes N1 and N2 in each case are slightly reduced, and the transistor T2 is turned off.

Then, at a time t7, the voltage applied to the BST line is reduced from 10 V to 0 V. This is the same action as that at the time t6 in FIG. 26.

In the case H, the potential VN1 (H) of the node N1 is slightly reduced for a reverse reason from that when the VN1 (H) is increased at the time t3. In addition, as for the potential VN2 (H) of the node N2, since the transistor T2 is off state at the point of the time t6, it is largely thrust downward in tandem with the reduction of the voltage applied to the BST line. Similar to the second embodiment, in the case where the ratio between the boost capacity Cbst and the whole capacity parasitic in the node N2 is 0.7, the VN2 (H) is reduced to a potential slightly lower than 5 V at the time t7.

In the case M, the potential VN1 (M) of the node N1 is slightly reduced for the same reason as that of the VN1 (H), and reaches a value slightly lower than 3 V. In addition, as for the potential VN2 (M) of the node N2, since the transistor T2 is in off state at the point at the time t6, similar to the case H, it is largely thrust downward in tandem with the reduction of the voltage applied to the BST line.

Here, it is to be noted that in the case M, since the VN2 (M) shows 3 V at the point of the time t7, it shows a negative potential lower than 0 V when the BST line is reduced by 10 V. However, at the moment the potential is largely reduced, the transistor T2 is turned on from the nodes N1 to N2, and the VN2 (M) is increased. Thus, similar to the second embodiment, when the threshold voltage of the transistor T2 is 2V, the potential of the VN2 (M) is increased to about −2 V which is lower than the voltage 0 V by 2 V, the voltage 0V being applied to the REF line and serving as a gate potential, and this is maintained.

In the case L, the potentials of the nodes N1 and N2 show the same behavior as those in the case M. As for the potential VN1 (L) of the node N1, it is slightly reduced for the same reason as that of the VN1 (H), and shows a value slightly lower than 0 V. In addition, the potential VN2 (L) of the node N2 is largely reduced instantaneously, but after that, the transistor T2 is turned on and the VN2 (L) is increased. Thus, similar to the VN2 (M), the VN2(L) is increased to about −2 V which is lower than the voltage 0 V by 2 V, the voltage 0V being applied to the REF line and serving as a gate potential, and this is maintained.

Then, at a time t8, the voltage applied to the REF line is increased from 0 V to 10 V. At this time, for the same reason as that when the voltage applied to the REF line is increased at the time t4, the potentials of the nodes N1 and N2 are slightly increased. That is, the VN1 (H) slightly lower than 5 V at a point just before the time t8 is increased to 5 V, the VN1 (M) slightly lower than 3 V is increased to 3 V, and the VN1 (L) slightly lower than 0 V is increased to 0 V.

In addition, when the voltage applied to the REF line is increased, the transistor T2 is turned on in each of the cases H, M, and L, and the potential VN2 of the node N2 is changed in a direction to the potential VN1 of the node N1. That is, the VN2 is also increased to the same potential as the VN1.

After that, the voltage application to the source line SL is stopped, and a process is moved to the standby step S2 similar to the second embodiment.

As described with reference to FIG. 26, according to the case of the self-refreshing method of the second embodiment, the action of increasing the voltage applied to the REF line is performed at the end of the refreshing step S1 to turn on the transistor T2. Thus, at the point just before this action, the potential VN1 (M) of the node N1 in the case M especially is set at 3 V which is the refreshing desired voltage. Therefore, it is likely that the VN1 (M) is slightly increased in tandem with the increasing action of the voltage applied to the REF line, and the refreshing action is completed in a state where the VN1 (M) is higher than the desired voltage of 3 V.

Meanwhile, according to the self-refreshing method in this embodiment, the actions are performed such that in the stage prior to the time t8 to perform the increasing action of the voltage applied to the REF line, the voltage applied to the REF line is reduced once at the time t6 to turn off the transistor T2 in each case, and the voltage applied to the BST line is reduced at the time t7. Therefore, at the point just before the voltage applied to the REF line is increased at the time t8, the VN1 (M) shows the potential slightly lower than 3 V which is the refreshing desired voltage, so that when the voltage applied to the REF line is increased at the time t8, the VN1 (M) is slightly increased and reaches the desired voltage of 3 V.

In addition, according to this embodiment, the voltage applied to the source line SL at the point of the time t5 shows the value slightly higher than the value (5.6 V here) provided by adding the turn-on voltage of the diode to the refreshing desired voltage in the case H. This is because the VN1 (H) is set to the value slightly higher than the desired potential on the assumption that the VN1 (H) is reduced when the voltage applied to the REF line is reduced from 6.5 V to 0 V at the time t6.

Fifth Embodiment

According to a fifth embodiment, a description will be given of the writing action in the constant display mode with reference to the drawings.

According to the writing action in the constant display mode, pixel data for one frame is divided with respect to each display line in the horizontal direction (row direction), and a voltage corresponding to each pixel data for the one display line is applied to the source line SL in each column. Here, similar to the second embodiment, three gradations are assumed as the pixel data. That is, a high level voltage (5 V), a middle level voltage (3 V), or a low level voltage (0 V) is applied to the source line SL. Thus, a selected row voltage 8 V is applied to the gate line GL of the selected display line (selected row) to turn on the first switch circuits 22 of all the pixel circuits belonging to the selected row, and the voltage of the source line SL in each column is transferred to the internal node N1 of each pixel circuit 2 in the selected row.

In addition, an unselected row voltage −5 V is applied to the gate line GL (unselected row) except for the selected display line to turn off the first switch circuits 22 of all the pixel circuits 2 in the selected row. In addition, the timing control of the voltage applied to each signal line in the writing action as will be described below is performed by the display control circuit 11, and individual voltage application is performed by the display control circuit 11, the opposite electrode drive circuit 12, the source driver 13, and the gate driver 14.

First, a description will be given of the pixel circuit belonging to the first type, in which the second switch circuit 23 is the series circuit composed of the transistor T1 and the diode D1 only.

FIG. 28 shows a timing chart of the writing action using the first type pixel circuit 2A (FIG. 7). FIG. 28 illustrates a voltage waveform of each of the two gate lines GL1 and GL2, the two source lines SL1 and SL2, the reference line REF, the auxiliary capacity line CSL, and the boost line BST for the one frame period, and a voltage waveform of the opposite voltage Vcom.

In addition, FIG. 28 also illustrates the waveforms of the potentials VN1 of the internal nodes N1 of the four pixel circuits 2A. These four pixel circuits 2A are the pixel circuit 2A (a) selected by the gate line GL1 and the source line SL1, the pixel circuit 2A (b) selected by the gate line GL1 and the source line SL2, the pixel circuit 2A (c) selected by the gate line GL2 and the source line SL1, and the pixel circuit 2A (d) selected by the gate line GL2 and the source line SL2. In the drawing, (a) to (d) are added behind the internal node potentials VN1 to be discriminated.

The one frame period is divided into the horizontal periods whose number corresponds to the number of the gate lines GL, and the gate lines GL1 to GLn to be selected in the horizontal periods are sequentially allocated to them. FIG. 28 illustrates voltage changes of the two gate lines GL1 and GL2 in the first two horizontal periods. In the one horizontal period, the selected row voltage 8 V is applied to the gate line GL1, and unselected row voltage −5 V is applied to the gate line GL2, and in the second horizontal period, the selected row voltage 8 V is applied to the gate line GL2, and the unselected row voltage −5 V is applied to the gate line GL1. In the following horizontal period, the unselected row voltage −5 V is applied to both gate lines GL1 and GL2.

The voltages (5 V, 3 V, and 0 V) which correspond to the pixel data of the display line corresponding to each horizontal period are applied to the source line SL in each column. FIG. 28 illustrates the two source lines SL1 and SL2 as a representative of the source line SL. In addition, FIG. 28 shows the voltages 5 V, 3 V, and 0 V of the two source lines SL1 and SL2 for the first two horizontal periods. After those periods, the three-valued voltage corresponding to the pixel data is applied thereto. In FIG. 28, “D” is illustrated to show that this is a voltage value depending on the data.

FIG. 28 shows a case, as one example, where the high level voltage is written in the pixel circuit 2A (a), and the low level voltage is written in the pixel circuit 2A (b) in the first horizontal period h1, and the middle level voltage is written in the pixel circuits 2A (c) and 2A (d) in the second horizontal period h2.

It is assumed that, as one example, the pixel circuits 2A (a) to 2A(d) at the point just before the writing action are written such that the 2A (a) is roughly to 0 V (low voltage state), 2A (b) and 2A (c) are roughly to 3 V (middle voltage state), and 2A (d) is roughly to 5 V (high voltage state). In addition, the term “roughly” is used in view of the potential change over time due to the leak current as described in the second embodiment.

That is, it is assumed that by the writing action of this embodiment, the pixel circuit 2A (a) is written from 0 V to 5 V, 2A (b) is written from 3 V to 0 V, 2A (c) is continuously written to 3 V, and 2A (d) is written from 5 V to 3 V.

During the writing action (one frame period), a voltage to constantly keep the transistor T2 in on state, regardless of the voltage state of the internal node N1, is applied to the reference line REF. Here, the voltage is 8 V. This voltage is to be a value greater than a value provided by adding the threshold voltage (2 V) of the transistor T2 to the potential VN1 (5 V) of the internal node N1 written in the high voltage state. Thus, the output node N2 and the internal node N1 are electrically connected, and the auxiliary capacitive element Cs connected to the internal node N1 can be used to stabilize the internal node potential VN1.

In addition, during the writing period, the boost thrusting action is not performed, so that the low level voltage (here, 0 V) is applied to the boost line BST. The auxiliary capacity line CSL is fixed to a predetermined fixed voltage (such as 0 V). As the opposite voltage Vcom is subjected to the opposite AC driving as described above, it is fixed to the high level voltage (5 V) or the low level voltage (0 V) during the one frame period. In FIG. 28, the opposite voltage Vcom is fixed to 0 V.

In the first horizontal period h1, the selected row voltage is applied to the gate line GL1, and the voltage corresponding to the pixel data is applied to the source line SL. In addition, 5 V is applied to the source line SL1 and 0 V is applied to the source line SL2 to write 5 V in the pixel circuit 2A (a) and 0 V in the pixel circuit 2A (b), respectively among the pixel circuits in which the control terminals of the transistors T3 are connected to the gate line GL 1. Similarly, the voltage according to the pixel data is applied to the other source line.

In the first horizontal period h1, the transistor T3 is turned on in each of the pixel circuits 2A (a) and 2A (b), so that the voltage applied to the source line SL is written to the internal node N1 through the transistor T3.

Meanwhile, in the first horizontal period h1, the transistor T3 is off in the pixel circuit whose control terminal of the transistor T3 is connected to the gate line GL except for the gate line GL1, so that the voltage applied to the source line SL is not applied to the internal node N1 through the first switch circuit 22.

Here, the pixel circuit 2A (c) selected by the gate line GL2 and the source line SL1 is to be focused on. As for the pixel circuit 2A (c), the control terminal of the transistor T3 is connected to the gate line GL2, so that the transistor T3 is off as described above, and the voltage (5 V) applied to the source line SL1 is not written in the internal node N1 through the first switch circuit 22.

Thus, the potential VN1 (c) of the internal node N1 shows roughly 3 V just before the writing, and the internal node N1 and the output node N2 show the same potential, so that the gate potential of the transistor T1 shows roughly 3 V. Since 5 V is applied to the source line SL1, the transistor T1 is turned off. Therefore, the voltage applied to the source line SL1 is not written in the internal node N1 through the second switch circuit 23.

Thus, the VN1 (c) still remains the potential at the point just before the writing action, in the first horizontal period h1.

Next, the pixel circuit 2A (d) selected by the gate line GL2 and the source line SL2 is to be focused on. As for the pixel circuit 2A (d) also, the control terminal of the transistor T3 is connected to the gate line GL2, similar to the pixel circuit 2A (c), so that the transistor T3 is off. Therefore, the voltage (0 V) applied to the source line SL2 is not applied to the internal node N1 through the first switch circuit 22.

Thus, the potential VN1 (d) of the internal node N1 shows roughly 5 V just before the writing. Since 0 V is applied to the source line SL2, a reversely-biased voltage is applied to the diode D1. Therefore, the voltage (0 V) applied to the source line SL2 is not applied to the internal node N1 through the second switch circuit 23.

Thus, the VN1 (d) also still remains the potential at the point just before the writing action, in the first horizontal period h1.

Meanwhile, in the second horizontal period h2, in order to write 3 V in the pixel circuit 2A (c) and 2A (d), the selected row voltage is applied to the gate line GL2, the unselected row voltage is applied to the other gate line GL, 3 V is applied to the source line SL1 and the SL2, and the voltage corresponding to the pixel data of the pixel circuit selected by the gate line GL2 is applied to the other source line SL. As for the pixel circuits 2A (c) and 2A (d), the voltage applied to the source line SL is applied to the internal node N1 through the first switch circuit 22. Thus, as for the pixel circuits 2A (a) and 2A (b), the first switch circuit 22 is off, and the diode D1 is in the reversely-biased state, or the transistor T1 is turned off in the second switch circuit 23, so that the voltage applied to the source line SL is not applied to the internal node N1.

Through the above voltage application, for only the selected pixel circuit, the voltage according to the pixel data is applied from the source line SL to the internal node N1 through the first switch circuit 22.

In addition, the description has been given assuming that the pixel circuit is the pixel circuit 2A shown in FIG. 7 in the above embodiment, the same writing action can be implemented in the pixel circuit 2A shown in FIG. 8, as a matter of course.

The second type assumes the pixel circuits 2B (FIGS. 9 to 11) in which the first switch circuit 22 is only composed of the transistor T3, and the pixel circuits 2C (FIGS. 12 to 15) in which the first switch circuit 22 is the series circuit composed of the transistors T3 and T4 (or T5), as described above.

As described in the first type, at the time of writing action, the second switch circuit 23 is turned off, and the voltage is applied from the source line SL to the internal node N1 through the first switch circuit 22. As for the pixel circuit 2B, the second switch circuit 23 can be surely off at the time of writing action, by constantly keeping the transistor T4 in the off state. In addition, as for the rest, the writing action can be implemented by the same method as that of the first type. FIG. 29 shows a timing chart of the writing action using the second type pixel circuit 2B (FIG. 9). In addition, in FIG. 29, in order to keep the transistor T4 in the off state during the writing action, −5 V is applied to the selection line SEL.

Meanwhile, as shown in FIGS. 12 to 15, in the case where the first switch circuit 22 is the series circuit composed of the transistors T3 and T4 (or T5), in order to turn on the first switch circuit 22, the transistor T4 (or T5) has to be turned on in addition to the transistor T3, at the time of writing action. In addition, as for the pixel circuit 2C shown in FIG. 15, the first switch circuit 22 is provided with the transistor T5, and the transistor T5 and the transistor T4 are connected through their control terminals, so that the conduction control of the first switch circuit 22 can be performed by controlling the conduction of the transistor T4, similar to the other pixel circuit 2C.

To summarize the above, as for the pixel circuit 2C, all the selection lines SEL are not collectively controlled like the pixel circuit 2B, but they need to be controlled individually with respect to each row like the gate line GL. That is, the selection lines SEL are provided in respective rows as many as the gate lines GL1 to GLn, and sequentially selected similar to the gate lines GL1 to GLn.

FIG. 30 shows a timing chart of the writing action using the second type pixel circuit 2C (FIG. 12). FIG. 30 illustrates voltage changes of the two selection lines SEL1 and SEL2 in the first two horizontal periods. In the first horizontal period, the selecting voltage 8 V is applied to the selection line SEL1, and non-selecting voltage −5 V is applied to the selection line SEL2, and in the second horizontal period, the selecting voltage 8 V is applied to the selection line SEL2, and the non-selecting voltage −5 V is applied to the selection line SEL1. In the following horizontal period, the non-selecting voltage −5 V is applied to both selection lines SEL1 and SEL2. The rest is the same as the timing chart of the writing action of the first type pixel circuit 2A shown in FIG. 28. Thus, the same voltage state as the first type pixel circuit 2A shown in FIG. 28 can be implemented. Detailed description is omitted.

The third type pixel circuit is different from the second type in that the selection line SEL is not provided, and the boost line BST is connected to the control terminal of the transistor T4. Therefore, the voltage may be applied to the boost line BST by the same method as that used for applying the voltage to the selection line SEL in the second type. FIG. 31 shows a timing chart of the writing action using the third type pixel circuit 2D (FIG. 16).

In addition, at this time, 8 V is applied to the reference line REF, and the transistor T2 is constantly on, so that even when the voltage applied to the boost line BST rises, the potential VN2 of the output node N2 hardly rises, and the transistor T1 is not turned on.

Sixth Embodiment

In a sixth embodiment, a description will be given of a relationship between the self-refreshing action and the writing action in the constant display mode.

In the constant display mode, after the writing action has been executed for the image data for the one frame, the writing action is not performed for a certain period and the display contents provided by the last writing action are maintained.

By the writing action, a voltage is applied to the internal node N1 (pixel electrode 20) in the pixel through the source line SL. Then, the gate line GL becomes low level, and the transistor T3 is turned off. However, the potential VN1 of the internal node N1 is maintained due to the presence of the electric charges accumulated in the pixel electrode 20 by the last writing action. That is, the voltage Vlc is maintained between the pixel electrode 20 and the opposite electrode 80. Thus, after the completion of the writing action, the voltage required to display the image data is kept applied to between both ends of the liquid crystal capacity Clc.

In the case where the potential of the opposite electrode 80 is fixed, the liquid crystal voltage Vlc depends on the potential of the pixel electrode 20. This potential fluctuates with time due to the generation of the leak current of the transistor in the pixel circuit 2. For example, in the case where the potential of the source line SL is lower than the potential of the internal node N1, the leak current generates from the internal node N1 to the source line SL, and the potential VN1 of the internal node N1 gradually decreases with time. On the other hand, in the case where the potential of the source line SL is higher than the potential of the internal node N1 (especially, in the case where the low voltage state is written), the leak current is generated from the source line SL toward the internal node N1, and the VN1 increases with time. That is, after the time has elapsed without externally executing the writing action, the liquid crystal voltage Vlc gradually changes, and as a result, a display image also changes.

In the normal display mode, the writing action is executed for all the pixel circuits 2 with respect to each frame even when the image is the still image. Therefore, the electric charge amount accumulated in the pixel electrode 20 needs to be held for only one frame period. Since the potential fluctuation amount of the pixel electrode 20 for the one frame period is very small, the potential fluctuation in this period does not affect the displayed image data to such a degree that it can be visually recognized. Therefore, in the normal display mode, the potential fluctuation of the pixel electrode 20 can be ignored.

Meanwhile, in the constant display mode, the writing action is not executed with respect to each frame. Therefore, while the potential of the opposite electrode 80 is fixed, it is necessary to hold the potential of the pixel electrode 20 over the several frames in some cases. However, when left over the several frames without executing the writing action, the potential of the pixel electrode 20 fluctuates intermittently due to the above-described generation of the leak current. As a result, the display image data could change to a degree that it can be visually realized.

In order to prevent this phenomenon from being generated, in the constant display mode, the self-refreshing action and the writing action are combined and executed in a manner shown in a flowchart in FIG. 32, so that while the potential fluctuation of the pixel electrode is suppressed, power consumption is considerably cut.

First, the writing action of the pixel data for the one frame in the constant display mode is executed in the manner described in the fifth embodiment (step #1).

After the writing action in the step #1, the self-refreshing action is executed in the manner described in the second embodiment (step #2). As described above, the self-refreshing action is composed of the refreshing step S1 and the stand-by step S2.

Here, when a request for the writing action of new pixel data (data rewriting), the external refreshing action, or the external polarity inverting action is received during the stand-by step S2 (YES in a step #3), the process returns to the step #1, and the writing action of the new pixel data or the previous pixel data is executed. When the above request is not received during the stand-by step S2 (NO in step #3), the process returns to the step #2, and the self-refreshing action is executed again. Thus, the display image is prevented from being changed due to the leak current.

When the refreshing action is performed by the writing action without performing the self-refreshing action, the power consumption is as expressed by the relational expression shown in the above formula 1, but in a case where the self-refreshing action is repeated at the same refreshing rate, and each pixel circuit holds three-valued pixel data, a variable number n in the formula 1 is 2 because the number of times to drive all the source lines is 2 like the fifth embodiment, so that when VGA is assumed as display resolution (pixel number), the number is such that m=1920, and n=480, and as a result, power consumption can be expected to be cut to about one-240th.

The reason why the self-refreshing action and the external refreshing action or the external polarity inverting action are combined in this embodiment is to deal with a case where even when the pixel circuit 2 normally operates at first, a defect is generated in the second switch circuit 23 or the control circuit 24 due to a change over time, and a state in which the writing action can be performed without any problem but the self-refreshing action cannot be normally executed is generated in some pixel circuits 2. That is, when only depending on the self-refreshing action, the display of the some pixel circuits 2 deteriorates, and it is fixed, but by combining with the external polarity inverting action, the display defect can be prevented from being fixed.

Seventh Embodiment

In a seventh embodiment, a description will be given of the writing action in the normal display mode, with reference to the drawing with respect to each type.

According to the writing action in the normal display mode, the pixel data for the one frame is divided with respect to each display line in the horizontal direction (row direction), a multi-gradation analog voltage corresponding to the pixel data for the one display line is applied to the source line SL of each row with respect to each horizontal period, and the selected row voltage 8 V is applied to the gate line GL of the selected display line (selected row) to turn on the first switch circuits 22 of all the pixel circuits 2 in the selected row and transfer the voltage of the source line SL of each row to the internal node N1 of each pixel circuit in the selected row. The unselected row voltage −5 V is applied to the gate line GL (unselected row) except for the selected display line to turn off the first switch circuits 22 of all the pixel circuits 2 in the unselected row.

In addition, unlike the constant display mode, according to the writing action in the normal display mode, the opposite voltage Vcom changes with respect to each horizontal period (opposite AC driving), so that the auxiliary capacity line CSL is driven so as to become the same voltage as the opposite voltage Vcom. This is because the pixel electrode 20 is capacitively coupled with the opposite electrode 80 through the liquid crystal layer, and also capacitively coupled with the auxiliary capacity line CSL through the auxiliary capacitive element Cs, so that when the voltage of the auxiliary capacitive element Cs is fixed, only the Vcom fluctuates in the formula 2, which induces fluctuation of the liquid crystal voltage Vlc of the pixel circuit 2 in the unselected row. Therefore, the voltages of the opposite electrode 80 and the pixel electrode 20 are changed in the same voltage direction by driving all the auxiliary capacity line CSL at the same voltage as the opposite voltage Vcom to offset the effect of the opposite AC driving.

The writing action in the normal display mode is the same as that in the constant display mode in principle except that the opposite AC driving is performed, and the analog voltage of the multi-gradation more than that of the constant display mode is applied from the source line SL, so that detailed description is omitted. FIG. 33 shows a timing chart of the writing action in the normal display mode for the first type pixel circuit 2A (FIG. 7). In addition, in FIG. 33, the analog voltage of the multi-gradation corresponding to the pixel data of the analog display line is applied to the source line SL, so that the applied voltage cannot be unambiguously specified between a minimum value VL and a maximum value VH, and this is expressed by a shaded part.

Similarly, FIG. 34 shows a timing chart of the writing action using the second type pixel circuit 2C (FIG. 12).

In this embodiment, a method to invert the polarity of each display line with respect to each horizontal period in the writing action in the normal display mode is used because the following inconvenience generated when the polarity is inverted with respect to each frame is to be solved. In addition, a method to solve such inconvenience includes a method to invert the polarity with respect to each column, and a method to invert the polarity with respect to each pixel in the row and column directions at the same time.

A case is assumed such that a positive liquid crystal voltage Vlc is applied to all the pixels in a certain frame F1, and a negative liquid crystal voltage Vlc is applied to all the pixels in the next frame F2. Even when the voltage having the same absolute value is applied to the liquid crystal layer 75, a slight difference is generated in some cases in optical transmittance depending on whether it is positive or negative. In a case where high-quality still image is displayed, this slight difference could generate a fine change in a display manner between the frame F1 and the frame F2. In addition, in a case where a moving image is displayed also, a fine change could be generated in its display manner, in a display region to display the same contents between the frames. In displaying the high-quality still or moving image, even such fine change could be visually recognized.

Thus, since such high-quality still or moving image is displayed in the normal display mode, the above fine change could be visually recognized. In order to avoid this phenomenon, the polarity is inverted with respect to each display line in the same frame in this embodiment. Thus, since the liquid crystal voltages Vlc having different polarities are applied between the display lines in the same frame, the display image data is prevented from being affected by the polarity of the liquid crystal voltage Vlc.

Other Embodiments

Hereinafter other embodiments will be described.

<1> The description has been given assuming that the constant display mode serving as the target of the self-refreshing action is smaller in display color number than the normal display mode. However, by increasing the gradation number to increase the display color number to a certain level, the liquid crystal display may be implemented only by the constant display mode. In this case, the full-color display cannot be implemented like the normal display mode, but the display process can be performed only by the constant display mode of the present invention, for a screen in which the required displayable color number is not so many.

In addition, when the gradation number increases, the number of times to apply the pulse increases in the self-refreshing action in the second embodiment, that is, the phase number also increases in the refreshing step S1. The second embodiment can be implemented with the phases P1 and P2 in the case of the three values, but three phases are needed in the case of four gradations, and four phases are needed in the case of five gradations.

Meanwhile, according to the method of the third embodiment, with the voltage to the boost line BST kept constant from the start of the phase P1, the number of the voltage applications to the reference line REF, and the number of the voltage application to the source line SL is changed to (gradation number−1).

In addition, as the values of the pixel data in the constant display mode, 5 V, 3 V, and 0 V are employed in the above embodiments, the values are not limited to the above voltage values, as a matter of course.

<2> As for the second type pixel circuits 2B (FIGS. 9 to 11), the low level voltage may be applied to the reference line REF at the time of writing actions in the normal display mode and the constant display mode to turn off the transistor T2. In this case, the internal node N1 and the output node N2 are electrically isolated, and as a result, the potential of the pixel electrode 20 is not affected by the voltage of the output node N2 before the writing action. Thus, the voltage of the pixel electrode 20 correctly reflects the voltage applied to the source line SL, and the image data can be displayed without an error.

<3> In the above embodiments, the second switch circuit 23 and the control circuit 24 are provided with respect to each pixel circuit 2 formed on the active matrix substrate 10. Meanwhile, in a case where two kinds of pixel parts such as a transmissive pixel part to perform a transmissive liquid crystal display, and a reflective pixel part to perform a reflective liquid crystal display are provided on the active matrix substrate 10, the second switch circuit 23 and the control circuit 24 may be provided only for the pixel circuit of the reflective pixel part, and the second switch circuit 23 and the control circuit 24 may not be provided for the pixel circuit of the transmissive display part.

In this case, the image is displayed by the transmissive pixel part in the normal display mode, and the image is displayed by the reflective pixel part in the constant display mode. In this configuration, the number of elements formed on the whole of the active matrix substrate 10 can be reduced.

<4> The pixel circuit 2 has the auxiliary capacitive element Cs in the above embodiments, but the auxiliary capacitive element Cs may not be provided. However, it is preferable to provide the auxiliary capacitive element Cs in order to further stabilize the potential of the internal node N1, and surely stabilize the display image.

<5> It is assumed that the display element part 21 of the pixel circuit 2 is only composed of the unit liquid crystal display element Clc in the above embodiments, but as shown in FIG. 35, an analog amplifier Amp (voltage amplifier) may be provided between the internal node N1 and the pixel electrode 20. In FIG. 35, as one example, the auxiliary capacity line CSL and a power supply line Vcc are inputted as a power supply line of the analog amplifier Amp.

In this case, the voltage applied to the internal node N1 is amplified at a amplification factor η set by the analog amplifier Amp, and the amplified voltage is supplied to the pixel electrode 20. Thus, a fine voltage change of the internal node N1 can be reflected on the display image.

In addition, in this configuration, the voltage of the internal node N1 is amplified at the amplification factor η and supplied to the pixel electrode 20, in the self-polarity-inverting action in the constant display mode, so that the voltages in the first and second voltage states supplied to the pixel electrode 20 can be conformed to the high level and low level voltages of the opposite voltage Vcom by adjusting a difference in voltage between the first and second states applied to the source line SL.

<6> The N channel type polycrystalline silicon TFT are assumed as the transistors T1 to T4 in the pixel circuit 2 in the above embodiments, but a P channel type TFT or amorphous silicon TFT may be used. In this case, the pixel circuit 2 can be operated in the same manner as the above embodiments by inverting a height relationship of the voltages or a rectifying direction of the diode D1, and the same effect can be provided.

<7> The description has been given of the liquid crystal display device in the above embodiments, but the present invention is not limited to this, and the present invention can be applied to any display device as long as it has capacity corresponding to the pixel capacity Cp for holding the pixel data, and displays an image based on a voltage held in the capacity.

For example, in a case of an organic EL (Electroluminescence) display device which displays an image by holding a voltage corresponding to pixel data in capacity corresponding to pixel capacity, the present invention can be applied to the self-refreshing action especially. FIG. 36 is a circuit diagram showing one example of a pixel circuit of the organic EL display device. In this pixel circuit, a voltage held in the auxiliary capacity Cs as the pixel data is applied to a gate terminal of a driving transistor Tdv composed of a TFT, and a current corresponding to the voltage flows to a light emitting element OLED through the driving transistor Tdv. Therefore, the auxiliary capacity Cs corresponds to the pixel capacity Cp in the above embodiments.

In addition, as for the pixel circuit shown in FIG. 36, unlike the liquid crystal display device which displays the image by controlling optical transmittance by applying the voltage to between electrodes, it displays an image by light emission of the element when a current flows in the element. Therefore, the polarity of the voltage applied to between both ends of the element cannot be inverted due to a rectifying property of the light emitting element, and what is more, it is not needed.

<8> In the second embodiment, the self-refreshing action of the second type pixel circuit has been described with reference to the timing charts in FIGS. 21 and 22. The second type pixel circuits 2B and 2C (FIGS. 9 to 15) are provided with the transistor T4, and also provided with the selection line SEL connected to the gate of the transistor T4 in addition to the boost line BST. Therefore, in this type pixel circuit, the voltage application timing to the boost line BST, and the turn-on timing of the T4 can be intentionally differentiated.

With this, in the case where the self-refreshing action is performed for the second type pixel circuits 2B and 2C, the voltage application timing to the selection line SEL may be delayed a little from the timing to apply the voltage to the reference line REF and the boost line BST.

As described above, as for the pixel having the gradation lower than the gradation serving as the refreshing target, the voltage that can turn on the T2 is applied to the reference line REF. Thus, even when the voltage is applied to the boost line BST in this state, the potential of the node N2 of the pixel is not boosted, and as a result, the transistor T1 is not turned on.

However, depending on an effect of another element such as an ability of the transistor or a parasitic capacitance of the node, even when the transistor T2 is on, the potential of the node N2 could be temporarily boosted when the voltage is applied to the boost line BST. In this case, the transistor T1 is turned on at that point, and as a result the pixel could be rewritten by the voltage having the different gradation.

Meanwhile, by delaying the turn-on timing of the transistor T4 a little from the voltage application timing to the boost line BST, even when the potential of the node N2 temporarily rises and the transistor T1 is on in this period, the transistor T4 is off, so that the source line SL and the node N1 cannot be connected by the transistor T4. In addition, even when the potential of the node N2 temporarily rises, the electric charge is absorbed into the parasitic capacitance of the node N1 after that, so that the potential of the node N2 falls. The transistor T1 is turned off at this time, so that even when the node T4 is turned on, the node N1 of the pixel circuit of the gradation lower than the refreshing target gradation is not rewritten by the voltage applied to the source line SL.

As described above, according to the second type pixel circuit especially, the voltage application timing to the selection line SEL can be controlled independently from the voltage application timing to the boost line BST, so that the error operation in which the wrong gradation is written can be surely prevented by delaying it a little from the application timing to the boost line BST.

This method can be applied to the timing chart shown in FIG. 25 in the third embodiment. That is, in FIG. 25, the voltage application timing to the selection line SE may be delayed a little from the time t3.

In addition, the refreshing action cannot be performed in the first type or the third type by this method, but probability the above error writing occurs is low from the beginning, so that the original gradation can be correctly restored by the refreshing action performed by the method described in the second embodiment.

<9> According to each of the above embodiments, the description has been given assuming that the pixel circuit has the second switch circuit 23 having one end connected to the source line SL and the other end connected to the internal node N1. However, as another configuration, even when a voltage supply line VSL is provided separately from the source line SL, and the second switch circuit 23 is connected to the voltage supply line VSL at one end in which the internal node N1 is not provided, the same action can be performed. Here, a voltage applied to the voltage supply line VSL is also controlled by the display control circuit 11 similar to the reference line REF and the boost line BST.

FIG. 37 shows one configuration example of the pixel circuit in this other embodiment. A pixel circuit 3A has a configuration in which one end of the second switch circuit 23 is connected to the voltage supply line VSL instead of being connected to the source line SL, compared with the pixel circuit 2A shown in FIG. 7. As for the pixel circuits 2A, 2B, 2C, 2D, and 2E shown in FIGS. 8 to 17, even when the one end of the second switch circuit 23 is connected to the voltage supply line VSL instead of being connected to the source line SL similarly, the same pixel circuit can be provided.

Thus, when the same voltage as that applied to the source line SL in each embodiment is applied to the voltage supply line VSL at the time of the self-refreshing action, the same voltage state as that in each embodiment can be provided. Thus, the self-refreshing action is executed for the pixel circuit in the other embodiment, based on all the same principle. In addition, since the transistor T3 is always off over the period of the self-refreshing action, the voltage applied to the source line SL has nothing to do with the self-refreshing action. In a view to cutting power consumption and excluding an effect of a leak current, the voltage applied to the source line SL is preferably set at 0 V over the period of the self-refreshing action. Its detailed description is omitted.

EXPLANATION OF REFERENCE

- 1: Liquid crystal display device
- 2: Pixel circuit
- 2A, 2B, 2C, 2D, 2E, 3A: Pixel circuit
- 10: Active matrix substrate
- 11: Display control circuit
- 12: Opposite electrode drive circuit
- 13: Source driver
- 14: Gate driver
- 20: Pixel electrode
- 21: Display element part
- 22: First switch circuit
- 23: Second switch circuit
- 24: Control circuit
- 74: Sealing material
- 75: Liquid crystal layer
- 80: Opposite electrode
- 81: Opposite substrate
- Amp: Analog amplifier
- BST: Boost line
- Cbst: Boost capacitive element
- Clc: Liquid crystal display element
- CML: Opposite electrode wiring
- CSL: Auxiliary capacity line
- Cs: Auxiliary capacitive element
- Ct: Timing signal
- D1: Diode element
- DA: Digital image signal
- Dv: Data signal
- GL (GL1, GL2, . . . , GLn): Gate line
- Gtc: Scan side timing control signal
- N1: Internal node
- N2: Output node
- OLED: Light emitting element
- P1, P2: Phase
- REF: Reference line
- S1, S2: Step
- Sc1, Sc2, . . . , Scm: Source signal
- SEL: Selection line
- SL (SL1, SL2, . . . , SLm): Source line
- Stc: Data side timing control signal
- T1, T2, T3, T4, T5: Transistor
- Tdv: Driving transistor
- Vcom: Opposite voltage
- Vlc: Liquid crystal voltage
- VN1: Internal node potential, Pixel electrode potential
- VN2: Output node potential

Number	Name	Date	Kind
20070128583	Miyazawa	Jun 2007	A1
20070222719	Wang et al.	Sep 2007	A1
20090027310	Kim	Jan 2009	A1
20090309816	Choi	Dec 2009	A1

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