The present disclosure claims priority to Japanese Patent Application No. 2019-022724 filed on Feb. 12, 2019, which is incorporated herein by reference in its entirety including specification, drawings and claims.
The present disclosure relates to a driving system.
A proposed configuration of a driving system includes a motor, an inverter configured to drive the motor, a smoothing capacitor mounted to a direct current side of the inverter, current sensors configured to detect electric currents of respective phases of the motor, and a voltage sensor configured to detect a voltage of the smoothing capacitor (as described in, for example, JP 2015-56919A). This driving system first specifies a target phase that is a phase of a second phase voltage command signal having a largest difference from a first phase voltage command signal having a signal level that is neither a maximum nor a minimum, based on three phase voltage command signals generated from the detected values of the current sensors. The driving system subsequently uses a BPF (band pass filter) to extract a voltage component of an identical frequency that is identical with a frequency of the three phase voltage command signals, from the detected value of the voltage sensor. The driving system then corrects the detected value of the current sensor with respect to the target phase, such that a voltage value of the extracted voltage component becomes equal to a desired voltage value.
While the motor is driven, not only the voltage component of the identical frequency (electrical first variation component) that is identical with the frequency of the three phase voltage command signals but an electrical second variation component, an electrical sixth variation component and the like are generated as variations in the voltage of the smoothing capacitor. It is, however, difficult to sufficiently remove the variation components of the orders other than the electrical first variation component by BPF. The driving system described above is thus likely to fail to adequately correct the detected value of the current sensor with respect to the target phase and thereby fail to sufficiently suppress a voltage variation of the smoothing capacitor and a torque variation of the motor.
A main object of a driving system of the present disclosure is to more effectively suppress a voltage variation of the smoothing capacitor and a torque variation of the motor.
In order to achieve the above primary object, the driving system of the present disclosure employs the following configuration.
The present disclosure is directed to a driving system, including: a motor; an inverter configured to drive the motor; a power storage device connected with the inverter via a power line; a smoothing capacitor mounted to the power line; a voltage sensor configured to detect a voltage of the smoothing capacitor; a current sensor configured to detect an electric current of each phase of the motor; and a control device configured to control the inverter, based on a detected value of the current sensor, wherein the control device performs Fourier series expansion of a detected value of the voltage sensor to calculate an electrical first variation component of the voltage of the smoothing capacitor, and the control device controls the inverter, such that the electrical first variation component of the voltage of the smoothing capacitor becomes equal to a value 0.
The driving system according to this aspect of the present disclosure performs Fourier series expansion of the detected value of the voltage sensor to calculate the electrical first variation component of the voltage of the smoothing capacitor and controls the inverter, such that the calculated electrical first variation component of the voltage of the smoothing capacitor becomes equal to the value 0. This configuration sufficiently removes components other than the electrical first variation component of the voltage of the smoothing capacitor (for example, electrical second variation component and electrical sixth variation component) and more effectively suppresses a voltage variation of the smoothing capacitor and a torque variation of the motor.
The following describes some aspects of the present disclosure with reference to several embodiments.
The motor 32 is configured as a synchronous generator motor and includes a rotor with permanent magnets embedded in a rotor core and a stator with three-phase coils wound on a stator core. The rotor of this motor 32 is connected with a driveshaft that is linked with drive wheels via a differential gear.
The inverter 34 is used to drive the motor 32. This inverter 34 is connected with the battery 36 via power lines 38 and includes six transistors T11 to T16 that serve as switching elements and six diodes D11 to D16 that are respectively connected in parallel with the six transistors T11 to T16. The transistors T11 to T16 are arranged in pairs, such that two transistors in each pair respectively serve as a source and as a sink relative to a positive bus bar and a negative bus bar of the power lines 38. The respective phases of the three-phase coils (U phase, V phase and W phase coils) of the motor 32 are connected with connection points of the respective pairs of the transistors T11 to T16. When a voltage is applied to the inverter 34, the electronic control unit 50 serves to regulate the rate of ON times of the respective pairs of the transistors T11 to T16 included in the inverter 34, such as to form a rotating magnetic field in the three-phase coils of the motor 32 and thereby rotate and drive the rotor of the motor 32.
The battery 36 is configured as, for example, a lithium ion rechargeable battery or a nickel metal hydride battery and is connected with the inverter 34 via the power lines 38 as described above. This battery 36 has an internal resistance 36r and an internal inductance 361. The smoothing capacitor 39 is mounted to the positive bus bar and the negative bus bar of the power lines 38.
The electronic control unit 50 is configured as a CPU 51-based microprocessor and includes a ROM 52 configured to store processing programs, a RAM 53 configured to temporarily store data, and input/output ports, in addition to the CPU 51. Signals from various sensors are input into the electronic control unit 50 via the input port. The signals input into the electronic control unit 50 include, for example, a rotational position θm of the rotor of the motor 32 from a rotational position detection sensor (for example, resolver) 32a configured to detect the rotational position of the rotor of the motor 32 and AD values (analog-to-digital converted voltage values) ADIV and ADIW corresponding to electric currents IV and IW of the V phase and the W phase of the motor 32 from current sensors 32v and 32w mounted to the V phase and the W phase of the motor 32. The input signals also include a voltage VH of the smoothing capacitor 39 (i.e., the power lines 38) from a voltage sensor 39a placed between terminals of the smoothing capacitor 39. The input signals further include an ignition signal from an ignition switch 60 and a shift position SP from a shift position sensor 62 configured to detect an operating position of a shift lever 61. The input signals also include an accelerator position Acc from an accelerator pedal position sensor 64 configured to detect a depression amount of an accelerator pedal 63, a brake pedal position BP from a brake pedal position sensor 66 configured to detect a depression amount of a brake pedal 65, and a vehicle speed V from a vehicle speed sensor 68.
Various control signals are output from the electronic control unit 50 via the output port. The signals output from the electronic control unit 50 include, for example, switching control signals to the transistors T11 to T16 included in the inverter 34. The electronic control unit 50 calculates an electrical angle θe, a mechanical angular velocity ωm an electrical angular velocity ωe and a rotation speed Nm of the motor 32, based on the rotational position θm of the rotor of the motor 32 input from the rotational position detection sensor 32a.
In the driving system 20 of the first embodiment configured as described above, the electronic control unit 50 sets a required torque Td* that is required for the driveshaft, based on the accelerator position Acc and the vehicle speed V, and sets a torque command Tm* of the motor 32, such that the set required torque Td* is output to the driveshaft. The electronic control unit 50 then performs switching control of the transistors T11 to T16 included in the inverter 34, such that the motor 32 is driven with the torque command Tm*. According to the first embodiment, the electronic control unit 50 controls the inverter 34 in a pulse width modulation (PWM) control mode.
The following describes operations of the driving system 20 according to the first embodiment having the configuration described above or more specifically a series of control of the inverter 34.
The electrical first variation component detector 71 serves to calculate a Fourier cosine coefficient aVH and a Fourier sine coefficient bVH as values relating to an electrical first variation component of the voltage VH of the smoothing capacitor 39, based on the electrical angle θe of the motor 32, a voltage phase ϕq with respect to a q axis of an output voltage of the inverter 34, and the voltage VH of the smoothing capacitor 39.
The inverter bus bar current estimator 72 serves to calculate a Fourier cosine coefficient aIm and a Fourier sine coefficient bIm as values relating to an electrical first variation component of a bus bar current (input current) Im of the inverter 34, based on the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39 calculated by the electrical first variation component detector 71 and the electrical angular velocity ωe of the motor 32.
The phase current offset estimator 73 serves to calculate current offset estimated values IVofs and IWofs of the V phase and the W phase, based on the Fourier cosine coefficient aIm and the Fourier sine coefficient bIm of the bus bar current Im of the inverter 34 calculated by the inverter bus bar current estimator 72 and a modulation degree Vr of the output voltage of the inverter 34.
The phase current offset controller 74 serves to calculate offset correction amounts ADVofs and ADWofs of the V phase and the W phase that are to be used by the phase current physical value calculator 76, based on the current offset estimated values IVofs and IVofs of the V phase and the W phase calculated by the phase current offset estimator 73.
The offset unit execution determiner 75 serves to determine whether the respective processes of the offset unit 70 (i.e., the processes of the electrical first variation component detector 71, the inverter bus bar current estimator 72, the phase current offset estimator 73 and the phase current offset controller 74) are to be performed, based on the electrical angular velocity ωe of the motor 32 and the modulation degree Vr of the output voltage of the inverter 34.
The phase current physical value calculator 76 serves to convert the AD values ADIV and ADIW input from the current sensors 32v and 32w into control currents IVcon and IWcon of the V phase and the W phase that are to be used by the current controller 77, by using the offset correction amounts ADVofs and ADWofs of the V phase and the W phase calculated by the phase current offset controller 74 or by setting both the offset correction amounts ADVofs and ADWofs of the V phase and the W phase to a value 0.
The current controller 77 serves to perform switching control of the transistors T11 to T16 included in the inverter 34, such that the motor 32 is driven with the torque command Tm*, based on the control currents IVcon and IWcon of the V phase and the W phase calculated by the phase current physical value calculator 76.
The following sequentially describes the electrical first variation component detector 71 to the current controller 77 more in detail. The electrical first variation component detector 71 is described first in detail.
When the electrical first variation component detecting process of
The electrical first variation component detector 71 subsequently updates a buffer index n by incrementing the buffer index n by a value 1 (step S110) and compares the updated buffer index n with the value N given above (step S120). When the buffer index n is smaller than the value N, the electrical first variation component detector 71 updates a sine component buffer VHSIN[n] and a cosine component buffer VHCOS[n] of the voltage VH of the smoothing capacitor 39 with regard to the buffer index n according to Expression (1-1) and Expression (1-2) given below by using the voltage VH of the smoothing capacitor 39, the electrical angle θe of the motor 32 and the voltage phase ϕq of the output voltage of the inverter 34 (steps S140 and S150).
VH SIN[n]←VH×sin(θe+φq+180°) (1-1)
VH COS[n]←VH×cos(θe+φq+180°) (1-2)
When the buffer index n is equal to or larger than the value N at step S120, on the other hand, the electrical first variation component detector 71 resets the buffer index n to a value 0 (step S130) and updates the sine component buffer VHSIN[n] and the cosine component buffer VHCOS[n] of the voltage VH of the smoothing capacitor 39 with regard to the buffer index n according to Expression (1-1) and Expression (1-2) given above (steps S140 and S150).
The electrical first variation component detector 71 subsequently calculates a Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39 according to Expression (1-3) given below by using the sine component buffer VHSIN[i] (where i=0, . . . , N−1) of the voltage VH of the smoothing capacitor 39 (step S160). The electrical first variation component detector 71 also calculates a Fourier cosine coefficient aVH of the voltage VH of the smoothing capacitor 39 according to Expression (1-4) given below by using the cosine component buffer VHCOS[i] of the voltage VH of the smoothing capacitor 39 (step S170) and then terminates the electrical first variation component detecting process of
The sine component buffer VHSIN[n] and the cosine component buffer VHCOS[n] of the voltage VH of the smoothing capacitor 39, the Fourier sine coefficient bVH and the Fourier cosine coefficient aVH are values relating to the electrical first variation component of the voltage VH of the smoothing capacitor 39. The following describes a method of deriving Expressions (1-1) to (1-4) given above. The definition of Fourier series expansion is described first. A periodic function f(x) having a period of 2n is expanded as shown by Equation (1-5) given below and is decomposable into frequency components of integral multiples of the original frequency. In Equation (1-5), “an” and “bn” are respectively expressed by Equation (1-6) and Equation (1-7) given below.
Extraction of the electrical first variation component of the voltage VH of the smoothing capacitor 39 is described next. The electrical first variation component of the voltage VH of the smoothing capacitor 39 is expressed by Equation (1-8) given below. In Equation (1-8), “aVH” and “bVH” are respectively referred to as a Fourier cosine coefficient and a Fourier sine coefficient of the voltage VH. The Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH are respectively expressed by Equation (1-9) and Equation (1-10) given below by substituting a value 1 into “n” of Equation (1-6) and Equation (1-7) described above, replacing “dt” with “dθ” and replacing “f(t) with “VH(θ)”.
Equation (1-11) and Equation (1-12) given below are obtained by changing the angle from “rad” to “deg” and discretizing Equation (1-9) and Equation (1-10) given above with respect to divisions of 360 degrees/N in the electrical angle θe of the motor 32. Expressions (1-1) to (1-4) are derived from these Equations (1-11) and (1-12). By taking into account the foregoing, the electrical first variation component detector 71 (i.e., the electrical first variation component detecting process of
As the result of experiments and analyses, the inventors of the present disclosure have found that a torque variation of the motor 32 can be reduced by extracting only the electrical first variation component of the voltage VH of the smoothing capacitor 39 and driving the motor 32, such that the electrical first variation component of the voltage VH of the smoothing capacitor 39 is decreased (or more preferably is made equal to zero) by decreasing offset amounts of electric currents IU, IV and IW of the respective phases of the motor 32 (or more preferably decreasing the offset amounts to zero), based on a relationship between the electrical first variation component of the voltage VH of the smoothing capacitor 39 and the offset amounts of electric currents IU, IV and IW of the respective phases of the motor 32. In general, when the motor 32 is driven, variation components of the orders other than the electrical first order (for example, electrical second order and electrical sixth order) are also generated as variation components of the voltage VH of the smoothing capacitor 39, as shown in
A technique employable to remove the components of the orders other than the electrical first order out of the variation components of the voltage VH of the smoothing capacitor 39 may use a band pass filter (BPF) as shown by Equation (1-13) given below. In this Equation (1-13), “ωe” denotes an electrical angular velocity of the motor 32, “s” represents the Laplacian operator, and “ξ” represents a constant designed to obtain a desired band width.
The following describes the inverter bus bar current estimator 72 in detail.
When the inverter bus bar current estimating process of
After obtaining the input data, the inverter bus bar current estimator 72 sets an amplitude ratio A and a phase difference Δθ between the bus bar electric current (electric current on the DC side as shown in
The inverter bus bar current estimator 72 subsequently calculates the Fourier cosine coefficient aIm and the Fourier sine coefficient bIm of the bus bar current Im of the inverter 34 according to Equation (2-1) and Equation (2-2) given below by using the amplitude A, the phase difference Δθ, and the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH (steps S220 and S230) and then terminates the inverter bus bar current estimating process of
a
Im
=A×(aVH cos Δθ+bVH sin Δθ) (2-1)
b
Im
=A×(bVH cos Δθ−aVH sin Δθ) (2-2)
The Fourier cosine coefficient aIm and the Fourier sine coefficient bIm of the bus bar current Im of the inverter 34 are values relating to the electrical first variation component of the bus bar current Im of the inverter 34 (shown in
For this conversion, the procedure of the first embodiment calculates a transfer function Im/VH with regard to a circuit shown in
The following describes a method of deriving Equation (2-1) and Equation (2-2) given above. The electrical first variation component of the voltage VH of the smoothing capacitor 39 is expressed by Equation (1-8) given above. According to the frequency characteristics of the transfer function Im/VH (expressed by Equation (2-3) given above), adding corrections of the amplitude ratio A and the phase difference Δθ to the variation in the voltage VH of the smoothing capacitor 39 provides a variation in the bus bar current Im of the inverter 34. Accordingly, an electrical first variation component Im1 of the bus bar current Im of the inverter 34 is expressed by Equation (2-4) given below.
I
m1
=A×a
VH cos(θ+Δθ)+A×bVH sin(θ+Δθ) (2-4)
Equation (2-5) given below is obtained by applying the addition theorem to Equation (2-4) given above and reorganizing Equation (2-4). A coefficient for cos θ in the first term on the right side of this Equation (2-5) is obtained as the Fourier cosine coefficient aIm of the bus bar current Im of the inverter 34 as shown by Equation (2-1) given above. A coefficient for sin θ, in the second term on the right side of Equation (2-5) is obtained as the Fourier sine coefficient bIm of the bus bar current Im of the inverter 34 as shown by Equation (2-2) given above. Accordingly, Equation (2-5) is rewritten as Equation (2-6) given below. By taking into account the foregoing, the inverter bus bar current estimator 72 (i.e., the inverter bus bar current estimating process of
I
m1
=A×(aVH cos Δθ+bVH sin Δθ)cos θ+A×(bVH cos Δθ−aVH sin Δθ)sin θ (2-5)
I
m1
=a
Im cos θ+bIm sin θ (2-6)
The following describes the phase current offset estimator 73 in detail.
When the phase current offset estimating process of
After obtaining the input data, the phase current offset estimator 73 calculates the current offset estimated values IWofs and IWofs of the V phase and the W phase according to Equation (3-1) and Equation (3-2) given below by using the input Fourier cosine coefficient aIm and the input Fourier sine coefficient bIm of the bus bar current Im of the inverter 34 and the input modulation degree Vr of the output voltage of the inverter 34 (steps S310 and S320) and then terminates the phase current offset estimating process of
Equation (3-1) and Equation (3-2) given above are equations of converting the electrical first variation components of the bus bar current Im of the inverter 34 into the current offset estimated values IVofs and IWofs of the V phase and the W phase. The following describes a method of driving Equation (3-1) and Equation (3-2). First, the electric currents IU, IV and IW of the respective phases are expressed by Equations (3-3) to (3-5) given below on the assumption of generation of offsets in the electric currents IU, IV and IW of the respective phases. In Equations (3-3) to (3-5), “I1” denotes a maximum value of a fundamental wave component of a three-phase AC current and is determined according to the specification of the motor 32, and “β” denotes a current phase on the basis of the U phase (IU=0 when β=0).
I
U
=I
Uofs
+I
1 sin(θe+β) (3-3)
I
V
=I
Vofs
+I
1 sin(θe+β−120°) (3-4)
I
W
=I
Wofs
+I
1 sin(θe+β+120°) (3-5)
Under the condition of balanced impedance of loads, the voltages applied to the loads differ due to the offset electric currents. Accordingly, offsets are generated in voltages VU, VV and VW of the respective phases. The voltages VU, VV and VW of the respective phases under this condition are expressed by Equations (3-6) to (3-8) given below. In Equations (3-6) to (3-8), “V1” denotes a maximum value of a fundamental wave component of a three-phase AC voltage and is determined according to the specification of the motor 32, and “α” denotes a voltage phase on the basis of the U phase (VU=0 when α=0).
V
U
=V
Uofs sin(θe+α) (3-6)
V
V
=V
Vofs
+V
1 sin(θe+α−120°) (3-7)
V
W
=V
Wofs+sin(θe+α+120°) (3-8)
The electric power is the product of the electric current and the voltage, so that a sum of electric powers PU, PV and PW of the respective phases of the motor 32 (i.e., the electric power of the motor 32) is expressed by Equation (3-9) given below and thereby by Equation (3-10) given below. The second term and the third term of Equation (3-10) may be regarded as an electrical first variation component P1 of the sum of the electric powers PU, PV and PW of the respective phases of the motor 32.
P
U
+P
V
+P
W
=V
U
I
U
+V
V
I
V
+V
W
I
W (3-9)
P
U
+P
V
+P
W=3/2V1I1 cos(α−β)+{V1IUofs sin(θe+α)+V1IVofs sin(θe+α−120°)+V1IWofs sin(θe+α+120°)}+{VUofsI1 sin(θe+β)+VVofsI1 sin(θe+β−120°)+VWofsI1 sin(θe+β+120°)}+{VUofsIUofs+VVofsIVofs+VWofsIWofs}(3-10)
When the electrical first variation component P1 of the sum of the electric powers PU, PV and PW of the respective phases of the motor 32 is divided into a component PV1 that is in synchronism with the voltage phase and a component PI1 that is in synchronism with the current phase as shown by Equation (3-11) given below (i.e., when the electrical first variation component P1 is regarded as the sum of the component PV1 and the component PI1), the component PV1 that is in synchronism with the voltage phase is expressed by Equation (3-12) given below and the component PI1 that is in synchronism with the current phase is expressed by Equation (3-13) given below.
P
1
=P
V1
+P
I1 (3-11)
P
V1
=V
1
I
Uofs sin(θe+α)+V1IVofs sin(θe+α−120°)+V1IWofs sin(θe+α+120°) (3-12)
P
I1
=V
Uofs
I
1 sin(θe+β)+VVofsI1 sin(θe+β−120°)+VWofsI1 sin(θe+β+120°) (3-13)
A value V1IUofs of the component PV1 and a value VUofsI1 of the component PI1 are compared with each other, in order to specify whether the component PV1 that is in synchronism with the voltage phase or the component PI1 that is in synchronism with the current phase is dominant. Expression (3-14) given below is expected to hold as a condition that the component PV1 that is in synchronism with the voltage phase becomes greater than the component PI1 that is in synchronism with the current phase. Under the condition of balanced impedance of loads of the three phases, Equation (3-15) given below is expected to hold by using resistance values R of the respective phases of the motor 32. Accordingly, Expression (3-16) and thereby Expression (3-17) given below are obtained from Expression (3-14). A value V1 is generally expressed by Equation (3-18) given below. Expression (3-19) given below is accordingly obtained from Expression (3-17) and Equation (3-18).
Accordingly, when the modulation degree Vr of the output voltage of the inverter 34 is equal to or greater than a certain value, the component PV1 that is in synchronism with the voltage phase out of the electrical first variation component P1 of the sum of the electric powers PU, PV and PW of the respective phases of the motor 32 is dominant. On the assumption that the left side of Expression (3-19) is sufficiently larger than the right side of Expression (3-19), Expression (3-20) given below holds. Equation (3-21) given below is obtained by rewriting Expression (3-19) on the assumption of the sum of the electric currents of the respective phases is equal to 0 (IUofs=−IVofs−IWofs).
P
1
=P
V1
=V
1
I
Uofs sin(θe+α)+V1IVofs sin(θe+α−120°)+V1IWofs sin(θe+α+120°) (3-20)
P
1=√{square root over (3)}V1IVofs sin(θe+α−150°)+√{square root over (3)}V1IWofs sin(θe+α−210°) (3-21)
The electrical first variation component P1 of the sum of the electric powers PU, PV and PW of the respective phases of the motor 32 is converted into an electrical first variation component Im1 of the bus bar current Im of the inverter 34 according to Equation (3-22) given below by using the voltage VH of the smoothing capacitor 39. Equation (3-23) given below is obtained by substituting Equation (3-18) into Equation (3-22).
Under the condition of “θe+α=150°”, Equation (3-25) and Equation (3-26) given below are obtained from this Equation (3-23) and Equation (3-24) given below, which is obtained on the basis of Equation (2-6) given above, so that Equation (3-2) given above is obtained. Under the condition of “θe+α=210°”, on the other hand, Equation (3-27) and Equation (3-28) given below are obtained from Equation (3-23) and Equation (3-24), so that Equation (3-1) given above is obtained. By taking into account the foregoing, the phase current offset estimator 73 (i.e., the phase current offset estimating process of
I
m1
=a
Im cos(θe+α)+bIm sin(θe+α) (3-24)
I
m1(150°)=√{square root over (2)}VrIWofs sin(−60°) (3-25)
I
m1(150°)=aIm cos(150°)+bIm sin(150°) (3-26)
I
m1(210°)=√{square root over (2)}VrIVofs sin(60°) (3-27)
I
m1(210°)=aIm cos(210°)+bIm sin(210°) (3-28)
The value “θe+α” means the sum of the electrical angle of the motor 32 and the voltage phase that provides the voltage Vu of the U phase equal to a value 0. The motor 32 is generally controlled by using the d axis and the q axis. An equation of “θe+α=θe+ϕq+180°” is given by conversion of this voltage phase into a voltage phase ϕq on the basis of the q axis. By taking into account the foregoing, according to the first embodiment, the electrical first variation component detector 71 (more specifically, the processing of steps S140 and S150 in the electrical first variation component detecting process of
The following describes the phase current offset controller 74 in detail.
When the phase current offset control process of
After obtaining the input data, the phase current offset controller 74 calculates the offset correction amount ADVofs of the V phase according to Expression (4-1) by using the input current offset estimated value IVofs of the V phase (step S410), calculates the offset correction amount ADWofs of the W phase according to Expression (4-2) by using the input current offset estimated value IWofs of the W phase (step S420) and then terminates the phase current offset control process of
AD
Vofs
←K
P(0−IVofs)+KI∫(0−IVofs)dt (4-1)
AD
Wofs
←K
P(0−IWofs)+KI∫(0−IWofs)dt (4-2)
The offset correction amounts ADVofs and ADWofs of the V phase and the W phase denote correction amounts respectively used to convert the AD values ADIV and ADIW input from the current sensors 32v and 32w of the V phase and the W phase into the control currents IVcon and IWcon of the V phase and the W phase by the phase current physical value calculator 76. Expression (4-1) and Expression (4-2) are relational expressions of feedback control respectively used to calculate the offset correction amounts ADVofs and ADWofs, such that the current offset estimated values IWofs and IWofs of the V phase and the W phase become equal to a value 0. In Expression (4-1) and Expression (4-2), “KP” denotes a gain of a proportional, and “KI” denotes a gain of an integral term.
The following describes the offset unit execution determiner 75 in detail.
When the offset unit execution determining process of
After obtaining the input data, the offset unit execution determiner 75 compares the input electrical angular velocity ωe of the motor 32 with a reference value ωeref (step S510) and also compares the input modulation degree Vr of the output voltage of the inverter 34 with a reference value Vrref (step S520). The reference value ωeref and the reference value Vrref are threshold values used to determine whether the respective processes of the offset unit 70 (i.e., the processes of the electrical first variation component detector 71, the inverter bus bar current estimator 72, the phase current offset estimator 73 and the phase current offset controller 74) are to be performed.
The processing of step S510 is described in detail. In order to perform the respective processes of the offset unit 70, there is a need for detecting the electrical first variation component of the voltage VH of the smoothing capacitor 39 (as shown in
The processing of step S520 is described in detail. The phase current offset estimator 73 derives Equations used for the processing of steps S310 and S320 (Equations (3-1) and (3-2)) in the phase current offset estimating process of
When it is determined at step S510 that the electrical angular velocity ωe of the motor 32 is equal to or larger than the reference value ωeref and it is determined at step S520 that the modulation degree Vr of the output voltage of the inverter 34 is equal to or larger than the reference value Vrref, the offset unit execution determiner 75 determines that the respective processes of the offset unit 70 (i.e., the processes of the electrical first variation component detector 71, the inverter bus bar current estimator 72, the phase current offset estimator 73 and the phase current offset controller 74) are to be performed (step S530) and then terminates the offset unit execution determining process of
When it is determined at step S510 that the electrical angular velocity ωe of the motor 32 is smaller than the reference value ωeref or when it is determined at step S520 that the modulation degree Vr of the output voltage of the inverter 34 is smaller than the reference value Vrref, on the other hand, the offset unit execution determiner 75 determines that the respective processes of the offset unit 70 are not to be preformed (step S540) and then terminates the offset unit execution determining process of
The following describes the phase current physical value calculator 76 in detail.
When the phase current physical value calculating process of
When it is determined that the respective processes of the offset unit 70 are being performed, the phase current physical value calculator 76 obtains the input data of the offset correction amounts ADVofs and ADWofs of the V phase and the W phase (step S620). The offset correction amounts ADVofs and ADWofs of the V phase and the W phase input here are values calculated by the phase current offset controller 74 (i.e., the phase current offset control process of
The phase current physical value calculator 76 subsequently converts the AD value ADIV into the control current IVcon of the V phase according to Equation (5-1) given below by using the offset correction amount ADVofs of the V phase (step S640), converts the AD value ADIW into the control current IWcon of the W phase according to Equation (5-2) given below by using the offset correction amount ADWofs of the W phase (step S650) and then terminates the phase current physical value calculating process of
I
Vcon=(ADIV−(2.5V+ADVofs))×ADVgain (5-1)
I
Wcon=(ADIW−(2.5V+ADWofs))×ADWgain (5-2)
Converting the AD values ADIV and ADIW into the control currents IVcon and IWcon of the V phase and the W phase by using the offset correction amounts ADVofs and ADWofs of the V phase and the W phase temporarily change the offset amounts of the control currents IVcon and IWcon of the V phase and the W phase. The current controller 77, however, performs control, such that the offset amounts of the control currents IVcon and IWcon of the V phase and the W phase become equal to a value 0 as described later. The offset amounts of the electric currents IV and IW of the V phase and the W phase (actual offset amounts) are thereby varied according to the offset correction amounts ADVofs and ADWofs of the V phase and the W phase.
When it is determined at step S610 that the respective processes of the offset unit 70 are not being performed, on the other hand, the phase current physical value calculator 76 sets previous offset correction amounts (previous ADWofs) and (previous ADWofs) of the V phase and the W phase respectively to the offset correction amounts ADWofs and ADWofs of the V phase and the W phase (step S630), converts the AD values ADIV and ADIW into the control currents IVcon and IWcon of the V phase and the W phase by using the offset correction amounts ADVofs and ADWofs of the V phase and the W phase (both equal to a value 0) by the processing of steps S640 and S650 described above and then terminates the phase current physical value calculating process of
The following describes the current controller 77 in detail.
The low pass filter 81 serves to generate a filtered torque command Tmf* by low pass filter processing of the torque command Tm* of the motor 32. The current command generator 82 serves to generate current commands Id* and Iq* of the d axis and the q axis by applying the filtered torque command Tmf* to a map that specifies a relationship of the filtered torque command Tmf* to current commands Id* and Iq* of the d axis and the q axis. The coordinate converter 83 serves to perform coordinate conversion (three phase to two phase conversion) of the control currents IVcon and IWcon of the V phase and the W phase of the motor 32 into electric currents Id and Iq of the d axis and the q axis by using the electrical angle θe of the motor 32 on the assumption that the sum of the electric currents flowing through the respective phases is equal to 0.
The subtractors 84d and 84q serve to calculate differences ΔId and ΔIq between the current commands Id* and Iq* and the electric currents Id and Iq of the d axis and the q axis. The feedback controllers (PI controllers) 85d and 85q serve to calculate voltage commands Vd* and Vq* of the d axis and the q axis by current feedback control such that the differences ΔId and ΔIq become equal to a value 0.
The coordinate converter 86 serves to perform coordinate conversion (two phase to three phase conversion) of the voltage commands Vd* and Vq* of the d axis and the q axis into voltage commands VU*, VV* and VW* of the respective phases. The PWM signal generator 87 serves to generate a PWM signal of the transistors T11 to T16 by using a triangular wave and the voltage commands VU*, VV* and VW* of the respective phases and perform switching control of the transistors T11 to T16. Offset amounts of the control currents IVcon and IWcon of the V phase and the W phase (the electric currents of the V phase and the W phase recognized by the electronic control unit 50) are provided as electrical first variation components of the electric currents Id and Iq of the d axis and the q axis by the coordinate conversion (three phase to two phase conversion). In the case where the motor 32 has sufficiently good responsibility in the current feedback control, the inverter 34 is controlled such that the electric currents Id and Iq of the d axis and the q axis become equal to the current commands Id* and Iq*. This sufficiently decrease the electrical first variations of the electric currents Id and Iq of the d axis and the q axis (ideally make the electrical first variations equal to zero).
The voltage phase calculator 88 serves to calculate the voltage phase ϕq with respect to the q axis of the output voltage of the inverter 34 according to Equation (6-1) given below by using the voltage commands Vd* and Vq* of the d axis and the q axis. The modulation degree calculator 89 serves to calculate the modulation degree Vr of the output voltage of the inverter 34 according to Equation (6-2) given below by using the voltage commands Vd* and Vq* of the d axis and the q axis and the voltage VH of the smoothing capacitor 39.
As described above, the driving system 20 of the first embodiment performs Fourier series expansion of the voltage VH of the smoothing capacitor 39 input from the voltage sensor 39a to calculate the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH, as the values relating to the electrical first variation component of the voltage VH of the smoothing capacitor 39. The driving system 20 of the first embodiment subsequently calculates the Fourier cosine coefficient aIm and the Fourier sine coefficient bIm, as the values relating to the electrical first variation component of the bus bar current (input current) Im of the inverter 34, based on the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39. The driving system 20 of the first embodiment subsequently calculates the current offset estimated values IVofs and IWofs of the V phase and the W phase, based on the Fourier cosine coefficient aIm and the Fourier sine coefficient bIm of the bus bar current Im of the inverter 34, and calculates the offset correction amounts ADVofs and ADWofs of the V phase and the W phase, such that the current offset estimated values IVofs and IWofs of the V phase and the W phase become equal to the value 0. The driving system 20 of the first embodiment also converts the AD values ADD/and ADIW corresponding to the electric currents IV and IW of the V phase and the W phase of the motor 32 input from the current sensors 32v and 32w into the control currents IVcon and IWcon of the V phase and the W phase accompanied with correction using the offset correction amounts ADVofs and ADWofs of the V phase and the W phase. The driving system 20 of the first embodiment then performs switching control of the transistors T11 to T16 included in the inverter 34, such that the motor 32 is driven with the torque command Tm*, based on the control currents IVcon and IWcon of the V phase and the W phase obtained by the conversion. The configuration of the first embodiment accordingly performs Fourier series expansion of the voltage VH of the smoothing capacitor 39 to extract the electrical first variation component of the voltage VH of the smoothing capacitor 39. This configuration thus sufficiently removes the components other than the electrical first variation component of the voltage VH of the smoothing capacitor 39 (for example, electrical second variation component and electrical sixth variation component) and more effectively suppresses a voltage variation of the smoothing capacitor 39 and a torque variation of the motor 32.
The following describes a driving system 120 according to a second embodiment.
The following describes a series of control of the inverter 34 performed by the electronic control unit 50 in the driving system 120 of the second embodiment.
The electrical first variation component detector 171 is described first.
When the electrical first variation component detecting process of
After calculating the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39 (steps S160 and S170), the electrical first variation component detector 171 updates a sine component buffer IBSIN[n] and a cosine component buffer IBCOS[n] of the electric current IB of the battery 36 according to Expression (7-1) and Expression (7-2) given below by using the electric current IB of the battery 36, the electrical angle θe of the motor 32 and the voltage phase ϕq of the output voltage of the inverter 34 (steps S172b and S174b).
IB SIN[n]←IB×sin(θe+φq+180°) (7-1)
IB COS[n]←IB×cos(θe+φq+180°) (7-2)
The electrical first variation component detector 171 subsequently calculates a Fourier sine coefficient bIB of the electric current IB of the battery 36 according to Expression (7-3) given below by using the sine component buffer IBSIN[i] (where i=0, . . . , N−1) of the electric current IB of the battery 36 (step S176b), calculates a Fourier cosine coefficient aIB of the electric current IB of the battery 36 according to Expression (7-4) given below by using the cosine component buffer IBCON[i] of the electric current IB of the battery 36 (step S178b) and then terminates the electrical first variation component detecting process of
The inverter bus bar current estimator 172 is described next.
After obtaining the input data, the inverter bus bar current estimator 172 subsequently calculates the Fourier cosine coefficient aIm of the bus bar current Im of the inverter 34 according to Equation (7-5) given below by using the input Fourier cosine coefficient aIB of the electric current IB of the battery 36, the input electrical angular velocity ωe of the motor 32 and the input Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39 (step S220b). The inverter bus bar current estimator 172 also calculates the Fourier sine coefficient bIm of the bus bar current Im of the inverter 34 according to Equation (7-6) given below by using the input Fourier sine coefficient bIB of the electric current IB of the battery 36, the input electrical angular velocity ωe of the motor 32 and the input Fourier cosine coefficient aVH of the voltage VH of the smoothing capacitor 39 (step S230b) and then terminates the inverter bus bar current estimating process of
a
Im
=a
IB−ωeCHbVH (7-5)
b
Im
=b
IB+ωeCHaVH (7-6)
The following describes a method of deriving Equation (7-5) and Equation (7-6) given above. Equation (7-7) given below holds between the electric current IB of the battery 36, the electric current Ic of the smoothing capacitor 39 and the bus bar current Im of the inverter 34 on the periphery of the smoothing capacitor 39. Equation (7-8) given below also holds between the electric current Ic and the voltage VH of the smoothing capacitor 39 by using the capacity value CH of the smoothing capacitor 39.
An electrical first variation component VH1 of the voltage VH of the smoothing capacitor 39 is expressed by Equation (7-9) given below. Accordingly, when the attention is focused on only the electrical first variation component VH1 with regard to the voltage VH of the smoothing capacitor 39, an electrical first variation component Ic1 of the electric current Ic of the smoothing capacitor 39 is expressed by Equation (7-10) given below, based on Equations (7-8) and (7-9). On definition of Equation (7-11) given below, Equation (7-12) given below is derived from Equation (7-10).
An electrical first variation component IB1 of the electric current Is of the battery 36 is expressed by Equation (7-13) given below. Accordingly, when the attention is focused on only the electrical first variation components IB1 and VH1 with regard to the electric current IB of the battery 36 and the voltage VH of the smoothing capacitor 39, the electrical first variation component Im1 of the bus bar current Im of the inverter 34 is expressed by Equation (7-14) given below, based on Equations (7-12) and (7-13).
I
B1
=a
IB cos θ+bIB sin θ (7-13)
I
m1
=a
IB cos θ+bIB sin θ−ωeCH(bVH cos θ−aVH sin θ) (7-14)
Equation (7-15) given below is derived by rewriting Equation (7-14). A coefficient for cos θ in the first term on the right side of this Equation (7-15) is obtained as the Fourier cosine coefficient aIm of the bus bar current Im of the inverter 34 as shown by Equation (7-5) given above. A coefficient for sine in the second term on the right side of Equation (7-15) is obtained as the Fourier sine coefficient bIm of the bus bar current Im of the inverter 34 as shown by Equation (7-6) given above. Accordingly, Equation (7-15) is rewritten as Equation (7-16) given below. By taking into account the foregoing, the inverter bus bar current estimator 172 (i.e., the inverter bus bar current estimating process of
I
m1=(aIB−ωeCHbVH)cos θ+(bIB+ωeCHaVH)sin θ (7-15)
I
m1
=a
Im cos θ+bIm sin θ (7-16)
Calculating the Fourier cosine coefficient aIm and the Fourier sine coefficient bIm of the bus bar current Im of the inverter 34 by using the electric current IB of the battery 36 (more specifically, the Fourier cosine coefficient am and the Fourier sine coefficient bIB of the electric current IB of the battery 36) as described above enables the Fourier cosine coefficient aIm and the Fourier sine coefficient bIm of the bus bar current Im of the inverter 34 to be calculated without using the resistance value RB of the internal resistance 36r and the inductance value LB of the internal inductance 361 of the battery 36, i.e., without being affected by variations in the resistance value RB and the inductance value LB.
As described above, like the driving system 20 of the first embodiment, the driving system 120 of the second embodiment performs Fourier series expansion of the voltage VH of the smoothing capacitor 39 input from the voltage sensor 39a to calculate the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39, and controls the inverter 34, based on the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39. Like the configuration of the driving system 20 of the first embodiment, the configuration of the driving system 120 of the second embodiment sufficiently removes the components other than the electrical first variation component of the voltage VH of the smoothing capacitor 39 (for example, electrical second variation component and electrical sixth variation component) and more effectively suppresses a voltage variation of the smoothing capacitor 39 and a torque variation of the motor 32.
Furthermore, the driving system 120 of the second embodiment calculates the Fourier cosine coefficient aIm and the Fourier sine coefficient bIm of the bus bar current Im of the inverter 34, based on the Fourier cosine coefficient aIB and the Fourier sine coefficient bIB of the electric current IB of the battery 36. This configuration enables the Fourier cosine coefficient aIm and the Fourier sine coefficient bIm of the bus bar current Im of the inverter 34 to be calculated without using the resistance value RB of the internal resistance 36r and the inductance value LB of the internal inductance 361 of the battery 36, i.e., without being affected by variations in the resistance value RB and the inductance value LB.
The following describes a driving system 220 according to a third embodiment.
The boost converter 240 is connected with the inverter 34 via the high voltage-side power lines 38a and is also connected with the battery 36 via the low voltage-side power lines 38b. This boost converter 240 includes two transistors T31 and T32 that serve as switching elements, two diodes D31 and D32 that are respectively connected in parallel with the two transistors T31 and T32, and a reactor 242. The transistor T31 is connected with a positive electrode line of the high voltage-side power lines 38a. The transistor T32 is connected with the transistor T31 and with negative electrode lines of the high voltage-side power lines 38a and of the low voltage-side power lines 38b. The reactor 242 is connected with a connection point between the transistors T31 and T32 and with a positive electrode line of the low voltage-side power lines 38b. This reactor 242 has a resistance component 242r and an inductance component 2421. In response to regulation of a ratio of ON times of the transistors T31 and T32 by the electronic control unit 50, the boost converter 240 serves to step up the voltage of electric power of the low voltage-side power lines 38b and supply the electric power of the stepped-up voltage to the high voltage-side power lines 38a and to step down the voltage of electric power of the high voltage-side power lines 38a and supply the electric power of the stepped-down voltage to the low voltage-side power lines 38b. The smoothing capacitor 246 is mounted to a positive bus bar and a negative bus bar of the low voltage-side power lines 38b.
Like in the driving system 20 of the first embodiment described above, in the driving system 220 of this embodiment, the electronic control unit 50 sets the required torque Td* that is required for the driveshaft, based on the accelerator position Acc and the vehicle speed V, sets the torque command Tm* of the motor 32, such that the set required torque Td* is output to the driveshaft, and performs switching control of the transistors T11 to T16 included in the inverter 34, such that the motor 32 is driven with the torque command Tm*. The electronic control unit 50 also sets a target voltage VH* of the smoothing capacitor 39 (high voltage-side power lines 38a), such that the motor 32 is driven with the torque command Tm*, sets a duty command D such that the voltage VH of the smoothing capacitor 39 becomes equal to a target voltage VH*, and performs switching control of the transistors T31 and T32 included in the boost converter 240 by using the set duty command D. The duty command D is defined as a product of a rate of the ON time of the transistor T31 to the sum of the ON time of the transistor T31 and the ON time of the transistor T32, and a value 100.
The following describes a series of control of the inverter 34 performed by the electronic control unit 50 in the driving system 220 of the third embodiment.
The inverter bus bar current estimator 272 is described first. Like the inverter bus bar current estimator 72, the inverter bus bar current estimator 272 performs the inverter bus bar current estimating process of
The offset unit execution determiner 275 is described next.
When the offset unit execution determining process of
After obtaining the input data, the offset unit execution determiner 275 compares the input electrical angular velocity ωe of the motor 32 with the reference value ωeref (step S510), compares the input modulation degree Vr of the output voltage of the inverter 34 with the reference value Vrref (step S520), and subsequently determines whether the duty command D is equal to 100% (step S522c).
When it is determined at step S510 that the electrical angular velocity ωe of the motor 32 is equal to or larger than the reference value ωeref, it is determined at step S520 that the modulation degree Vr of the output voltage of the inverter 34 is equal to or larger than the reference value Vrref, and it is determined at step S522c that the duty command D is equal to 100%, the offset unit execution determiner 275 determines that the respective processes of the offset unit 270 (i.e., the processes of the electrical first variation component detector 71, the inverter bus bar current estimator 272, the phase current offset estimator 73 and the phase current offset controller 74) are to be performed (step S530) and then terminates the offset unit execution determining process of
When it is determined at step S510 that the electrical angular velocity ωe of the motor 32 is smaller than the reference value ωeref, when it is determined at step S520 that the modulation degree Vr of the output voltage of the inverter 34 is smaller than the reference value Vrref or when it is determined at step S522c that the duty command D is not equal to 100%, on the other hand, the offset unit execution determiner 275 determines that the respective processes of the offset unit 270 are not to be preformed (step S540) and then terminates the offset unit execution determining process of
The processing of step S522c is performed, since the inverter bus bar current estimator 272 can appropriately set the amplitude ratio A and the phase difference Δθ by using the map of
As described above, like the driving system 20 of the first embodiment, the driving system 220 of the third embodiment performs Fourier series expansion of the voltage VH of the smoothing capacitor 39 input from the voltage sensor 39a to calculate the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39, and controls the inverter 34, based on the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39. Like the configuration of the driving system 20 of the first embodiment, the configuration of the driving system 220 of the third embodiment sufficiently removes the components other than the electrical first variation component of the voltage VH of the smoothing capacitor 39 (for example, electrical second variation component and electrical sixth variation component) and more effectively suppresses a voltage variation of the smoothing capacitor 39 and a torque variation of the motor 32.
The following describes a driving system 320 according to a fourth embodiment.
The following describes a series of control of the inverter 34 performed by the electronic control unit 50 in the driving system 320 of the fourth embodiment.
The electrical first variation component detector 371 is described first.
When the electrical first variation component detecting process of
After calculating the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39 (steps S160 and S170), the electrical first variation component detector 371 updates a sine component buffer ILSIN[n] and a cosine component buffer ILCOS[n] of the electric current IL of the reactor 242 according to Expression (9-1) and Expression (9-2) given below by using the electric current IL of the reactor 242, the electrical angle θe of the motor 32 and the voltage phase ϕq of the output voltage of the inverter 34 (steps S172d and S174d).
IL SIN[n]←IL×sin(θe+φq+180°) (9-1)
IL COS[n]←IL×cos(θe+φq+180°) (9-2)
The electrical first variation component detector 371 subsequently calculates a Fourier sine coefficient bIL of the electric current IL of the reactor 242 according to Expression (9-3) given below by using the sine component buffer ILSIN[i] (where i=0, . . . , N−1) of the electric current IL of the reactor 242 (step S176d), calculates a Fourier cosine coefficient aIL of the electric current IL of the reactor 242 according to Expression (9-4) given below by using the cosine component buffer ILCOS[n] of the electric current IL of the reactor 242 (step S178d) and then terminates the electrical first variation component detecting process of
The inverter bus bar current estimator 372 is described next.
After obtaining the input data, the inverter bus bar current estimator 372 subsequently calculates the Fourier cosine coefficient aIm of the bus bar current Im of the inverter 34 according to Equation (9-5) given below by using the input Fourier cosine coefficient aIL of the electric current IL of the reactor 242, the input electrical angular velocity ωe of the motor 32 and the input Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39 (step S220d). The inverter bus bar current estimator 372 also calculates the Fourier sine coefficient bIm of the bus bar current Im of the inverter 34 according to Equation (9-6) given below by using the input Fourier sine coefficient bIL of the electric current IL of the reactor 242, the input electrical angular velocity ωe of the motor 32 and the input Fourier cosine coefficient aVH of the voltage VH of the smoothing capacitor 39 (step S230d) and then terminates the inverter bus bar current estimating process of
a
Im
=Da
IL−ωeCHbVH (9-5)
b
Im
=Db
IL+ωeCHaVH (9-6)
The following describes a method of deriving Equation (9-5) and Equation (9-6) given above. Equation (9-7) given below holds between supplied electric current ICNV of the boost converter 240, the electric current Ic of the smoothing capacitor 39 and the bus bar current Im of the inverter 34 on the periphery of the smoothing capacitor 39. Equation (9-8) given below also holds between the electric current Ic and the voltage VH of the smoothing capacitor 39 by using the capacity value CH of the smoothing capacitor 39.
The electrical first variation component VH1 of the voltage VH of the smoothing capacitor 39 is expressed by Equation (9-9) given below. Accordingly, when the attention is focused on only the electrical first variation component VH1 with regard to the voltage VH of the smoothing capacitor 39, the electrical first variation component Ic1 of the electric current Ic of the smoothing capacitor 39 is expressed by Equation (9-10) given below, based on Equations (9-8) and (9-9). On definition of Equation (9-11) given below, Equation (9-12) given below is derived from Equation (9-10).
The supplied electric current ICNV of the boost converter 240 is approximated by Equation (9-13) given below by using the electric current IL of the reactor 242 and the duty command D. An electrical first variation component IL1 of the electric current IL of the reactor 242 is expressed by Equation (9-14) given below. Accordingly, when the attention is focused on only electrical first variation components ICNV1 and VH1 with regard to the supplied electric current ICNV of the boost converter 240 (i.e., the electric current IL of the reactor 242) and the voltage VH of the smoothing capacitor 39, the electrical first variation component Im1 of the bus bar current Im of the inverter 34 is expressed by Equation (9-15) given below, based on Equation (9-7) and Equations (9-12) to (9-14).
I
CNV
=DI
L (9-13)
I
L1
=a
IL cos θ+binIL sin θ (9-14)
I
m1
=Da
IL cos θ+DbIL sin θ−ωeCH(bVH cos θ−aVH sin θ) (9-15)
Equation (9-16) given below is derived by rewriting Equation (9-15). A coefficient for cos θ in the first term on the right side of this Equation (9-16) is obtained as the Fourier cosine coefficient aIm of the bus bar current Im of the inverter 34 as shown by Equation (9-5) given above. A coefficient for sine in the second term on the right side of Equation (9-16) is obtained as the Fourier sine coefficient bIm of the bus bar current Im of the inverter 34 as shown by Equation (9-6) given above. Accordingly, Equation (9-16) is rewritten as Equation (9-17) given below. By taking into account the foregoing, the inverter bus bar current estimator 372 (i.e., the inverter bus bar current estimating process of
I
m1=(DaIL−ωeCHbVH)cos θ+(DbIL+ωeCHaVH)sin θ (9-16)
I
m1
=a
Im cos θ+bIm sin θ (9-17)
Calculating the Fourier cosine coefficient aIm and the Fourier sine coefficient bIm of the bus bar current Im of the inverter 34 by using the electric current IL of the reactor 242 (more specifically, the Fourier cosine coefficient aIL and the Fourier sine coefficient bIL of the electric current IL of the reactor 242) as described above enables the Fourier cosine coefficient aIm and the Fourier sine coefficient bIm of the bus bar current Im of the inverter 34 to be calculated without using the resistance value RB of the internal resistance 36r and the inductance value LB of the internal inductance 361 of the battery 36 and the resistance value RR of the resistance component 242r and the inductance value LR of the inductance component 2421 of the reactor 242, i.e., without being affected by variations in the resistance value RB, the inductance value LB, the resistance value RR and the inductance value LR.
As described above, like the driving system 20 of the first embodiment, the driving system 320 of the fourth embodiment performs Fourier series expansion of the voltage VH of the smoothing capacitor 39 input from the voltage sensor 39a to calculate the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39, and controls the inverter 34, based on the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39. Like the configuration of the driving system 20 of the first embodiment, the configuration of the driving system 320 of the fourth embodiment sufficiently removes the components other than the electrical first variation component of the voltage VH of the smoothing capacitor 39 (for example, electrical second variation component and electrical sixth variation component) and more effectively suppresses a voltage variation of the smoothing capacitor 39 and a torque variation of the motor 32.
Furthermore, the driving system 320 of the fourth embodiment calculates the Fourier cosine coefficient aIm and the Fourier sine coefficient bIm of the bus bar current Im of the inverter 34 by using the electric current IL of the reactor 242 (more specifically, the Fourier cosine coefficient aIL and the Fourier sine coefficient bIL of the electric current IL of the reactor 242). This configuration enables the Fourier cosine coefficient aIm and the Fourier sine coefficient bIm of the bus bar current Im of the inverter 34 to be calculated without using the resistance value RB of the internal resistance 36r and the inductance value LB of the internal inductance 361 of the battery 36 and the resistance value RR of the resistance component 242r and the inductance value LR of the inductance component 2421 of the reactor 242, i.e., without being affected by variations in the resistance value RB, the inductance value LB, the resistance value RR and the inductance value LR.
The following describes a driving system 420 according to a fifth embodiment. The driving system 420 of the fifth embodiment has an identical hardware configuration with the hardware configuration of the driving system 120 of the second embodiment shown in
The following describes a series of control of the inverter 34 performed by the electronic control unit 50 in the driving system 420 of the fifth embodiment.
The inverter bus bar power estimator 472 is described first.
When the inverter bus bar power estimating process of
After calculating the Fourier cosine coefficient aIm and the Fourier sine coefficient bIm of the bus bar current Im of the inverter 34 (steps S220 and S230), the inverter bus bar power estimator 472 extracts a 0-th variation component IB0 of the electric current IB of the battery 36 by low pass filter processing of the electric current IB of the battery 36 (step S240e) and sets the extracted 0-th variation component IB0 of the electric current IB of the battery 36 to a 0-th variation component Imo of the bus bar current Im of the inverter 34 (step S242e). The smoothing capacitor 39 does not allow the DC current to flow, so that the processing of step S242e is performed on the basis of “IB=Im” in a limited ultralow frequency domain.
The inverter bus bar power estimator 472 subsequently extracts a 0-th variation component VH0 of the voltage VH of the smoothing capacitor 39 by low pass filter processing of the voltage VH of the smoothing capacitor 39 (step S244e). The inverter bus bar power estimator 472 calculates a Fourier cosine coefficient aPm of a bus bar power Pm of the inverter 34 according to Equation (10-1) given below by using the Fourier cosine coefficient aIm of the bus bar current Im of the inverter 34, the Fourier cosine coefficient aVH and the 0-th variation component VH0 of the voltage VH of the smoothing capacitor 39 and the 0-th variation component Imo of the bus bar current Im of the inverter 34 (step S246e). The inverter bus bar power estimator 472 also calculates a Fourier sine coefficient bPm of the bus bar power Pm of the inverter 34 according to Equation (10-2) given below by using the Fourier sine coefficient bIm of the bus bar current Im of the inverter 34, the Fourier sine coefficient bVH and the 0-th variation component VH0 of the voltage VH of the smoothing capacitor 39 and the 0-th variation component Imo of the bus bar current Im of the inverter 34 (step S248e) and then terminates the inverter bus bar power estimating process of
a
Pm
=V
H0
a
Im
+I
m0
a
VH (10-1)
b
Pm
=V
H0
b
Im
+I
m0
b
VH (10-2)
The following describes a method of deriving Equation (10-1) and Equation (10-2) given above. An electrical first variation component P1 of the bus bar power Pm of the inverter 34 is expressed by Equation (3-21) given above in the case of generation of the offsets in the electric currents IU, IV and IW of the respective phases of the motor 32. When each of the voltage VH of the smoothing capacitor 39 and the bus bar current Im of the inverter 34 is divided into an electrical 0-th component and an electrical first component, the voltage VH of the smoothing capacitor 39 and the bus bar current Im of the inverter 34 are respectively expressed by Equation (10-3) and by Equation (10-4). The Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39 in Equation (10-3) are expressed by Expressions (1-1) to (1-4) given above. The Fourier cosine coefficient aIm and the Fourier sine coefficient bIm of the bus bar current Im of the inverter 34 in Equation (10-4) are expressed by Equations (2-1) and (2-2) given above.
V
H
=V
H0+αVH sin(θe+α)+bVH cos(θe+α) (10-3)
I
m
=I
m0
+a
Im cos(θe+α)+bIm sin(θe+α) (10-4)
The electrical first variation component P1 of the product of the voltage VH of the smoothing capacitor 39 and the bus bar current Im of the inverter 34 is the sum of the products of the 0-th variation components and the first variation components of Equation (10-3) and Equation (10-4). This gives Equation (10-5) and thereby Equation (10-6).
P
1
=V
H0(aIm cos(θe+α)+bIm sin(θe+α))+Im0(aVH cos(θe+α)+bVH sin(θe+α)) (10- 5)
P
1=(VH0aIm+Im0aVH)cos(θe+α)+(VH0bIm+Im0bVH)sin(θe+α) (10-6)
A coefficient for cos (θe+α) in the first term on the right side of this Equation (10-6) is obtained as the Fourier cosine coefficient aPm of the bus bar power Pm of the inverter 34 as shown by Equation (10-1) given above. A coefficient for sin(θe+α) in the second term on the right side of Equation (10-6) is obtained as the Fourier sine coefficient bPm of the bus bar power Pm of the inverter 34 as shown by Equation (10-2) given above. Accordingly, Equation (10-6) is rewritten as Equation (10-7) given below. By taking into account the foregoing, the inverter bus bar power estimator 472 (i.e., the inverter bus bar power estimating process of
P
1
=a
P m cos(θe+α)+bPm sin(θe+α) (10-7)
The phase current offset estimator 473 is described next.
After obtaining the input data, the phase current offset estimator 473 calculates a fundamental wave voltage amplitude V1 that is a maximum value of a fundamental wave component of a three-phase AC voltage according to Equation (10-8) given below by using the modulation degree Vr of the output voltage of the inverter 34 and the voltage VH of the smoothing capacitor 39 (step S302e). The phase current offset estimator 473 subsequently calculates current offset estimated values IVofs and IWofs of the V phase and the W phase according to Equation (10-9) and Equation (10-10) given below by using the Fourier cosine coefficient aPm and the Fourier sine coefficient bPm of the bus bar power Pm of the inverter 34 and the fundamental wave voltage amplitude V1 (steps S310e and S320e) and then terminates the phase current offset estimating process of
The following describes a method of deriving Equation (10-9) and Equation (10-10) given above. Under the condition of” θe+α=150°”, Equation (10-11) and Equation (10-12) given below are obtained from Equation (3-21) and Equation (10-7) given above, so that Equation (10-10) given above is obtained. Under the condition of” θe+α=210°”, on the other hand, Equation (10-13) and Equation (10-14) given below are obtained from Equation (3-21) and Equation (10-7), so that Equation (10-9) given above is obtained. By taking into account the foregoing, the phase current offset estimator 473 (i.e., the phase current offset estimating process of
P
1(150°)=√{square root over (3)}V1IWofs sin(−60°) (10-11))
P
1(150°)=aP cos(150°)+bP sin(150°) (10-12)
P
1(210°)=√{square root over (3)}V1IVofs)sin(60°) (10-13))
P
1(210°)=aP cos(210°)+bP sin(210°) (10-14)
As described above, like the driving system 20 of the first embodiment, the driving system 420 of the fifth embodiment performs Fourier series expansion of the voltage VH of the smoothing capacitor 39 input from the voltage sensor 39a to calculate the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39, and controls the inverter 34, based on the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39. Like the configuration of the driving system 20 of the first embodiment, the configuration of the driving system 420 of the fifth embodiment sufficiently removes the components other than the electrical first variation component of the voltage VH of the smoothing capacitor 39 (for example, electrical second variation component and electrical sixth variation component) and more effectively suppresses a voltage variation of the smoothing capacitor 39 and a torque variation of the motor 32.
Furthermore, the driving system 420 of the fifth embodiment calculates the Fourier cosine coefficient aPm and the Fourier sine coefficient bPm of the bus bar power Pm of the inverter 34 and calculates the current offset estimated values IWofs and IWofs of the V phase and the W phase based on the calculated Fourier cosine coefficient aPm and the calculated Fourier sine coefficient bPm of the bus bar power Pm of the inverter 34. This procedure performs the calculation by taking into account both a variation in the bus bar current Im of the inverter 34 and a variation in the voltage VH of the smoothing capacitor 39. This accordingly increases the accuracies of the current offset estimated values IVofs and IWofs of the V phase and the W phase.
The driving system 420 of the fifth embodiment has the similar hardware configuration to the hardware configuration of the driving system 120 of the second embodiment shown in
The following describes a driving system 520 according to a sixth embodiment.
The voltage sensor 540 includes an amplifier 541 and a low pass filter 542. The amplifier 541 serves to amplify the voltage between terminals of the smoothing capacitor 39 and output the amplified voltage. The low pass filter 542 includes a resistance element 543 and a capacitor 544. The resistance element 543 has one terminal that is connected with an output side of the amplifier 541 and the other terminal that is connected with the electronic control unit 50. The capacitor 544 has one terminal that is connected with the other terminal of the resistance element 543 and the other terminal that is grounded. The low pass filter 542 processes the output of the amplifier 541 by low pass filter processing and outputs the processed output as a voltage VH of the smoothing capacitor 39 to the electronic control unit 50.
The following describes a series of control of the inverter 34 performed by the electronic control unit 50 in the driving system 520 of the sixth embodiment.
When the electrical first variation component detecting process of
The electrical first variation component detector 571 updates the sine component buffer VHSIN[n] and the cosine component buffer VHCOS[n] of the voltage VH of the smoothing capacitor 39 (steps S140 and S150) and subsequently calculates a Fourier sine coefficient temporary value bVHtemp that is a temporary value (tentative value) of the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39 according to Expression (11-1) given below by using the sine component buffer VHSIN[i] (where i=0, . . . , N−1) of the voltage VH of the smoothing capacitor 39 (step S160f). The electrical first variation component detector 571 also calculates a Fourier cosine coefficient temporary value aVHtemp that is a temporary value of the Fourier cosine coefficient aVH of the voltage VH of the smoothing capacitor 39 according to Expression (11-2) given below by using the cosine component buffer VHCOS [i] of the voltage VH of the smoothing capacitor 39 (step S170f).
The electrical first variation component detector 571 subsequently sets a correction amplitude ratio AVH and a correction phase difference ΔθVH, which are used to correct a deviation of the voltage VH of the smoothing capacitor 39 input from the voltage sensor 540 relative to the actual voltage VHact of the smoothing capacitor 39, based on the electrical angular velocity ωe of the motor 32 (step S172f). A procedure employed to set the correction amplitude ratio AVH and the correction phase difference ΔθVH according to the embodiment specifies and stores in advance relationships of the correction amplitude ratio AVH and the correction phase difference ΔθVH to the electrical angular velocity ωe of the motor 32 in the form of a map in the ROM 52. When a value of the electrical angular velocity ωe of the motor 32 is given, the procedure reads to set values of the correction amplitude ratio AVH and the correction phase difference ΔθVH corresponding to the given value of the electrical angular velocity ωe from this map.
The electrical first variation component detector 571 subsequently calculates the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39 according to Equation (11-4) and Equation (11-5) given below by using the Fourier cosine coefficient temporary value aVHtemp and the Fourier sine coefficient temporary value bVHtemp of the voltage VH of the smoothing capacitor 39, the correction amplitude ratio AVH and the correction phase difference ΔθVH (steps S174f and S176f) and then terminates the electrical first variation component detecting process of
a
VH
=A
VH×(aVHtmp cos ΔθVH+bVHtmp sin ΔθVH) (11-4)
b
VH
=A
VH×(bVHtmp cos ΔθVH−aVHtmp sin ΔθVH) (11-5)
The processing of step S172f to S176f described above converts the voltage VH input from the voltage sensor 540 into a value closer to the actual voltage VHact by taking into account the characteristics of the low pass filter 542 to calculate the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39.
As described above, like the driving system 20 of the first embodiment, the driving system 520 of the sixth embodiment performs Fourier series expansion of the voltage VH of the smoothing capacitor 39 input from the voltage sensor 540 to calculate the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39, and controls the inverter 34, based on the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39. Like the configuration of the driving system 20 of the first embodiment, the configuration of the driving system 520 of the sixth embodiment sufficiently removes the components other than the electrical first variation component of the voltage VH of the smoothing capacitor 39 (for example, electrical second variation component and electrical sixth variation component) and more effectively suppresses a voltage variation of the smoothing capacitor 39 and a torque variation of the motor 32.
Furthermore, the driving system 520 of the sixth embodiment calculates the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39 by taking into account the frequency characteristics of the voltage sensor 540. This configuration enables the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39 to be calculated with the higher accuracy.
The driving system 520 of the sixth embodiment having the configuration shown in
The following describes a driving system 620 according to a seventh embodiment. The driving system 620 of the seventh embodiment has an identical hardware configuration with the hardware configuration of the driving system 20 of the first embodiment shown in
The following describes a series of control of the inverter 34 performed by the electronic control unit 50 in the driving system 620 of the seventh embodiment.
The phase current offset controller 674 is described first.
After obtaining the input data, the phase current offset controller 674 calculates a voltage offset correction amount VVofs of the V phase according to Expression (12-1) given below by using the input current offset estimated value IVofs of the V phase (step S410g), calculates a voltage offset correction amount VWofs of the W phase according to Expression (12-2) given below by using the input current offset estimated value IVofs of the W phase (step S420g), and then terminates the phase current offset control process of
V
Vofs
←K
P2(0−IVofs)+KI2∫(0−IVofs)dt (12-1)
V
Wofs
←K
P2(0−IWofs)+KI2∫(0−IWofs)dt (12-2)
The voltage offset correction amounts VVofs and VWofs of the V phase and the W phase respectively denote correction amounts used by the current controller 677 for correction of voltage commands VV* and VW* of the V phase and the W phase. Expression (12-1) and Expression (12-2) are relational expressions of feedback control respectively used to calculate the voltage offset correction amounts VVofs and VWofs of such that the current offset estimated values IVofs and IWofs of the V phase and the W phase become equal to a value 0. In Expression (12-1) and Expression (12-2), “KP2” denotes a gain of a proportional, and “KI2” denotes a gain of an integral term.
The phase current physical value calculator 676 is described next.
When the phase current physical value calculating process of
I
Vcon=(ADIV−2.5V)×ADVgain (12-3)
I
Wcon=(ADIW−2.5V)×ADWgain (12-4)
Equation (12-3) given above is equivalent to Equation (5-1) with setting the offset correction amount ADVofs of the V phase to a value 0. Equation (12-4) is equivalent to Equation (5-2) with setting the offset correction amount ADWofs of the W phase to a value 0. Accordingly, when offsets are generated in the electric currents IU, IV and IW of the respective phases, the offset amounts (actual offset amounts) included in the electric currents IV and IW of the V phase and the W phase are reflected on the AD values ADIV and ADIW from the current sensors 32v and 32w and thereby on the control currents IVcon and VWcon of the V phase and the W phase (i.e., the electric currents of the V phase and the W phase recognized by the electronic control unit 50).
The current controller 677 is described below.
The adders 90v and 90w serve to correct the voltage commands VV* and VW* of the V phase and the W phase from the coordinate converter 86 by adding the voltage offset correction amounts VVofs and VWofs to the voltage commands VV* and VW* of the V phase and the W phase and output the corrected voltage commands VV* and VW* of the V phase and the W phase to the PWM signal generator 87.
As described above, like the driving system 20 of the first embodiment, the driving system 620 of the seventh embodiment performs Fourier series expansion of the voltage VH of the smoothing capacitor 39 input from the voltage sensor 39a to calculate the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39, and controls the inverter 34, based on the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39. Like the configuration of the driving system 20 of the first embodiment, the configuration of the driving system 620 of the seventh embodiment sufficiently removes the components other than the electrical first variation component of the voltage VH of the smoothing capacitor 39 (for example, electrical second variation component and electrical sixth variation component) and more effectively suppresses a voltage variation of the smoothing capacitor 39 and a torque variation of the motor 32.
The following describes a driving system 720 according to an eighth embodiment. The driving system 720 of the eighth embodiment has an identical hardware configuration with the hardware configuration of the driving system 20 of the first embodiment shown in
The following describes a series of control of the inverter 34 performed by the electronic control unit 50 in the driving system 720 of the eighth embodiment.
The phase current offset controller 774 is described first.
After obtaining the input data, the phase current offset controller 774 calculates a pulse width correction amount θVofs of the V phase according to Expression (13-1) given below by using the input current offset estimated value IVofs of the V phase (step S410h), calculates a pulse width correction amount θWofs of the W phase according to Expression (13-2) given below by using the input current offset estimated value IWofs of the W phase (step S420h), and then terminates the phase current offset control process of
θVofs←KP3(0−IVofs)+KI3∫(0−IVofs)dt (13-1)
θWofs←KP3(0−IWofs)+KI3∫(0−IWofs)dt (13-2)
The pulse width correction amounts θVofs and θWofs of the V phase and the W phase respectively denote correction amounts used by the torque controller 777 for correction of pulse widths of rectangular wave pulse signals of the V phase and the W phase. Expression (13-1) and Expression (13-2) are relational expressions of feedback control respectively used to calculate the pulse width correction amounts θVofs and θWofs, such that the current offset estimated values IVofs and IWofs of the V phase and the W phase become equal to a value 0. In Expression (13-1) and Expression (13-2), “KP3” denotes a gain of a proportional, and “KI3” denotes a gain of an integral term.
The phase current physical value calculator 776 is described next. The phase current physical value calculator 776 performs the phase current physical value calculating process of
The torque controller 777 is described below.
The low pass filter 781 serves to process the torque command Tm* of the motor 32 by low pass filter processing and generate a filtered torque command Tmf*. The coordinate converter 782 serves to perform coordinate conversion (three phase to two phase conversion) of the control currents IVcon and IWcon of the V phase and the W phase of the motor 32 into electric currents Id and Iq of the d axis and the q axis by using the electrical angle θe of the motor 32 on the assumption that the sum of the electric currents of the respective phases is equal to zero. The torque estimator 783 serves to determine a torque estimated value Tmes of the motor 32, based on the electric currents Id and Iq of the d axis and the q axis. The subtractor 784 serves to calculate a difference ΔTm between the filtered torque command Tmf* and the torque estimated value Tmes of the motor 32.
The feedback controller 785 serves to perform torque feedback control and calculate a tentative voltage phase ϕqtemp that is a tentative value of the voltage phase ϕq of the output voltage of the inverter 34, such that the difference ΔTm becomes equal to a value 0. The upper/lower limit restrictor 786 serves to set the voltage phase ϕq by applying upper limit and lower limit guarding to the tentative voltage phase ϕqtemp.
The rectangular wave pulse generator 787 serves to generate rectangular wave pulses of the transistors T11 to T16 by using the electrical angle θe of the motor 32, the voltage phase ϕq of the output voltage of the inverter 34 and the pulse width correction amounts θVofs and θWofs of the V phase and the W phase and to perform switching control of the transistors T11 to T16.
As described above, like the driving system 20 of the first embodiment, the driving system 720 of the eighth embodiment performs Fourier series expansion of the voltage VH of the smoothing capacitor 39 input from the voltage sensor 39a to calculate the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39, and controls the inverter 34, based on the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39. Like the configuration of the driving system 20 of the first embodiment, the configuration of the driving system 720 of the eighth embodiment sufficiently removes the components other than the electrical first variation component of the voltage VH of the smoothing capacitor 39 (for example, electrical second variation component and electrical sixth variation component) and more effectively suppresses a voltage variation of the smoothing capacitor 39 and a torque variation of the motor 32.
The following describes a driving system 820 according to a ninth embodiment. The driving system 820 of the ninth embodiment has an identical hardware configuration with the hardware configuration of the driving system 20 of the first embodiment shown in
The following describes a series of control of the inverter 34 performed by the electronic control unit 50 in the driving system 820 of the ninth embodiment.
In the electrical first variation component detecting process of
When the buffer index n is smaller than the product of the value N and the value M at step S120i, the electrical first variation component detector 871 updates the sine component buffer VHSIN[n] and the cosine component buffer VHCOS[n] of the voltage VH of the smoothing capacitor 39 with regard to the buffer index n (steps S140 and S150). When the buffer index n is equal to or larger than the product of the value N and the value M, on the other hand, the electrical first variation component detector 871 resets the buffer index n to the value 0 (step S130) and updates the sine component buffer VHSIN[n] and the cosine component buffer VHCOS[n] of the voltage VH of the smoothing capacitor 39 with regard to the buffer index n (steps S140 and S150).
The electrical first variation component detector 871 subsequently calculates the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39 according to Expression (14-1) given below by using the sine component buffer VHSIN[i] (where i=0, . . . , N×M−1) of the voltage VH of the smoothing capacitor 39 (step S160i). The electrical first variation component detector 871 also calculates the Fourier cosine coefficient aVH of the voltage VH of the smoothing capacitor 39 according to Expression (14-2) given below by using the cosine component buffer VHCOS[i] of the voltage VH of the smoothing capacitor 39 (step S170i) and then terminates the electrical first variation component detecting process of
As described above, like the driving system 20 of the first embodiment, the driving system 820 of the ninth embodiment performs Fourier series expansion of the voltage VH of the smoothing capacitor 39 input from the voltage sensor 39a to calculate the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39, and controls the inverter 34, based on the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39. Like the configuration of the driving system 20 of the first embodiment, the configuration of the driving system 820 of the ninth embodiment sufficiently removes the components other than the electrical first variation component of the voltage VH of the smoothing capacitor 39 (for example, electrical second variation component and electrical sixth variation component) and more effectively suppresses a voltage variation of the smoothing capacitor 39 and a torque variation of the motor 32.
Furthermore, the driving system 820 of the ninth embodiment calculates the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39 by using the data of the period M (where M≥2) at the electrical angle θe of the motor 32. This configuration enables the Fourier cosine coefficient aVH and the Fourier sine coefficient bVH of the voltage VH of the smoothing capacitor 39 to be calculated with the higher accuracy.
In the driving systems 20, 120, 220, 320, 420, 520, 620, 720 and 820 of the first to the ninth embodiments described above, the offset unit execution determining process of
In the driving system of the present invention, the control device may control the inverter by regulating an offset amount of the detected value of the current sensor, such that the electrical first variation component of the voltage of the smoothing capacitor becomes equal to the value 0. This configuration more effectively suppresses the electrical first variation component of the voltage of the smoothing capacitor by regulating the offset amount of the detected value of the current sensor.
In the driving system of the present invention, the control device may calculate an electrical first variation component of a bus bar current or a bus bar power of the inverter, based on the electrical first variation component of the voltage of the smoothing capacitor, and the control device may control the inverter, based on the calculated electrical first variation component of the bus bar current or the bus bar power of the inverter, such that the electrical first variation component of the voltage of the smoothing capacitor becomes equal to the value 0. This configuration calculates and uses the electrical first variation component of the bus bar current or the bus bar power of the inverter and thereby more effectively suppresses the electrical first variation component of the voltage of the smoothing capacitor.
In the driving system of the present disclosure calculating an electrical first variation component of a bus bar current or a bus bar power of the inverter, based on the electrical first variation component of the voltage of the smoothing capacitor, the control device may calculate the electrical first variation component of the bus bar current or the bus bar power of the inverter, based on the electrical first variation component of the voltage of the smoothing capacitor and a frequency characteristic of a circuit from the power storage device to the smoothing capacitor. This configuration enables the electrical first variation component of the bus bar current or the bus bar power of the inverter to be calculated by taking into account the frequency characteristic of the circuit from the power storage device to the smoothing capacitor. In the driving system of this aspect, the control device may use an amplitude ratio and a phase difference between the bus bar current of the inverter and the voltage of the smoothing capacitor as the frequency characteristic of the circuit from the power storage device to the smoothing capacitor.
In the driving system of the present disclosure calculating an electrical first variation component of a bus bar current or a bus bar power of the inverter, based on the electrical first variation component of the voltage of the smoothing capacitor, the driving system may include a boost converter that is provided between the inverter along with the smoothing capacitor and the power storage device on the power line and that may be configured to include switching elements of an upper arm and of a lower arm and a reactor, wherein the control device may calculate the electrical first variation component of the bus bar current or the bus bar power of the inverter, based on the electrical first variation component of the voltage of the smoothing capacitor and a frequency characteristic of a circuit from the power storage device to the smoothing capacitor, when the upper arm is kept on. This configuration enables the electrical first variation component of the bus bar current or the bus bar power of the inverter to be calculated by taking into account the frequency characteristic of the circuit from the power storage device to the smoothing capacitor, when the upper arm of the boost converter is kept on. In the driving system of this aspect, the control device may use an amplitude ratio and a phase difference between the bus bar current of the inverter and the voltage of the smoothing capacitor as the frequency characteristic of the circuit from the power storage device to the smoothing capacitor.
In the driving system of the present disclosure calculating an electrical first variation component of a bus bar current or a bus bar power of the inverter, based on the electrical first variation component of the voltage of the smoothing capacitor, the driving system may include a second current sensor configured to detect an electric current of the power storage device, wherein the control device may perform Fourier series expansion of a detected value of the second current sensor to calculate an electrical first variation component of the electric current of the power storage device, and the control device may calculate the electrical first variation component of the bus bar current or the bus bar power of the inverter, based on the electrical first variation component of the voltage of the smoothing capacitor and the electrical first variation component of the electric current of the power storage device. This configuration enables the electrical first variation component of the bus bar current or the bus bar power of the inverter to be calculated by taking into account the electrical first variation component of the electric current of the power storage device.
In the driving system of the present disclosure calculating an electrical first variation component of a bus bar current or a bus bar power of the inverter, based on the electrical first variation component of the voltage of the smoothing capacitor, the driving system may include a boost converter that is provided between the inverter along with the smoothing capacitor and the power storage device on the power line and that may be configured to include switching elements of an upper arm and of a lower arm and a reactor; and a third current sensor configured to detect an electric current of the reactor, wherein the control device may perform Fourier series expansion of a detected value of the third current sensor to calculate an electrical first variation component of the electric current of the reactor, and the control device may calculate the electrical first variation component of the bus bar current or the bus bar power of the inverter, based on the electrical first variation component of the voltage of the smoothing capacitor, the electrical first variation component of the electric current of the reactor and a duty command used to control the boost converter. This configuration enables the electrical first variation component of the bus bar current or the bus bar power of the inverter to be calculated by taking into account the electrical first variation component of the electric current of the reactor included in the boost converter.
In the driving system of the present disclosure calculating an electrical first variation component of a bus bar current or a bus bar power of the inverter, based on the electrical first variation component of the voltage of the smoothing capacitor, the control device may estimate an offset amount of the detected value of the current sensor, based on the electrical first variation component of the bus bar current or the bus bar power of the inverter, the control device may calculate a control current of each phase of the motor by correcting the detected value of the current sensor, such that the offset amount of the detected value of the current sensor becomes equal to a value 0, and the control device may control the inverter, based on the calculated control current of each phase of the motor.
In the driving system of the present disclosure calculating an electrical first variation component of a bus bar current or a bus bar power of the inverter, based on the electrical first variation component of the voltage of the smoothing capacitor, when the control device controls the inverter in a pulse width modulation control mode, the control device may estimate an offset amount of the detected value of the current sensor, based on the electrical first variation component of the bus bar current or the bus bar power of the inverter, and the control device may control the inverter with correcting a voltage command of each phase of the motor, such that the offset amount of the detected value of the current sensor becomes equal to a value 0.
In the driving system of the present disclosure calculating an electrical first variation component of a bus bar current or a bus bar power of the inverter, based on the electrical first variation component of the voltage of the smoothing capacitor, the inverter may have switching elements of an upper arm and a lower arm with respect to each phase, and when the control device controls the inverter in a rectangular wave control mode, the control device may estimate an offset amount of the detected value of the current sensor, based on the electrical first variation component of the bus bar current or the bus bar power of the inverter, and the control drive may control the inverter with correcting a pulse width of the upper arm and the lower arm with respect to each phase, such that the offset amount of the detected value of the current sensor becomes equal to a value 0.
In the driving system of the present disclosure, when an electrical angular velocity of the motor is equal to or higher than a predetermined angular velocity and a modulation degree of an output voltage of the inverter is equal to or higher than a predetermined modulation degree, the control device may perform Fourier series expansion of the detected value of the voltage sensor to calculate the electrical first variation component of the voltage of the smoothing capacitor, and the control device may control the inverter, such that the electrical first variation component of the voltage of the smoothing capacitor becomes equal to the value 0.
In the driving system of the present disclosure, the control device may calculate the electrical first variation component of the voltage of the smoothing capacitor by taking into account a frequency characteristic of the voltage sensor. This configuration enables the electrical first variation component of the voltage of the smoothing capacitor to be calculated with the higher accuracy.
In the driving system of the present disclosure, the control device may perform Fourier series expansion of the detected value of the voltage sensor by using data of multiple periods at an electrical angle of the motor. This configuration enables the electrical first variation component of the voltage of the smoothing capacitor to be calculated with the higher accuracy.
The following describes a correspondence relationship between the primary components of the respective embodiments described above and the primary components in the respective aspects of the present disclosure described in Summary. The motor 32, the inverter 34, the smoothing capacitor 39, the current sensors 32v and 32w, the electronic control unit 50 and the voltage sensor 39a of the embodiments respectively correspond to the “motor”, the “inverter”, the “smoothing capacitor”, the “current sensor”, the “control device”, and the “voltage sensor” of the respective aspects.
The correspondence relationship between the primary components of the embodiment and the primary components of the disclosure, regarding which the problem is described in Summary, should not be considered to limit the components of the disclosure, regarding which the problem is described in Summary, since the embodiment is only illustrative to specifically describes the aspects of the disclosure, regarding which the problem is described in Summary. In other words, the disclosure, regarding which the problem is described in Summary, should be interpreted on the basis of the description in the Summary, and the embodiment is only a specific example of the disclosure, regarding which the problem is described in Summary.
The aspect of the disclosure is described above with reference to the embodiment. The disclosure is, however, not limited to the above embodiment but various modifications and variations may be made to the embodiment without departing from the scope of the disclosure.
The technique of the disclosure is preferably applicable to the manufacturing industries of the driving system and so on.
Number | Date | Country | Kind |
---|---|---|---|
2019-022724 | Feb 2019 | JP | national |