Equalization based on digital signal processing in downsampled domains

Abstract

This invention relates to a device, a method, a software application program, a software application program product and an audio device for processing a digital signal, wherein the digital signal is separated and downsampled into at least two downsampled subband signals, wherein at least one of the at least two downsampled subband signals is equalized, and wherein the at least two downsampled subband signals are upsampled and combined into a digital output signal.

Description

BRIEF DESCRIPTION OF THE FIGURES

In the figures show:

FIG. 1: an illustration of a target equalizer magnitude response in the frequency domain;

FIG. 2: an illustration of subbands of downsampled subband signals and a target equalizer magnitude response in the frequency domain according to a preferred embodiment of the present invention;

FIG. 3: a schematic presentation of components of a device for equalization according to a preferred embodiment of the present invention;

FIG. 4: a schematic presentation of components of a quadrature mirror filter analysis filter and an quadrature mirror filter synthesis filter according to a preferred embodiment of the present invention;

FIG. 5: a magnitude transfer function of the quadrature mirror filter analysis filter and the quadrature mirror filter synthesis filter of FIG. 4;

FIG. 6: a schematic presentation of components of an audio device according to a preferred embodiment of the present invention;

FIG. 7: a flowchart of a method for equalizing in downsampled subband domains according to a preferred embodiment of the present invention.

DETAILED DESCRIPTION OF THE INVENTION

The present invention proposes to equalize a digital signal by separating and downsampling said digital signal into at least two downsampled subband signals; by equalizing at least one of said at least two downsampled subband signals; and by upsampling and combining said at least two downsampled subband signals into an output digital signal.

In the following, the present invention will be described for a preferred embodiment.

In this preferred embodiment, a digital audio signal is equalized according to the present invention. Said equalizing may be performed according to a target equalizer transfer function, wherein said equalizer (EQ) target transfer function is represented by a target EQ magnitude response 100,203.

FIG. 2 schematically depicts the separating of an available frequency region of said digital audio signal into three subbands 200,201,202, wherein said available frequency range of said digital audio signal spans a frequency range from f₁=0 Hz to f₂=F_S/2, and wherein F_S denotes the sampling rate of said digital audio signal and f₂=F_S/2 denotes the corresponding the Nyquist frequency.

The equalization of said digital audio signal will be performed on three downsampled subband signals corresponding to said three subbands, wherein said three downsampled subband signals are downsampled and separated from said digital audio signal by said separator and downsampler 302,322 as can be seen from FIG. 3. A first subband 200 spans a frequency range from f_1,1=0 Hz to f_2,1=F_S/8, a second subband 201 spans a frequency range from f_1,2=F_S/8 to f_2,2=F_S/4, and a third subband 202 spans a frequency range from f_1,3=F_S/4 to f_2,3=F_S/2.

Furthermore, FIG. 2 schematically depicts the magnitude of a given equalizer (EQ) target transfer function 203, wherein this target EQ magnitude response 100, 203 is separated into n=9 target equalizer bands in the frequency domain which are approximately logarithmically distributed.

FIG. 3 depicts a schematic presentation of the components of a device for equalization according to the preferred embodiment of the present invention. The operation of equalization will now be explained in detail and will be referenced to the method according to the present invention depicted as a flow chart in FIG. 7. It should be noted that this flowchart is of rather general nature and is not limited to the preferred embodiment.

Said device comprises a separator and downsampler 300 wherein said separator and downsampler 300 separates and downsamples a digital signal x into said three downsampled subband signals x_02,x₁₂,x₁₁ in accordance with step 700, wherein said digital signal x represents a digital audio signal according to the preferred embodiment. Furthermore, said device comprises an equalizer 340 in order to equalize said three downsampled subband signals according to step 700 as depicted in FIG. 7. Further, said device comprises a first delay module 350 and a second delay module 351, wherein said first delay module 350 introduces a delay to downsampled subband signal x₁₂, and said second delay module 351 introduces a delay to downsampled subband signal x₁₁ in conformity with step 703. Said device comprises an upsampler and combiner 310, wherein said upsampler and combiner 310 upsample and combine said three downsampled subband signals according to step 704, after being signal processed by said equalizer 340 and by said delay module 350,351 into a digital output signal y.

In the following, the separating and downsampling is explained in details:

The separator and downsampler 300 comprises a first analysis filter 301 and a second analysis filter 302, wherein said analysis filters 301,302 are arranged in a non-symmetric tree structure. In this preferred embodiment, said first analysis filter 301 is represented by a first quadrature mirror filter (QMF) analysis filter 301 and said second analysis filter 302 is represented by a second quadrature mirror filter (QMF) analysis filter 302, but in general, said analysis filters are not restricted to said QMF analysis filters.

The upsampler and combiner 310 comprise a first synthesis filter 311 and a second synthesis filter 312, wherein said synthesis filters 311,312 are arranged in a non-symmetrical tree structure, wherein said non-symmetric tree structure corresponds to said tree structure of said analysis filter. In this preferred embodiment, said first synthesis filter 311 is represented by a first quadrature mirror filter synthesis filter 311 and said second synthesis filter 312 is represented by a second quadrature mirror filter (QMF) synthesis filter 312, but in general, said synthesis filters are not restricted to said QMF synthesis filters.

Said first QMF analysis filter 301 separates and downsamples said digital audio signal x into two downsampled subband signals x₀₁ and x₁₁, wherein the subband of x₀₁ spans a frequency range from 0 Hz to F_S/4, and wherein the subband of x₁₁ spans a frequency range from F_S/4 to F_S/2, and wherein x₀₁ and x₁₁ have a sampling rate being half of the sampling rate F_S of said digital audio signal x. For instance, assuming a sampling rate of F_S=48000 Hz associated with said digital audio signal would lead to a sampling rate of 24000 Hz for said two downsampled subband signals x₀₁ and x₁₁. Said subband signal x₁₁ is outputted from a first output 321 of said first QMF analysis filter 301, and said subband signal x₀₁ is output from a second output 322 of said first QMF analysis filter 301.

According to the non-symmetric tree structure of said analysis filters 301,302, the downsampled subband signal x₀₁ is separated and downsampled by said second QMF 302 into a first downsampled subband signal x₀₂ and into a second downsampled subband signal x₁₂, wherein the subband of x₀₂ corresponds to said first subband 200 spanning a frequency range from f₁₁=0 Hz to f_2,1=F_S/8, and wherein the subband of x₁₂ corresponds to said second subband 201 spanning a frequency range from f_1,2=F_S/8 to f_2,2=F_S/4. The sampling rate F_S,1 of said first downsampled subband signal x₀₂ is F_S,1=F_S/4, and the sampling rate F_S,2 of said second downsampled subband signal x₁₂ is also F_S,2=F_S/4. The downsampled subband signal x₁₁, which is outputted from said first QMF 301, is not fed to another analysis filter, furthermore, the subband of signal x₁₁ corresponds to said third subband 202 spanning a frequency range from f_1,3=F_S/4 to f_2,3=F_S/2.

Thus, said separator and downsampler 300 separates and downsamples said digital audio signal x into said first downsampled subband signal x₀₂, and into said second downsampled subband signal x₁₂, and into a third downsampled subband signal x₁₁, wherein the subbands of said three downsampled subband signals correspond to said three subbands 200,201,202. Said separator and downsampler 300 may comprise more than two analysis filters in order to separate and downsample the digital signal in more than three subbands. Furthermore, the structure of said analysis filters is not restricted to a non-symmetric tree structure. For instance, said analysis filter may be arranged in form of a symmetric tree structure or in form of a filter bank.

In the following, the design of QMF analysis filter and QMF synthesis filter with respect to the characteristic of the target equalizer magnitude response 100,203 is explained in detail.

FIG. 4 depicts a schematic presentation of the components of a QMF analysis filter 401 and a corresponding QMF synthesis filter 402, wherein said QMF analysis filter 401 and said corresponding QMF synthesis filter 402 represents a QMF analysis and synthesis couple. For instance, at least one of said analysis filters 301,302 and the corresponding synthesis filters 311,312 to said at least one of said analysis filters could be represented by at least one of said QMF analysis and synthesis couple, wherein said corresponding synthesis filter may correspond to said at least one of said analysis filters via said non-symmetric tree structure of said analysis filter and of said synthesis filter. For example said first QMF analysis filter 301 and said first QMF synthesis filter 311 could be implemented by said QMF analysis and synthesis couple, and/or said second QMF analysis filter 302 and said second QMF synthesis filter 312 could be implemented by said QMF analysis and synthesis couple.

In order to achieve a sufficient stopband attenuation of the QMF analysis and synthesis couple even in case of large magnitude response level variations (e.g., +/±15 dB) of the target equalizer magnitude response 100,203 it is proposed to implement second order or higher order allpass filters for the realisation of the filters a₀(z) 410,421 and the filters a₁(z) 411,420 of said QMF analysis filter 401 and said QMF synthesis filter 402.

In the preferred embodiment, said QMF analysis filter 401 comprises a first second order allpass filter 410 and a second second order allpass filter 411, wherein said first second order allpass filter 410 has a first transfer function a₀(z) and said second second order allpass filter 411 has a second transfer function a₁(z). Furthermore, said corresponding QMF synthesis filter 402 comprises a third second order allpass filter 421 and a fourth second order allpass filter 420, wherein said third second order allpass filter 421 has said first transfer function a₀(z) and said fourth second order allpass filter 420 has said second transfer function a₁(z). Said first transfer function a₀(z) has the form

$a_0\left(z\right)=\frac{{\alpha }_{01}+{\alpha }_{02}z^{-1}+1z^{-2}}{1+{\alpha }_{02}z^{-1}+{\alpha }_{01}z^{-2}},$

and said second transfer function a₁(z) has the form

$a_1\left(z\right)=\frac{{\alpha }_{11}+{\alpha }_{12}z^{-1}+1z^{-2}}{1+{\alpha }_{12}z^{-1}+{\alpha }_{11}z^{-2}},$

wherein said allpass filters 410,411,420,421 are polyphase components of 9^th order elliptic filters whose poles are on the imaginary axis. However, said allpass filter design is not restricted to second order allpass filters. For instance, higher order allpass filter may be implemented, which may lead to a higher stopband attenuation, and, further, even first order allpass filter may be implemented which may lead to decreased implementation costs. Furthermore, said allpass filters are not restricted being polyphase components of 9^th order elliptic filters whose polse are on the imaginary axis. For instance, said allpass filters may be implemented being polyphase components of decreased or increased order elliptic filter compared to the 9^th order, e.g. being polyphase components of 8^th order, 12^th order or 13^th order.

A criteria for the QMF allpass filter design includes the minimum stopband attenuation and the stopband edge frequency f_st,L of the magnitude response of the low-frequency branch of said QMF analysis filter 401 and said QMF synthesis filter 402, and it includes the minimum stopband attenuation and the stopband edge frequency f_st,H of the magnitude response of the high-frequency branch of said QMF analysis filter 401 and said QMF synthesis filter 402. For the use of said QMF analysis filter 401 and said corresponding QMF synthesis filter 402 in a device for equalization, it is desirable to have the stopband edge frequencies f_st,L and f_st,H as close as possible to F_S/4, wherein F_S denotes the sampling frequency associated with the QMF analysis filter and the corresponding QMF synthesis filter, as the stopband edge frequencies f_st,L and f_st,H define the width (from f_st,H to f_st,L) of the QMF bank transition band 210,211,501, as depicted in FIG. 5. The low-frequency branch's stopband edge frequency f_st,L and the high-frequency branch's stopband edge frequency f_st,H of a QMF bank are related via f_st,H=0.5·F_S−f_st,L. Within each QMF transition band 210,211,501 said target EQ magnitude response 100,203 should be constant in order to avoid aliasing. Thus, each of the crossover frequencies 231,232,233,234,235,236,237,238,239,240 of the n subbands of said target equalizer magnitude response 100,203 has to be chosen being outside of each of the QMF transition bands 210,211.

On the other hand, by allowing larger values of f_st,L, and thus smaller values of f_st,H, it is possible to increase the stopband attenuation, which reduces the risk of audible aliasing of frequency components, which is caused by large variations in the levels of the target EQ magnitude response 100,203, and too small stopband attenuation.

Thus, the allpass filter coefficients of said first, second, third and fourth second order allpass filter of said QMF analysis and synthesis couple, i.e. said QMF bank, are designed so that the stopband edge frequency of the magnitude response of the low-frequency frequency branch is set to f_st,L≈0.316·F_S and that the stopband edge frequency of the magnitude response of the high-frequency branch is set to f_st,H≈0.184·F_S for this preferred embodiment. FIG. 5 depicts the magnitude response 502 of the low-frequency branch 431 associated with said QMF analysis and synthesis couple, and FIG. 5 depicts the magnitude response 503 of the high-frequency branch 432 associated with said QMF analysis and synthesis couple.

Furthermore, FIG. 5 depicts the QMF transition band 501 of said QMF synthesis and analysis couple, wherein said QMF transition band 501 spans a frequency range from 0.184·F_S to 0.316·F_S according to said stopband edge frequencies f_st,H≈0.184·F_S and f_st,L≈0.316·F_S. Within said QMF transition band, said equalizer (EQ) target magnitude response 100,203 must remain constant. FIG. 2 depicts the QMF transition bands 210,211 according to the first preferred embodiment, wherein said QMF synthesis and analysis couple is applied for said first QMF analysis filter 301 and said first QMF synthesis means 311, and wherein said QMF synthesis and analysis couple is applied for said second QMF analysis filter 302 and for said second QMF synthesis filter 312. A first QMF transition band 210 is caused by said first QMF analysis filter 301 and said first QMF synthesis filter 311, and a second QMF transition band 210 is caused by said second QMF analysis filter 302 and said second QMF synthesis filter 311. Within said first QMF transition band and said second QMF transition band said target EQ magnitude response 100,203 must remain constant, as depicted in FIG. 2, in order to maintain a high audio quality.

Assuming a sampling rate of F_S=48000 Hz for said digital audio signal x, said first QMF transition band 211 is in the frequency range from 4416 Hz to 7584 Hz, and said second QMF transition band 210 is in the frequency range from 8832 Hz to 15168 Hz.

In the following, the details of equalization will be explained, in particular the calculation of the filter coefficients in dependency on the target EQ magnitude response.

The equalization of said three downsampled subband signals is performed by said equalizer 340, wherein said equalizer 340 comprises a first finite impulse response (FIR) filter 341, wherein said first FIR filter 341 equalizes said first downsampled subband signal x₀₂, and wherein said equalizer 340 comprises a second FIR filter 342, wherein said second FIR filter 342 equalizes said second downsampled subband signal x₁₂, and wherein said equalizer 340 comprises a third FIR filter 343, wherein said third FIR filter 343 equalizes said third downsampled subband signal x₁₁. Thus, said equalization is performed in downsampled frequency subband domains, which reduces the computational complexity and the memory consumption of said equalization compared to an equalization performed on the full sampling rate and the full bandwidth.

In order to obtain the filter coefficients of said first FIR filter 341, a linear phase frequency-domain representation is formed according to a target subband magnitude transfer function, wherein said target subband magnitude transfer function is separated from the target equalizer magnitude response 100,203 within said first subband 200, and wherein the inverse discrete fourier transformation of said linear phase frequency-domain representation is calculated in order to obtain said filter coefficients of said first FIR filter 341. As depicted in FIG. 2, said target subband magnitude transfer function is represented by bands 1 to k of said target equalizer magnitude response 203. The coefficients of the remaining FIR filters 342, 343 may be calculated in the same way. This calculation of the filter coefficients is performed by the filter calculator 360, as depicted in FIG. 3. Therefore, the target EQ magnitude response may be fed to said filter calculation means. Furthermore, said target EQ magnitude response may by interactively obtained from a user via an interface 607, as depicted in FIG. 6.

In the following, the details of said delay module 350,351 for delaying of at least one of said downsampled subband signals will be explained in detail.

According to the first preferred embodiment, the length of said first FIR filter 341 may be larger than the length of said second FIR filter 342, as said first FIR filter 341 requires a higher order than said FIR filter 342 as there are more target equalizer bands in said first subband 200 than in said second subband 201 due to the logarithmic distribution of said target equalizer bands. Thus, a delay module 350 may be needed for delaying said second downsampled subband signal in order to compensate for the delay mismatch introduced by different group delays of said first FIR filter 341 and said second FIR filter 342. This step of delaying is depicted as step 703 in the flowchart in FIG. 7. For the present preferred embodiment, the use of symmetric, linear-phase FIR filters is suggested, which leads to a constant group delay for all frequencies for each of said symmetric, linear-phase FIR filters. Thus, a simple delay line 355 (without any fractional delays) is sufficient for the FIR delay compensations.

Furthermore, said non-symmetric tree structure of said analysis filters 301,302 and said synthesis filters 311,312 may introduce a delay mismatch between a first signal path and a second signal path, wherein said first signal path begins at a first output 321 of a first analysis filter 301, and wherein said first signal path end at a first input 331 of a first synthesis filter 311, and wherein said second signal path begins at a second output 322 of said first analysis filter 301, and wherein said second signal path ends at a second input 332 of said second synthesis filter 311, and wherein said first synthesis filter 311 corresponds to said first analysis filter 301 via said non-symmetric tree structure. Said second signal path comprises a second analysis filter 302 and a second synthesis filter 312, wherein said second analysis filter 302 and said second synthesis filter 312 introduce a group delay associated with said second signal path, which has to be compensated in said first signal path in order to reconstruct output signal y correctly by applying said first synthesis filter 311. To perform this compensation, a second delay module 351 is placed in said first signal path which comprises a group delay module 357 in order to delay said third downsampled subband signal. This step of delaying corresponds to step 703 in the flowchart in FIG. 7.

To avoid strong aliasing of downsampled signal components, it is crucial that the group delay has to be matched especially in the transition region of the QMF filter bank. If said second analysis filter 302 and said second synthesis filter 312 is represented by said QMF analysis and synthesis couple, wherein said QMF analysis and synthesis couple comprises said second order allpass filters, said group delay module 354 may be performed by implementing a filter with the transfer function

$T\left(z\right)=\frac{z^{-1}a_0\left(z^2\right)a_1\left(z^2\right)}{2}.$

Said transfer function

$T\left(z\right)=\frac{z^{-1}a_0\left(z^2\right)a_1\left(z^2\right)}{2}$

may also be used for obtaining an exact group delay compensation caused by non second order allpass filters. Filter with a different/simpler transfer function than the above transfer function T(z) may also be used for group delay compensation in the transition region of the QMF filter bank, which is not as complete as that of T(z), and does not necessarily compensate the group delay of the QMF bank elsewhere in the frequency domain but in the said transition region.

Furthermore, said first signal path may further be delayed by a delay module 356, which is located in said second delay module 351 in order to compensate for a delay mismatch between said third FIR filter 343 and said first FIR filter 341, and/or for compensating for a delay mismatch between said third FIR filter 343 and said second FIR filter 342 and said delay module 352. Furthermore, said group delay module 357 for compensating for the delay mismatch introduced by said QMF analysis and synthesis couple may also be performed by a delay line, but this introduces the drawback of reduced audio quality. In particular, when the group delay introduced by said QMF analysis and synthesis couple is an integer number at half the Nyquist frequency, said delay line may be used for performing said group delay module 357.

In particular, if a subband of a downsampled subband signal corresponds to a plurality of EQ bands, wherein said plurality of EQ bands has very different magnitude levels at adjacent bands (causing steep level changes close to the crossover frequencies between adjacent bands), the computational complexity of the corresponding equalizer increases. Thus, in this preferred embodiment said first FIR filter 341 has a higher computational complexity compared to the second FIR filter 342 or compared to the third FIR filter 343. Due to the present invention, the second FIR filter 342 and the third FIR filter 343 can be implemented with a low computational complexity, as there exists only two corresponding bands of the target EQ magnitude response 203 each associated with said second or said third subband, and, due to the wider bandwidths of the frequency bands that the second and the third FIR filters implement, the magnitude change between adjacent bands may occur over wider frequency range than in the lowest EQ bands, which is implemented by the first FIR filter.

FIG. 6 schematically depicts the main components of an audio device according to another preferred embodiment of the present invention, wherein said audio device comprises an interface 607 which may be used to obtain said target EQ magnitude interactively by a user. Furthermore said interface 607 may be a graphic user interface 607, which is able to display the actual target EQ magnitude response. For this case, this audio device for equalizing a digital audio signal represents a graphic audio equalizer, as it enables visual and interactive way of frequency balance modification of audio in real time. Said interface is connected to the filter calculator 606,360 which calculates and adjusts the filter coefficients of at least one FIR filter 605, wherein said at least one FIR filter 605 is located inside the equalizer 602.

The main advantage of the present invention are the reduced complexity and smaller memory requirements for the implementation compared to an implementation where the signal processing is performed at the full sampling rate. Furthermore, it enables a flexible design of the target EQ magnitude response with respect to the crossover frequencies and bandwidths of the EQ bands.

The invention has been described above by means of preferred embodiments. It should be noted that there are alternative ways and variations which are obvious to a skilled person in the art and can be implemented without deviating from the scope and spirit of the appended claims. In particular, the present invention is not restricted to equalization of an audio signal. It may equally well applied in systems that have to equalize any digital signal, for instance in order to equalize a received digital signal that has been distorted. Said distortion may for instance be caused by a transmission of said digital signal over an intersymbol-interference channel. Furthermore, it should be noted that the present invention is not restricted to non-symmetric tree structures concerning the separator and downsampler 300 and concerning the upsampler and combiner 310. As a matter of course also symmetric tree structures may be applied for the separator and downsampler 300 and for the upsampler and combiner 310. For instance, this symmetric tree structure could be used to separate the digital signal into a plurality of subsignals each having the same bandwidth. Further, at least one delay module 350, 351 may also be applied to at least one frequency branch in order to compensate for different group delays of different frequency branches.

While there have been shown and described and pointed out fundamental novel features of the invention as applied to preferred embodiments thereof, it will be understood that various omissions and substitutions and changes in the form and details of the devices and methods described may be made by those skilled in the art without departing from the spirit of the invention. For example, it is expressly intended that all combinations of those elements and/or method steps which perform substantially the same function in substantially the same way to achieve the same results are within the scope of the invention. Moreover, it should be recognized that structures and/or elements and/or method steps shown and/or described in connection with any disclosed form or embodiment of the invention may be incorporated in any other disclosed or described or suggested form or embodiment as a general matter of design choice. It is the intention, therefore, to be limited only as indicated by the scope of the claims appended hereto. Furthermore, in the claims means-plus-function clauses are intended to cover the structures described herein as performing the recited function and not only structural equivalents, but also equivalent structures. Thus although a nail and a screw may not be structural equivalents in that a nail employs a cylindrical surface to secure wooden parts together, whereas a screw employs a helical surface, in the environment of fastening wooden parts, a nail and a screw may be equivalent structures.

Claims

1. An apparatus comprising: a separator and downsampler for separating and downsampling a digital signal into at least two downsampled subband signals;an equalizer for equalizing at least one of said at least two downsampled subband signals; andan upsampler and combiner for upsampling and combining said at least two downsampled subband signals into a digital output signal.

2. The device according to claim 1, wherein said separator and downsampler comprises at least one analysis filter.

3. The device according to claim 2, wherein at least one of said at least one analysis filter is a quadrature mirror filter analysis filter.

4. The device according to claim 1, wherein said separator and downsampler comprise at least one synthesis filter.

5. The device according to claim 4, wherein at least one of said at least one synthesis filter is an quadrature mirror filter synthesis filter.

6. The device according to claim 1, wherein said digital signal is a digital audio signal.

7. The device according to claim 1, wherein said separator and downsampler comprises N analysis filters with N≧2, wherein said analysis filters are arranged in a non-symmetrical tree structure; and wherein said upsampler and combiner comprise N synthesis filters, wherein said synthesis filter are arranged in a non-symmetrical tree structure corresponding to said non-symmetrical tree structure of said N analysis filters.

8. The device according to claim 7, wherein at least one of said N analysis filters is a quadrature mirror filter analysis filter and wherein at least one of said N synthesis filters is a quadrature mirror filter synthesis filter.

9. The device according to claim 1, wherein said device comprises at least one delay module for delaying at least one of said at least two downsampled subband signals.

10. The device according to claim 9, wherein at least one of said at least one delay module comprises a group delay module.

11. The device according to claim 7, wherein a first of said N analysis filters comprises at least two outputs for outputting at least two digital signals, and wherein a first of said N synthesis filters comprises at least two inputs for inputting at least two digital signals, wherein said first synthesis filter corresponds to said first analysis filter via said non-symmetric tree structure; and wherein a first signal path begins at a first output of said at least two outputs of said first analysis filter, wherein said first signal path ends at a first input of said at least two inputs of said first synthesis filter, andwherein a second signal path begins at a second output of said at least two outputs of said first analysis filter, and wherein said second signal path ends at a second input of said at least two inputs of said first synthesis filter; andwherein said device comprises at least one delay module for delaying at least one of said at least two subband signals, wherein at least one of said at least one delay module comprises a group delay module, and wherein said group delay module is arranged for compensating for different group delays between said first signal path and said second signal path.

12. The device according to claim 11, wherein at least one of said at least one analysis filter is a quadrature mirror filter analysis filter, and wherein at least one of said at least one synthesis filter is a quadrature mirror filter synthesis filter.

13. The device according to claim 1, wherein said equalizer comprises at least one finite impulse response filter.

14. The device according to claim 13, wherein at least one of said at least one finite impulse response filter is a symmetric, linear-phase finite impulse response filter.

15. The device according to claim 3, wherein at least one of said at least one quadrature mirror filter analysis filter comprises first or higher order allpass filters.

16. The device according to claim 5, wherein at least one of said at least one quadrature mirror filter synthesis filter comprises first or higher order allpass filters.

17. The device according to claim 15, wherein at least one of said at least one quadrature mirror filter analysis filter comprises a first allpass filter and a second allpass filter, and wherein said at least one of said at least one quadrature mirror filter analysis filter is associated with a sampling rate FS, and wherein the magnitude response of the low-frequency branch of said at least one of said at least one quadrature mirror filter analysis filter has a stopband edge frequency fst,L relatively close to FS/4, and wherein the magnitude response of the high-frequency branch of said at least one of said at least one quadrature mirror filter analysis filter has a stopband edge frequency fst,H relatively close to FS/4.

18. The device according to claim 15, wherein at least one of said at least one quadrature mirror filter analysis filter comprises a first allpass filter and a second allpass filter, and wherein said at least one of said at least one quadrature mirror filter analysis filter is associated with a sampling rate FS, and wherein the magnitude response of the low-frequency branch of said at least one of said at least one quadrature mirror filter analysis filter has a stopband edge frequency fst,L≈0.316·FS, and wherein the magnitude response of the high-frequency branch of said at least one of said at least one quadrature mirror filter analysis filter has a stopband edge frequency fst,H≈0.184·FS.

19. The device according to claim 16, wherein at least one of said at least one quadrature mirror filter synthesis filter comprises a first allpass filter and a second allpass filter, and wherein said at least one of said at least one quadrature mirror filter synthesis filter is associated with a sampling rate FS, and wherein the magnitude response of the low-frequency branch of said at least one of said at least one quadrature mirror filter synthesis filter has a stopband edge frequency fst,L relatively close to FS/4, and wherein the magnitude response of the high-frequency branch of said at least one of said at least one quadrature mirror filter synthesis filter has a stopband edge frequency fst,H relatively close to FS/4.

20. The device according to claim 16, wherein at least one of said at least one quadrature mirror filter synthesis filter comprises a first allpass filter and a second allpass filter, and wherein said at least one of said at least one quadrature mirror filter synthesis filter is associated with a sampling rate FS, and wherein the magnitude response of the low-frequency branch of said at least one of said at least one quadrature mirror filter synthesis filter has a stopband edge frequency fst,L≈0.316·FS, and wherein the magnitude response of the high-frequency branch of said at least one of said at least one quadrature mirror filter synthesis filter has a stopband edge frequency fst,H≈0.184·FS.

21. The device according to claim 1, wherein said separator and downsampler said digital signal comprises at least one analysis filter, and wherein said upsampler and combiner for upsampling and combining said digital signal comprise at least one synthesis filter; and wherein at least one of said at least one analysis filter is a quadrature mirror filter analysis filter; and wherein at least one of said at least one synthesis filter is a quadrature mirror filter synthesis filter; and wherein at least one of said at least one quadrature mirror filter analysis filter comprises a first second order allpass filter and a second second order allpass filter, wherein said first second order allpass filter has a first transfer function a0(z) and said second second order allpass filter has a second transfer function a1(z), andwherein at least one of said at least one quadrature mirror filter synthesis filter comprises a third second order allpass filter and a fourth second order allpass filter, wherein said third second order allpass filter has said first transfer function a0(z) and said fourth second order allpass filter has said second transfer function a1(z),wherein said first second order allpass filter, said second second order allpass filter, said third second order allpass filter and said fourth second order allpass filter are polyphase components of 9th order elliptic filters whose poles are on the imaginary axis.

22. The device according to claim 12, wherein at least one of said at least one quadrature mirror filter analysis filter comprises a first second order allpass filter and a second second order allpass filter, wherein said first second order allpass filter has a first transfer function a0(z) and said second second order allpass filter has a second transfer function a1(z), and wherein at least one of said at least one quadrature mirror filter synthesis filter comprises a third second order allpass filter and a fourth second order allpass filter, wherein said third second order allpass filter has said first transfer function a0(z) and said fourth second order allpass filter has said second transfer function a1(z),wherein said first second order allpass filter, said second second order allpass filter, said third second order allpass filter and said fourth second order allpass filter are polyphase components of 9th order elliptic filters whose poles are on the imaginary axis, andwherein said at least one of said at least one quadrature mirror filter analysis filter corresponds to said at least one of said at least one quadrature mirror filter synthesis filter via said non-symmetric tree structure; andwherein at least one of said at least one group delay module has the following transfer function:

23. The device according to claim 21, wherein the magnitude response of a low-frequency branch of said at least one of said at least one quadrature mirror filter analysis filter has a stopband edge frequency fst,L≈0.316·FS, and wherein the magnitude response of a high-frequency branch of said at least one of said at least one quadrature mirror filter analysis filter has a stopband edge frequency fst,H≈0.184·FS, wherein FS denotes the sampling rate associated with said at least one of said at least one quadrature mirror filter analysis filter; and wherein the magnitude response of a low-frequency branch of said at least one of said at least one quadrature mirror filter synthesis filter has a stopband edge frequency fst,L≈0.316·FS, and wherein the magnitude response of a high-frequency branch of said at least one of said at least one quadrature mirror filter synthesis filter has a stopband edge frequency fst,H≈0.184·FS, wherein FS denotes the sampling rate associated with said at least one of said at least one quadrature mirror filter synthesis filter.

24. The device according to claim 21, wherein the magnitude response of a low-frequency branch of said at least one of said at least one quadrature mirror filter analysis filter has a stopband edge frequency fst,L relatively close to FS/4, and wherein the magnitude response of a high-frequency branch of said at least one of said at least one quadrature mirror filter analysis filter has a stopband edge frequency fst,H relatively close to FS/4, wherein FS denotes the sampling rate associated with said at least one of said at least one quadrature mirror filter analysis filter; and wherein the magnitude response of a low-frequency branch of said at least one of said at least one quadrature mirror filter synthesis filter has a stopband edge frequency fst,L relatively close to FS/4, and wherein the magnitude response of a high-frequency branch of said at least one of said at least one quadrature mirror filter synthesis filter has a stopband edge frequency fst,H relatively close to FS/4, wherein FS denotes the sampling rate associated with said at least one of said at least one quadrature mirror filter synthesis filter.

25. The device according to claim 13, wherein said device comprises a filter calculator for calculating the filter coefficients of said at least one finite impulse response filter by using a target equalizer magnitude response, and wherein said filter calculator is fed with said target equalizer magnitude response.

26. The device according to claim 25, wherein a first finite impulse response filter of said at least one finite impulse response filter is associated with a first set of filter coefficients, wherein said first finite impulse response filter equalizes a first of said at least two downsampled subband signals; and wherein said filter calculator calculates said first set of filter coefficients by forming a linear phase frequency-domain representation according to a target subband magnitude transfer function, wherein said target subband magnitude transfer function is separated from said target equalizer magnitude response within a frequency band corresponding to said first subband signal, and wherein the inverse discrete fourier transformation of said linear phase frequency-domain representation is calculated in order to obtain said first set of filter coefficients.

27. The device according to claim 1, wherein said separator and downsampler comprises N analysis filters with N≧1, wherein said analysis filters are arranged in a symmetrical tree structure; and wherein said upsampler and combiner comprise N synthesis filters, wherein said synthesis filters are arranged in a symmetrical tree structure corresponding to said symmetrical tree structure of said N analysis filters.

28. The device according to claim 27, wherein at least one of said N analysis filters is a quadrature mirror filter analysis filter, and wherein at least one of said N synthesis filters is a quadrature mirror filter synthesis filter.

29. The device according to claim 1, wherein said equalizer comprises at least one infinite impulse response filter.

30. A method: separating and downsampling a digital signal into at least two downsampled subband signals;equalizing at least one of said at least two downsampled subband signals; andupsampling and combining said at least two downsampled subband signals into a digital output signal.

31. The method according to claim 30, wherein said separating and downsampling comprises analysis filtering.

32. The method according to claim 31, wherein said analysis filtering comprises quadrature mirror filter analysis.

33. The method according to claim 30, wherein said upsampling and combining comprises synthesis filtering.

34. The method according to claim 33, wherein said synthesis filtering comprises quadrature mirror filter synthesis.

35. The method according to claim 30, wherein said digital signal is a digital audio signal.

36. The method according to claim 30, wherein said separating and downsampling comprises N times analysis filtering with N≧2, wherein said N times analysis filtering being performed according to a non-symmetrical tree structure; and wherein said upsampling and combining comprises N times synthesis filtering, wherein said N times synthesis filtering being performed according to a non-symmetrical tree structure according to said non-symmetrical tree structure of said N times analysis filtering.

37. The method according to claim 36, wherein said analysis filtering comprises quadrature mirror filter analysis, and wherein said synthesis filtering comprises quadrature mirror filter synthesis.

38. The method according to claim 30, wherein said method comprises delaying of at least one of said at least two downsampled subband signals.

39. The method according to claim 38, wherein said delaying comprises group delaying.

40. The method according to claim 36, wherein said method comprises delaying of at least one of said at least two downsampled subband signals, and wherein said delaying comprises group delaying, wherein said group delaying is performed to compensate different group delays caused by said non-symmetric tree structure of said N times analysis filtering and the corresponding non-symmetric tree structure of said N times synthesis filtering.

41. The method according to claim 40, wherein said analysis filtering comprises quadrature mirror filter analysis, and wherein said synthesis filtering comprises quadrature mirror filter synthesis.

42. The method according to claim 30, wherein said equalizing comprises finite impulse response filtering.

43. The method according to claim 42, wherein said finite impulse response filtering comprises linear-phase Finite Impulse filtering, and wherein the filter coefficients used for said linear-phase finite impulse response filtering are symmetric.

44. The method according to claim 32, wherein said quadrature mirror filter analysis comprises first or higher order allpass filtering.

45. The method according to claim 34, wherein said quadrature mirror filter synthesis comprises first or higher order allpass filtering.

46. The method according to claim 44, wherein said quadrature mirror filter analysis comprises a first quadrature mirror filter analysis, and wherein said first quadrature mirror filter analysis is associated with a sampling rate FS, and wherein said first quadrature mirror filter analysis comprises allpass filtering for obtaining a stopband edge frequency fst,L relatively close to FS/4 in the magnitude response of a low-frequency branch of said first quadrature mirror filter analysis and for obtaining a stopband edge frequency fst,H relatively close to FS/4 in the magnitude response of a high-frequency branch of said first quadrature mirror filter analysis.

47. The method according to claim 44, wherein said quadrature mirror filter analysis comprises a first quadrature mirror filter analysis, and wherein said first quadrature mirror filter analysis is associated with a sampling rate FS, wherein said first quadrature mirror filter synthesis comprises allpass filtering for obtaining a stopband edge frequency fst,L≈0.316·FS in the magnitude response of a low-frequency branch of said first quadrature mirror filter analysis and for obtaining a stopband edge frequency fst,H≈0.184·FS in the magnitude response of a high-frequency branch of said first quadrature mirror filter analysis.

48. The method according to claim 45, wherein said quadrature mirror-filter synthesis comprises a first quadrature mirror filter synthesis, and wherein said first quadrature mirror filter synthesis is associated with a sampling rate FS, and wherein said first quadrature mirror filter synthesis comprises allpass filtering for obtaining a stopband edge frequency fst,L relatively close to FS/4 in the magnitude response of a low-frequency branch of said first quadrature mirror filter synthesis and for obtaining a stopband edge frequency fst,H relatively close to FS/4 in the magnitude response of a high-frequency branch of said first quadrature mirror filter synthesis.

49. The method according to claim 45, wherein said quadrature mirror filter synthesis comprises a first quadrature mirror filter synthesis, and wherein said first quadrature mirror filter synthesis is associated with a sampling rate FS, and wherein said first quadrature mirror filter synthesis comprises allpass filtering for obtaining a stopband edge frequency fst,L≈0.316·FS in the magnitude response of a low-frequency branch of said first quadrature mirror filter synthesis and for obtaining a stopband edge frequency fst,H≈0.184·FS in the magnitude response of a high-frequency branch of said first quadrature mirror filter synthesis.

50. The method according to claim 30, wherein said separating and downsampling comprises analysis filtering, and wherein said analysis filtering comprises quadrature mirror filter analysis, and wherein said upsampling and combining comprises synthesis filtering, and wherein said synthesis filtering comprises quadrature mirror filter synthesis; and wherein said quadrature mirror filter analysis comprises a first quadrature mirror filter analysis, wherein said first quadrature mirror filter analysis comprises a first second order allpass filtering and a second second order allpass filtering, wherein said first second order allpass filtering being performed by a first transfer function a0(z), and wherein said second second order allpass filtering being performed by a second transfer function a1(z); andwherein said quadrature mirror filter synthesis comprises a first quadrature mirror filter synthesis, wherein said first quadrature mirror filter synthesis comprises a third second order allpass filtering and a fourth second order allpass filtering, wherein said third second order allpass filtering being performed by said first transfer function a0(z), and wherein said fourth second order allpass filtering being performed by said second transfer function a1(z); andwherein said transfer functions a0(z) and a1(z) represent second order allpass filters with polyphase components of 9th order elliptic filters whose poles are on the imaginary axis.

51. The method according to claim 41, wherein said quadrature mirror filter analysis comprises a first quadrature mirror filter analysis, wherein said first quadrature mirror filter analysis comprises a first second order allpass filtering and a second second order allpass filtering, wherein said first second order allpass filtering being performed by a first transfer function a0(z), and wherein said second second order allpass filtering being performed by a second transfer function a1(z); and wherein said quadrature mirror filter synthesis comprises a first quadrature mirror filter synthesis, wherein said first quadrature mirror filter synthesis comprises a third second order allpass filtering and a fourth second order allpass filtering, wherein said third second order allpass filtering being performed by said first transfer function a0(z), and wherein said fourth second order allpass filtering being performed by a second transfer function a1(z); andwherein said transfer functions a0(z) and a1(z) represent second order allpass filters with polyphase components of 9th order elliptic filters whose poles are on the imaginary axis; andwherein said first quadrature mirror filter analysis corresponds to said first quadrature mirror filter synthesis via said non-symmetric tree structure; andwherein said group delaying is performed by filtering, wherein said filtering corresponds to the following transfer function:

52. The method according to claim 51, wherein said first quadrature mirror filter analysis is associated with a sampling rate FS, and wherein the magnitude response of a low-frequency branch of said first quadrature mirror filter analysis has a stopband edge frequency fst,L≈0.316·FS, and wherein the magnitude response of a high-frequency branch of said first quadrature mirror filter analysis has a stopband edge frequency fst,H≈0.184·FS; and wherein said first quadrature mirror filter synthesis is associated with a sampling rate FS, wherein the magnitude response of a low-frequency branch of said first quadrature mirror filter synthesis has a stopband edge frequency fst,L≈0.316·FS, and wherein the magnitude response of a high-frequency branch of said first quadrature mirror filter analysis has a stopband edge frequency fst,H≈0.184·FS.

53. The method according to claim 51, wherein said first quadrature mirror filter analysis is associated with a sampling rate FS, wherein the magnitude response of a low-frequency branch of said first quadrature mirror filter analysis has a stopband edge frequency fst,L close to FS/4, and wherein the magnitude response of a high-frequency branch of said first quadrature mirror filter analysis has a stopband edge frequency fst,H close to FS/4; and wherein said first quadrature mirror filter synthesis is associated with a sampling rate FS, wherein the magnitude response of a low-frequency branch of said first quadrature mirror filter synthesis has a stopband edge frequency fst,L close to FS/4, and wherein the magnitude response of a high-frequency branch of said first quadrature mirror filter synthesis has a stopband edge frequency fst,H close to FS/4.

54. The method according to claim 30, wherein said separating and downsampling comprises N times analysis filtering with N≧1, wherein said N times analysis filtering being performed according to a symmetrical tree structure; and wherein said upsampling and combining comprises N times synthesis filtering, wherein said N times synthesis filtering being performed according to a symmetrical tree structure according to said symmetrical tree structure of said N times analysis filtering.

55. The method according to claim 54, wherein said analysis filtering comprises quadrature mirror filter analysis, and wherein said synthesis filtering comprises quadrature mirror filter synthesis.

56. The method according to claim 42, wherein said finite impulse response filtering comprises a first finite impulse response filtering associated with a first set of filter coefficients, wherein said first finite impulse response filtering equalizes a first subband signal of said at least two downsampled subband signals, wherein a linear phase frequency-domain representation is formed according to a target subband magnitude transfer function, wherein said target subband magnitude transfer function is separated from a target equalizer magnitude response within a frequency band corresponding to said first subband signal, and wherein the inverse discrete fourier transformation of said linear phase frequency-domain representation is calculated in order to obtain said first set of filter coefficients.

57. The method according to claim 42, wherein said finite impulse response filtering comprises a first finite impulse response filtering associated with a first set of filter coefficients, wherein said first finite impulse response filtering equalizes a first of said at least two downsampled subband signals, wherein a linear phase frequency-domain representation is formed according to a target subband magnitude transfer function, wherein said target subband magnitude transfer function is separated from a target equalizer magnitude response within a frequency band corresponding to said first subband signal, and wherein the Remez filter design algorithm is applied to said linear phase frequency-domain representation in order to calculate said first set of filter coefficients.

58. The method according to claim 56, wherein said target equalizer magnitude response is separated into n subbands in the frequency domain with n≧2.

59. The method according to claim 58, wherein said separating and downsampling comprises analysis filtering, and wherein said analysis filtering comprises quadrature mirror filter analysis, and wherein said upsampling and combining comprises synthesis filtering, and wherein said synthesis filtering comprises quadrature mirror filter synthesis; and wherein said first quadrature mirror filter synthesis corresponds to said first quadrature mirror filter synthesis; andwherein said first quadrature mirror filter analysis and said first quadrature mirror filter synthesis are associated with a sampling rate FS, and wherein the magnitude response of a low frequency branch of said quadrature mirror filter analysis and synthesis has a stopband edge frequency fst,L≧FS/4, and wherein the magnitude response of a high frequency branch of said quadrature mirror filter analysis and synthesis has the stopband edge frequency fst,H≦FS/4; andwherein said target equalizer magnitude response is constant in the frequency region between fst,H and fst,L.

60. The method according to claim 59, wherein said n subbands of said target equalizer magnitude response correspond to n−1 crossover frequencies, and wherein said n−1 crossover frequencies are arranged so that none of said n−1 crossover frequencies lies in said frequency region between fst,H and fst,L.

61. The method according to claim 60, wherein said n subbands of said target equalizer magnitude response are distributed logarithmically.

62. The method according to claim 30, wherein said equalizing comprises finite impulse response filtering.

63. A computer program product for equalizing a digital signal comprising program code stored on a readable medium for execution by a processor, such that when executed said program code: separates and downsamples said digital signal into at least two downsampled subband signals; andequalizes at least one of said at least two downsampled subband signals; andupsamples and combines said at least two downsampled subband signals into a digital output signal.

64. An audio device comprising a device according to claim 1.

65. The audio device according to claim 64, wherein said equalizer comprises at least one finite impulse response filter; and wherein said audio device comprises a filter calculator for calculating the filter coefficients of said at least one finite impulse response filter by using a target equalizer magnitude response,wherein said audio device comprises a user interface in order to obtain said target equalizer magnitude response, wherein said user interface is connectable to said filter calculator to transmit said target equalizer magnitude response to said filter calculator.

66. The audio device according to claim 64, wherein said equalizer comprises at least one infinite impulse response filter; and wherein said audio device comprises a filter calculator for calculating the filter coefficients of said at least one infinite impulse response filter by using a target equalizer magnitude response,wherein said audio device comprises a user interface in order to obtain said target equalizer magnitude response, wherein said user interface is connectable to said filter calculator to transmit said target equalizer magnitude response to said filter calculator.

67. An apparatus comprising: means for separating and downsampling said digital signal into at least two downsampled subband signals;means for equalizing at least one of said at least two downsampled subband signals; andmeans for upsampling and combining said at least two downsampled subband signals into a digital output signal.

68. The apparatus according to claim 67, wherein said separator and downsampler comprises at least one analysis filter.