Patent Application
20150009505
The invention relates to a fiber-optic measurement device allowing to measure the variation of a parameter that produces non-reciprocal disturbances in a SAGNAC ring interferometer.
The SAGNAC interferometer and the physical phenomena involved thereby are well known. Reference may be made for example about that to “The Fiber-Optic Gyroscope”, H. Lefèvre (Artech House, 1993).
In such an interferometer, a splitting plate or any other splitting device splits an incident wave at the input of the interferometer into two waves. The two thus-created waves are referred to as “counter-propagating waves”. They indeed propagate in opposite directions along a same closed optical path, then recombine with each other, producing interferences at the time of their recombination. The interference state between the two counter-propagating waves then depends on the relative phase difference between them. The luminous power P measured at the output of a SAGNAC interferometer is of the form: P(Δφ)=P0[1+cos(Δφ)], where Δφ is the relative phase difference between the two counter-propagating waves. Hence, the power measured at the output of the interferometer takes values between a minimum (it is then talked about “dark” fringe) and a maximum (“bright” fringe) as a function of the value of the phase difference Δφ.
It is known that some physical phenomena are liable to introduce so-called non-reciprocal phase shifts, in the counter-propagating waves, hence generating a phase difference Δ p between these waves and modifying the interference state during the recombination thereof. Hence, the measurement of this non-reciprocal phase shift Δφp allows to quantify the phenomenon that has been generated thereby.
The main physical phenomenon liable to create non-reciprocal disturbances is the SAGNAC effect produced by the rotation of the interferometer about an axis perpendicular to the plane of its closed optical path. A second effect, the FARADAY effect—or collinear magneto-optic effect—is also known for producing non-reciprocal effects of this type.
It is known that a SAGNAC interferometer can include a fiber-optic coil, which is preferably single-mode and of the polarization-maintaining type. The multiple turns of an optical fiber form a closed optical path of very long length, up to several kilometres.
A proper frequency fp of the SAGNAC interferometer is commonly defined. The proper frequency fp of a SAGNAC ring interferometer including a single-mode fiber-optic coil (silica fiber having a refractive index close to 1.5 in the operating wavelength range) of 1 kilometre long is of the order of 100 kilohertz (kHz). The extension of the coil length and hence of the optical path has for advantage to provide the interferometer with a greater sensitivity.
Moreover, it has been shown that the measurement accuracy is improved by the use of a so-called “phase cancellation” method, also called closed-loop operation, instead of a simple open-loop operation.
According to this method, an additional so-called “feedback” phase difference Δφcr is introduced by means of a phase modulator between the two counter-propagating waves, so as to compensate for the phase shift Δφp produced by the parameter measured. The sum of the two phase-shifts Δφp and Δφcr is kept at zero, which allows to make the interferometer operate with a better accuracy. The measurement of the parameter to be measured is made thanks to the use of the signal necessary to the production of the feedback phase difference Δφcr.
However, the sensitivity of the response P(Δφ) of the interferometer in the vicinity of the zero phase difference (Δφ=0) is low, because the signal measured at the output of the interferometer is a cosine-wave function of the phase difference Δφ.
It is known that it is possible to displace the operating point of the interferometer towards a point offering a greater sensitivity. It has notably been proposed to introduce an additional so-called “biasing” phase-difference modulation Δφb, by means of the phase modulator. The total phase difference Δφt between the two counter-propagating waves is then equal to the sum of the different phase differences: Δφt=Δφp+Δφcr+Δφb.
A simple-to-implement solution to perform this biasing consists in a square-wave periodic modulation at a biasing modulation frequency fb, the modulation having for example levels +π/2 and −π/2. This biasing phase-difference modulation Δφb produces at the output of the interferometer a square-wave periodic modulated electrical signal at the biasing modulation frequency fb whose amplitude is a sine-wave function of the sum of the two phase-shifts Δφp and Δφcr, in the case of a closed-loop measurement as described above. The response provided by the SAGNAC interferometer can hence be used with a greater sensitivity.
Moreover, in order to improve the stability of the measurement of a non-reciprocal parameter by means of a SAGNAC interferometer, the document EP0430747 proposes a device in which the biasing phase-difference modulation MOO introduced between the two counter-propagating waves is periodic at the frequency fb.
At each period of the modulation, the level of the phase-difference modulation Δφb(t) is hence equal to:
The values of a and φ0 are chosen so as to verify the relation: cos(φ0)=cos(aφ0).
The device according to the document EP0430747 also includes a signal processing system using the four values taken by the modulated electrical signal delivered by the interferometer during one modulation period. The signal processing system then allows to maintain constant the gain of the modulation chain so as to compensate for the slow drifts of the different components of the device (for example: variation as a function of the temperature).
To reduce the effects of the modulation chain defects on the measurement, it is known that the biasing modulation frequency fb has to be equal to the proper frequency fp of the interferometer or to one of its odd multiples.
In particular, the so-called “four states” modulation generated by the biasing described in the document EP0430747 introduces defect-bearing peaks on the modulated electrical signal measured at the output of the interferometer, these defects being eliminated when the biasing modulation frequency fb is equal to the proper frequency fp of the SAGNAC interferometer, or to one of its odd multiples.
Moreover, the number of these peaks increases with the biasing modulation frequency fb.
To reduce the response time of the interferometer and to ensure that the feedback loop of measurement does not break in case of a rapid variation of the parameter to be measured, the biasing modulation frequency fb is increased. However, the measurement accuracy is hence substantially degraded due to a greater number of peaks in the signal detected.
The object of the present invention is to propose a fiber-optic measurement device wherein a parameter to be measured generates a phase difference between two counter-propagating waves, wherein the response time is improved while keeping a good accuracy over the measurement range.
For that purpose, the invention relates to a fiber-optic measurement device of the type in which a parameter to be measured generates a phase difference Δφp between two counter-propagating waves, including:
According to the invention, said fiber-optic measurement device is characterized in that said biasing means are adapted to generate said biasing signal such that said phase modulator generates a biasing phase-shift modulation φb(t), when it is subjected to said biasing modulation voltage Vb(t), said biasing phase-shift modulation φb(t) being the sum of:
Hence, said device according to the invention allows to perform a “biasing” around π, thanks to said first biasing phase-difference modulation component Δφb1(t) offering an optimal signal-to-noise ratio for the detection chain. Said device also allows to increase the frequency of demodulation of the signal function of said phase difference Δφp and of said parameter to be measured thanks to said second biasing phase-difference modulation component Δφb2(t) to reduce the response time of the interferometer without thereby increasing the number of cumbersome peaks in the modulated electrical signal. The stability of the closed loop is hence improved, that is to say that the measurement device according to the invention is capable of measuring a parameter generating a non-reciprocal effect, even if said parameter to be measured varies very rapidly.
Moreover, other advantageous and non-limitative characteristics of the device according to the invention are as follows:
The measurement device according to the invention is particularly well adapted for the making of a gyrometer. In this case, the parameter to be measured is a component of the rotational speed of the ring interferometer.
Hence, the invention also relates to a gyrometer, characterized in that it is compliant with the fiber-optical measurement device according to the invention, the parameter to be measured being a component of the rotational speed of the ring interferometer.
This gyrometer advantageously enters into the making of navigation or inertial-stabilization systems.
Hence, the invention also proposes a navigation or inertial-stabilization system including at least one gyrometer according to the invention.
Embodiments of the invention will be described in detail with reference to the drawings in which:
The fiber-optic measurement device 10 first includes a light source 11 herein comprising a laser diode.
As a variant, the light source may comprise, for example, a super-luminescent diode or a doped-fiber light source of the “ASE” (“Amplified Spontaneous Emission”) type.
The device 10 also comprises a first splitting element 12. This first splitting element 12 is herein a semi-reflective plate having a transmittance of 50% and a reflectance of 50%.
As a variant, the splitting element may, for example, be a 3-decibel 2x2 coupler or an optical circulator.
The luminous wave emitted by the light source 11 is hence transmitted in part by the first splitting element 12 towards an optical filter 13 at the output of which the luminous wave has been filtered. The optical filter 13 preferably includes a polarizer and a spatial filter. This spatial filter is herein a single-mode optical fiber, preferably of the polarization-maintaining type.
The device 10 also includes a SAGNAC ring interferometer 20 comprising a fiber-optic coil 21 wound around itself. It is herein an optical fiber, preferably of the single-mode and polarization-maintaining type.
This SAGNAC ring interferometer 20 also comprises a second splitting element 23 allowing to split the wave exiting from the optical filter 13 into two counter-propagating waves 24, 25 on the two “arms” of the ring interferometer 20, these two arms defining two optical paths 24A and 25A. The second splitting element 23 is herein a semi-reflective plate having a transmittance of 50% and a reflectance of 50%.
The second splitting element 23 also allows to recombine the two counter-propagating waves 24, 25 at the output of the ring interferometer 20.
As a variant, the second splitting element may be, for example, a 3-decibel 2×2 coupler or a “Y”-junction in integrated optics.
The two counter-propagating waves 24, 25 then pass through the optical filter 13 and are reflected by the first splitting element 12 towards an electromagnetic radiation detector 14.
This detector 14 is herein a semi-conductor photodiode.
The detector 14 is sensitive to the luminous power received, which is herein function of the interference state between the two counter-propagating waves 24, 25 during their recombination at the output of the SAGNAC ring interferometer 20. It hence delivers an electrical signal that is representative of the total phase difference Δφt between the two counter-propagating waves 24, 25. It will be seen in the following of the description that this electrical signal is a modulated electrical signal.
The device 10 also includes a modulation chain 30 comprising a digital/analog converter 31, an amplifier 32 and a phase modulator 33.
The digital/analog converter 31 processes a digital control signal delivered by the control means 140, the decomposition of this signal being described in detail hereinafter. The digital/analog converter 31 delivers as an output an analog control signal.
The amplifier 32 then processes this analog control signal to deliver a modulation control voltage Vm(t) to the phase modulator 33.
The phase modulator 33 is placed in the ring interferometer 20 and is thus also a part thereof. It is herein arranged at one end of the optical path of the SAGNAC ring interferometer 20. The phase modulator 33 is herein of the electro-optical type (said of “Pockels effect” type) in proton-exchange lithium-niobate integrated optics.
The phase modulator 33 allows, when the time-dependant control modulation voltage Vm(t) is applied at the input thereof, to generate a phase-shift modulation φm(t) that is proportional, and thus with the same time dependency, on a luminous wave passing through it at the given instant t in one direction or another.
In the case of the SAGNAC ring interferometer 20 shown in
Hence, the phase-shift modulation φm(t) generated by the phase modulator 33 controlled by the modulation control voltage Vm(t) introduces between the two counter-propagating waves 24, 25, a phase-difference modulation Δφm(t) such that: Δφm(t)=φm(t)−φm(t−Δτg).
The transit-time difference Δτg also defines a proper frequency fp of the SAGNAC ring interferometer 20 by the relation: fp=1/(2Δτg).
This proper frequency fp thus depends on the length of the coil 21 in the SAGNAC ring interferometer 20. With the fiber-optic coil 21 used herein, a coil having a length of 1 kilometre, the proper frequency fp of the SAGNAC ring interferometer 20 is of about 100 kilohertz (kHz), corresponding to a transit-time difference Δτg of 5 microseconds (μs).
The luminous power P(Δφ1) received by the detector 14 is also modulated and the electrical signal delivered by the detector 14 will thus be a modulated electrical signal (38), examples of which are given in
The modulated electrical signal 38 is transmitted to electronic means 100 that process it to deliver a signal function of the phase difference Δφp and of the parameter to be measured.
For that purpose, the electronic means 100 comprise signal processing means 110, as shown in
This digitization operation is performed at a synchronization frequency fixed by the clock 101. The synchronization frequency of the clock 101 is preferably a multiple of the proper frequency fp of the SAGNAC ring interferometer 20.
The signal processing means 110 also comprise a digital processing unit 112 configured to process the digital electrical signal provided at the output of the analog/digital converter 111. The digital processing unit 112 also includes a digital demodulator, a control-loop digital filter fed with a first demodulated digital signal exiting from the digital demodulator and a register.
The digital processing unit 112 delivers a signal function of the phase difference Δφp and of the parameter to be measured for any desired external use.
The electronic means 100 also control in return the modulation chain 30.
For that purpose, the electronic means 100 include, on the one hand, feedback means 120 and, on the other hand, biasing means 130.
The feedback means 120 receive as an input the signal function of the phase difference Δφp of the parameter to be measured provided by the digital processing unit 112. The feedback means 120 generate as an output a feedback signal whose action on the modulation chain 30 will be described in more detail hereinafter.
The feedback means 120 herein include an accumulator.
The biasing means 130 are configured to generate a biasing signal at precise instants, synchronized by the frequency of the clock 101. The action of this biasing signal on the modulation chain 30 will be described in more detail hereafter.
The electronic means 100 further include control means 140 that have two inputs and one output. The control means 140 receive as an input, on the one hand, the feedback signal, and on the other hand, the biasing signal. These signals are then processed by the control means 140. The control means 140 deliver as an output a digital control signal that is then transmitted to the digital/analog converter 31 of the modulation chain 30.
The control means 140 herein include a digital adder. The operation performed by the control means 140 consists in the digital addition of the feedback signal provided by the feedback means 120 and of the biasing signal provided by the biasing means 130. The digital control signal is the signal resulting from this addition.
The digital control signal is then transmitted to the modulation chain 30. It is converted into an analog control signal by the digital/analog converter 31, then transmitted to the amplifier 32 that delivers a modulation control voltage Vm(t) to the phase modulator 33.
The modulation chain 30 thus receives as a input the digital control signal and produces as an output, by means of the phase modulator 33, a phase-shift modulation φm(t) modulated in time, which will be introduced in the counter-propagating waves 24, 25 propagating in the SAGNAC ring interferometer 20.
The modulation chain 30 is then characterized electronically by its transfer function between the input and the output. This transfer function is the ratio between the value (in radians) of the phase-shift effectively generated by the modulation chain 30 via the phase modulator 33 and the value (with no unity) of the digital control signal transmitted to the modulation chain 30.
In order to keep the transfer function of the modulation chain 30 adjusted, the electronic means 100 also include gain control means 150.
These gain control means 150 include another digital processing unit (not shown) using the digital electrical signal delivered by the analog/digital converter 111 so as to provide a signal function of the transfer function of the modulation chain 30.
This signal is filtered by a control-loop digital integrating filter that feeds another digital/analog converter controlling the variable gain G of the amplifier 32 or the analog reference voltage of the digital/analog converter 31. Hence, the transfer function of the modulation chain 30 is kept correctly adjusted, as well as the modulation control voltage Vm(t) delivered by the amplifier 32 to the phase modulator 33.
It is meant by this that a given value of the digital control signal at the input of the modulation chain 30 will always give the same value (in radians) of phase-shift modulation φm generated by the phase modulator 33, and hence the same value (in radians) of the phase-difference modulation Δφm introduced between the two counter-propagating waves 24, 25 in the SAGNAC ring interferometer 20.
The digital control signal being the sum of the feedback signal and the biasing signal, the modulation control voltage Vm(t) at the input of the phase modulator 33 is decomposed into the sum of a feedback modulation voltage Vcr(t) and a biasing modulation voltage Vb(t).
The feedback modulation voltage Vcr(t) at the input of the phase modulator 33 results, at the output of the phase modulator 33, in a feedback phase-shift modulation φcr(t) on the wave passing through it.
In the case of the SAGNAC ring interferometer 20, the effect of this feedback phase-shift modulation φcr(t) is the introduction of a feedback phase-difference modulation Δφcr(t) between the two counter-propagating waves 24, 25, allowing to compensate for the phase difference Δφp generated by the parameter to be measured and hence to keep at zero the sum of the phase difference Δφp and the phase difference Δφcr.
This feedback allows to make the device 10 operate in closed loop so as to reach a good linearity and stability of the measurement of the parameter generating the phase difference Δφp.
According to the prior art,
The feedback signal generated by the feedback means 120 is a stair-step digital signal.
For a SAGNAC ring interferometer 20, of proper frequency fp, the prior art teaches a duration of Δτg for each step, the passage from one step to another being synchronized thanks to the clock 101 present in the electronic means 100.
This is translated in
Likewise, as described hereinabove, the height of the step is such that the phase-difference modulation Δφcr(t) introduced between the two counter-propagating waves 24, 25 compensate for the phase difference Δφp due to the parameter measured.
Moreover, the feedback phase-shift modulation φcr(t) is a stair-step ramp modulation such that this modulation falls down by 2π, as shown in
It is known that this “falling down to 2π” is made necessary by the fact that the value of the feedback modulation voltage Vcr(t) cannot increase indefinitely. The use of digital means, such as the digital/analog converter 31, allows to make simply this falling down to 2π.
As mentioned above, the biasing means 130 are configured to generate a biasing signal, this biasing signal being transmitted to the control means 140 piloting the modulation chain 30.
This biasing signal is associated with the biasing modulation voltage Vb(t), through the digital/analog converter 31 and the amplifier 32.
This biasing modulation voltage Vb(t) at the output of the amplifier 32 and at the input of the phase modulator 33 results at the output of the phase modulator 33 in a biasing phase-shift modulation φb(t) on a wave passing through it.
In the case of the SAGNAC ring interferometer 20, the effect of this biasing phase-shift modulation φb(t) is the introduction of a biasing phase-difference modulation Δφb(t) between the two counter-propagating waves 24, 25.
The modulation control voltage Vm(t) at the input of the phase modulator 33 being decomposed into the sum of the feedback modulation voltage Vcr(t) and the biasing modulation voltage Vb(t), the phase-shift modulation φm(t) (respectively the phase-difference modulation Δφm(t)) is the sum of the feedback phase-shift modulation φcr(t) (respectively the feedback phase-difference modulation Δφcr(t)) and the biasing phase-shift modulation φb(t) (respectively the biasing phase-difference modulation Δφb(t)), such that: φm(t)=φcr(t)+φb(t), and Δφm(t)=Δφcr(t)+Δφb(t).
According to the invention, the biasing phase-shift modulation φb(t) is the sum of:
For that purpose, the biasing means 130 are arranged so that the biasing digital signal is the sum of a first biasing component and a second biasing component.
The first biasing component is associated with a first voltage component Vb1(t), through the digital/analog converter 31 and the amplifier 32.
Likewise, the second biasing component is associated with a second voltage component Vb2(t), through the digital/analog converter 31 and the amplifier 32.
Therefore, the biasing modulation voltage Vb(t) is decomposed into the sum of a first voltage component Vb1(t) and a second voltage component Vb2(t), generated through the amplifier 32 and the digital/analog converter 31, respectively from the first biasing component and the second biasing component.
According to the invention, the first biasing phase-shift modulation component φb1(t) 35A, generated from the first voltage component Vb1(t) thanks to the phase modulator 33, introduces a first biasing phase-difference modulation component Δφb1(t) between the counter-propagating waves 24, 25 of the SAGNAC ring interferometer 20.
According to the invention, this first biasing phase-difference modulation component Δφb1(t) is a square-wave periodic modulation at a first biasing modulation frequency fb1 such that fb1=(2k1+1)fp, k1 being a natural number and fp the proper frequency.
According to a preferred embodiment of the invention, the first biasing modulation frequency fb1 is herein equal to the proper frequency fp(k1=0) of the SAGNAC ring interferometer 20.
As a variant, the first biasing modulation frequency fb1 may be, for example, an odd multiple of the proper frequency fp, such that k1>0.
An example of this first biasing phase-difference modulation component Δφb1(t) is shown and denoted by 34 in
According to the invention, the first biasing phase-difference modulation component Δφb1(t) 34 has extreme levels of values +π and −π. This first modulation component will hence be referred to hereinafter “π-modulation”.
In a preferred embodiment of the invention, this π-modulation, denoted by 34, has herein a duty factor of 50%, i.e. the duration of the level +π (respectively the level −π) represents 50% (respectively 50%) of the total duration of the period of π-modulation, denoted by 34.
The first biasing modulation frequency fb1 being herein equal to the proper frequency fp=1/(2Δτg) of the SAGNAC ring interferometer 20, the period of the π-modulation, denoted by 34, is equal to 1/fb1=2Δτg, the π-modulation, denoted by 34, remaining at its level +π during a half-period of duration Δτg, and at its level −π during another half-period of duration Δτg.
The first voltage component Vb1(t) produces a first biasing phase-shift modulation component φb1(t) 34A as shown in
Therefore, it is understood how the first biasing phase-difference modulation component Δφb1(t) 34, shown in
In a particular embodiment of the invention, where each stair step of the feedback phase-shift modulation φcr(t) has a duration Δτg, the feedback phase-shift modulation φcr(t) created through the modulation chain 30 is synchronous with the first biasing phase-shift modulation component φb1(t) which is herein at the proper frequency fp.
It will be defined herein that the feedback phase-shift modulation φcr(t) and the first biasing phase-shift modulation component φb1(t) are in phase with each other when the passage of the feedback phase-shift modulation φcr(t) from one step to another occurs during a transition of the first biasing phase-shift modulation component φb1(t) from one extreme level to another.
According to this particular embodiment, the falling down to 2π of the feedback phase-shift modulation φcr(t) is then synchronized with a transition of the π-modulation, denoted by 34.
According to the invention, the second biasing phase-shift modulation component φb2(t), generated from the second voltage component Vb2(t) thanks to the phase modulator 33, introduces a second biasing phase-difference modulation component φb2(t) between the counter-propagating waves 24, 25 of the SAGNAC ring interferometer 20.
According to the invention, this second biasing phase-difference modulation component φb2(t) is a periodic modulation at a second biasing modulation frequency fb2, such that fb2=(2k2+1)fp, k2 being a non-zero natural number such that k2>k1 and fp being the proper frequency.
The second biasing modulation frequency fb2 is hence a frequency strictly higher than the first biasing modulation frequency fb1.
According to a particular embodiment of the invention, the second biasing modulation frequency fb2 is such that fb2=3fp (i.e. k2=1). It is hence effectively strictly higher than the first biasing modulation frequency fb1, which is herein such that fb1=fp.
In another embodiment, the second biasing modulation frequency fb2 is preferentially such that k2>2, and still more preferentially such that k2>4.
According to a particular embodiment of the invention, the second biasing phase-difference modulation component Δφb2(t) is herein a square-wave modulation.
An example of this second biasing phase-difference modulation component Δφb2(t) is shown and denoted by 35 in
Generally, the second biasing phase-difference modulation component Δφb2(t) may have extreme levels of values +π/a and −π/a, a being a real number verifying the condition |a|>1. The second modulation component is then generally referred to as “π/8-modulation”.
As shown in
The second biasing modulation frequency fb2 being herein equal to 3fp=3/(2Δτg) of the SAGNAC ring interferometer 20, the period of π/8-modulation, denoted by 35, is equal to 1/fb2=(2/3)Δτg, the π/8-modulation, denoted by 35, remaining at its level +π/8 during a half-period of duration (1/3)Δτg, and at its level −π/8 during another half-period of duration (1/3)Δτg.
According to another embodiment of the invention, the second biasing phase-difference modulation component Δφb2(t) is a sine-wave periodic modulation, of amplitude π/a, such that a is a non-zero real number verifying the condition |a|>1.
The second biasing modulation voltage Vb2(t) produces a second biasing phase-shift modulation component φb2(t) 35A as shown in
Therefore, it is understood how is generated the second biasing phase-difference modulation component Δφb2(t) 35, shown in
According to the particular embodiment described hereinabove, the second biasing modulation frequency fb2 is an odd multiple of the first biasing modulation frequency fb1. Indeed, the first biasing modulation frequency fb1 being such that fb1=fp, the second biasing modulation frequency fb2 is such that fb2=3fp=3fb1=(2k21+1)fb1, with k21=1.
Moreover, the first biasing phase-difference modulation component Δφb1(t) 34 and the second biasing phase-difference modulation component Δφb2(t) 35 are herein in phase quadrature.
It will be defined herein that the first biasing phase-difference modulation component Δφb1(t) 34 and the second biasing phase-difference modulation component Δφb2(t) 35 are in phase quadrature when a transition of the first biasing phase-difference modulation component Δφb1(t) 34 from one extreme level to another one occurs at equal distance from two successive zeroes of the second biasing phase-difference modulation component Δφb2(t) 35.
As illustrated in
In
In each of the
For more simplicity, the reasoning is made for these
The luminous power P(Δφ) 37 received by the detector 14 is indeed of the form: P(Δφ)=P0[1+cos(Δφ)]. It is hence zero when Δφ=+π or −π, (because cos(+π)=cos(−π)=−1) and it is maximum and equal to 2P0 when Δφ=0 (because cos(0)=1).
The reasoning may be transposed to the case of the closed loop.
The first biasing phase-difference modulation component Δφb1(t) 34 (modulation π) being herein at the frequency fp and the second biasing phase-difference modulation component Δφb2(t) 35 (π/8-modulation) being at the frequency 3fp, the biasing phase-difference modulation Δφb(t) 36, which is the sum of the two previous modulations, is hence a periodic modulation at the proper frequency fp.
As described hereinabove, the modulation π, denoted by 34, has two extreme levels +π and −π and the π/8-modulation, denoted by 35, has two extreme levels +π/8 and −π/8, so that the biasing phase-difference modulation Δφb(t) has sequentially four different levels defining four different modulation states, which are:
These four distinct modulation states E1, E2, E3, E4 are preferably close to a dark fringe of the SAGNAC ring interferometer 20, where the signal-to-noise ratio is optimum.
The luminous power P(Δφ) 37 received by the detector 14 is hence modulated according to these four distinct modulation states and the modulated electrical signal S(t) 38 delivered by the detector 14 takes sequentially four values S1, S2, S3 and S4 respectively associated with the four modulation states E1, E2, E3 and E4.
When the parameter to be measured generates a zero phase difference Δφp, as it is the case in
Δφ=Δφb+Δφp=Δφb=Δφb1+Δφb2=+π−π/8=8π/8
Δφ=Δφb+Δφp=Δφb=Δφb1+Δφb2=+π+π/8=9π/8
Δφ=Δφb+Δφp=Δφb=Δφb1+Δφb2=−π+π/8=−7π/8
Δφ=Δφb+Δφp=Δφb=Δφb1+Δφb2=−π−π/8=−9π/8
The luminous power P(Δφ) 37 received by the detector 14 being a cosine-wave function, as explained above, it is herein, in the case of
From the preceding situation, described in
This may be shown in
Δφ=Δφb+Δφp=[+π−π/8]+π/16=15π/16
Δφ=Δφb+Δφp=[+π+π/8]+π/16=19π/16
Δφ=Δφb+Δφp=[−π+π/8]+π/16=−13π/16
Δφ=Δφb+Δφp=[−π−π/8]+π/16=−17π/16
Hence, as can be seen in
The detector 14 then delivers a modulated electrical signal S(t) 38 as shown in
It is also observed in
These peaks 39 are cumbersome insofar as they introduce non wanted defects in the modulated electrical signal S(t) 38.
This modulated electrical signal S(t) 38 is then digitized by the analog/digital converter 111 that delivers and transmits a digital electrical signal to the digital processing unit 112.
This digital electrical signal is also modulated and takes four digital values E1, E2, E3 and E4 according to the four modulation states E1, E2, E3 and E4 of the biasing phase-difference modulation Δφb(t) 36 with which the values are associated.
The digital processing unit 112 demodulates the digital electrical signal in phase with the second biasing phase-difference modulation component Δφb2(t) 35 independently of the first biasing phase-difference modulation component Δφb1(t) 34.
It is meant by this that the digital processing unit 112 delivers a first demodulated digital signal Σp based on the four digital values Σ1, Σ2, Σ3 and Σ4 respectively associated with the four modulation states E1, E2, E3 and E4, by performing a calculation operation of the type: Σp=−Σ1+Σ2+Σ3−Σ4 where the “weight” of each digital value in the preceding expression depends on the extreme level of the π/8-modulation, denoted by 35, in the modulation state associated with this digital value, but does not depend on the level of the modulation π, denoted by 34, in this modulation state.
The digital processing unit 112 hence produces a first demodulated digital signal Σp depending of the phase-shift Δφp and representative of the value of the parameter to be measured in the SAGNAC ring interferometer 20.
It is herein observed that the first demodulated digital signal Σp is a signal at the same frequency that the π/8-modulation, denoted by 35, i.e. 3fp. Nevertheless, it is also observed that the defect-bearing peaks 39 in the modulated electrical signal S(t) 38 transmitted by the detector 14 occur at the frequency equal to the double of the frequency of the modulation π, denoted by 34, i.e. herein at the frequency 2fp.
Hence, it is herein possible, by using a first biasing phase-difference modulation component Δφb1(t) 34 at the frequency fp and a second biasing phase-difference modulation component Δφb2(035 at the frequency 3fp, to obtain a signal representative of the parameter to be measured at a frequency 3fp and to limit the number of defect-bearing peaks 39 in the modulated electrical signal S(t) 38 delivered by the detector 14. The device 10 may then detect the rapid variations of the parameter to be measured, without thereby degrading the accuracy of the measurement.
Generally, according to the invention, the number of defect-bearing peaks 39 is function of the first biasing modulation frequency fb1 of the first biasing phase-difference modulation component Δφb1(t) 34 and the frequency of the signal function of the parameter to be measured provided by the digital processing unit 112 is equal to the second biasing modulation frequency fb2 of the second biasing phase-difference modulation component Δφb2(t) 35. Hence, the response time of the fiber-optic measurement device 10 according to the invention is substantially reduced and the accuracy of the measurement is maintained.
In a closed-loop operation, the first demodulated digital signal Σp serves as an error signal to control the total phase difference Δφt to zero by compensating the non-reciprocal phase-shift Δφp with the opposite phase-shift Δφcr introduced by the phase modulator 33 controlled by the feedback means 120.
This phase-shift Δφcr being generated through the same modulation chain 30 as the biasing phase-difference modulation Δφb, the control of the modulation chain 30, whose operation is described in detail hereinafter, thus allows to have a stable and controlled measurement of Δφcr, and hence finally of Δφp, which is opposite thereto and which is the parameter that is desired to be measured.
In practice, the transfer function, which depends on the characteristics of both the digital/analog converter 31 via its analog reference voltage and the amplifier 32 via its variable gain G, may undergo variations as a function of the measurement conditions, for example the temperature of operation of the device 10 of the electrical drift of certain electronic components included in the electronic means 100. Generally, the parameters influencing the transfer function generate low and slow variations of the latter, so that the gain control means 150 operate easily and rapidly so as to keep adjusted the transfer function of the modulation chain 30.
In the case shown in
On the other hand, the first biasing modulation frequency fb1 of the first biasing phase-difference modulation component Δφb1(t) 34 remains unchanged and herein equal to the proper frequency fp.
Likewise, as shown in
These changes of extreme levels of modulation has for consequence that the biasing phase-difference modulation Δφb(t) 36 is also modified by the multiplicative factor (1+ε)=16/15.
Hence, this homothetic transformation on the four modulation levels causes a change of the four modulation states E1, E2, E3 and E4 on which is modulated the signal received by the detector 14, which is function of the luminous power P(Δφ) 37 received depending on the open-loop phase difference Δφ the output of the SAGNAC ring interferometer 20.
The four levels of the biasing phase-difference modulation Δφb(t) 36 associated to the four modulation states are hence now:
Δφ=Δφb+Δφp=(1+ε)[π−π/8]=14π/15
Δφ=Δφb+Δφp=(1+ε)[π+π/8]=18π/15
Δφ=Δφb+Δφp=(1+ε)[−π+π/8]=−14π/15
Δφ=Δφb+Δφp=(1+ε)[−π−π/8]=−18π/15
Hence, the luminous power P(Δφ) 37 received by the detector 14 in the modulation states E1 and E4 is identical, but lower than the luminous power received when the transfer function of the modulation chain 30 is correctly adjusted, as in
Likewise, the luminous power P(Δφ) 37 received by the detector 14 in the modulation states E2 and E3 is identical, but higher than the luminous power received when the transfer function of the modulation chain 30 is correctly adjusted, as in
The detector 14 then delivers a modulated electrical signal S(t) 38 as shown in
The four values Σ1, Σ2, Σ3 and Σ4 of the digital electrical signal respectively associated with the four modulation states E1, E2, E3 and E4 being also identical two-by-two with Σ1=Σ3 and Σ2=Σ4, the first demodulated digital signal Σp, calculated according to the operation Σp=−Σ1+Σ2+Σ3−Σ4, is zero, which indicates that the value of the phase difference Δφp due to the parameter to be measured is also zero.
Moreover, the digital electrical signal delivered by the analog/digital converter 111 is transmitted to the gain control means 150 as shown in
The gain control means 150 demodulate the digital electrical signal so as to provide a signal function of the transfer function of the modulation chain 30.
More precisely, the other digital processing unit of the gain control means 150 performs a calculation operation of the type: ΣG=Σ1−Σ2+Σ3−Σ4, so as to produce a second demodulated digital signal ΣG independent of the phase difference Δφp generated by the parameter to be measured but significant of the transfer function of the modulation chain 30.
In particular, in the case shown in
The second demodulated digital signal ΣG then serves as an error signal for a control loop of the transfer function of the modulation chain 30.
For that purpose, the second demodulated digital signal ΣG is filtered by a control-loop digital integrating filter that then feeds the digital/analog converter 31 to control the analog reference voltage or the amplifier 32 to control the variable gain G thereof.
Hence, the transfer function of the modulation chain 30 is kept correctly adjusted between the value of the digital control signal and the value of the phase-shift modulation effectively applied by the phase modulator 33.
It will be observed that, in the case of
Indeed, in this case:
In the particular case of
Hence, the four levels of the biasing phase-difference modulation Δφb(t) 36 associated with the four modulation states are hence herein:
Δφ=Δφb+Δφp=(1+ε)[π−π/8]+π/24=117π/120
Δφ=Δφb+Δφp=(1+ε)[π+π/8]+π/24=149π/120
Δφ=Δφb+Δφp=(1+ε)[−π+π/8]+π/24=−107π/120
Δφ=Δφb+Δφp=(1+ε)[−π−π/8]+π/24=−139π/120
As can be observed in
Hence, the first demodulated digital signal ΣP representative of the phase difference Δφp and of the parameter to be measured in the SAGNAC ring interferometer 20 is non-zero.
Likewise, the second demodulated digital signal ΣG significant of the transfer function of the modulation chain 30 is non-zero, showing that the latter is effectively incorrectly adjusted.
Hence, the measurement of the phase difference Δφp due to the parameter to be measured and that of the transfer function of the modulation chain (30) are performed independently of each other.
The measurement device of the invention is particularly well adapted to the making of a gyrometer. In this case, the parameter to be measured is a component of the rotational speed of the ring interferometer 20.
This gyrometer hence advantageously enters in the making of navigation or inertial-stabilization systems.
Such an arrangement is also well adapted to the making of a magnetic-field and electrical-current measurement device, taking advantage of the FARADAY effect.
| Number | Date | Country | Kind |
|---|---|---|---|
| 1251541 | Feb 2012 | FR | national |
| Filing Document | Filing Date | Country | Kind |
|---|---|---|---|
| PCT/FR2013/050351 | 2/20/2013 | WO | 00 |
