Patent Grant
10317521
Field
The aspects of the disclosed embodiment relates to Doppler radar technology.
Background
Doppler radar has been the effective tool for detection of moving target range and velocity. However the maximum detectable range and velocity of current radar technology is limited by range-Doppler ambiguity. For atmosphere remote sensing research, Doppler velocity measurements have been especially challenging for radars onboard fast moving platforms (such as spacecrafts including satellites) or radars with rapid scanning antennas due to Doppler spectrum broadening caused by, for example, the ground speed of the spacecraft or motion of the rapid scanning antenna.
Millimeter wave radars have been widely used for atmospheric remote sensing and tracking hard-targets from airborne platforms. In large part, the popularity of these millimeter wave radars is because fine antenna beam widths can be achieved while still allowing for physically small and lightweight designs as well as higher backscattering efficiency for weather targets such as cloud particles. However, the product of the unambiguous range and Doppler velocity is limited by the radar wavelength (i.e. cλ/8). Airborne millimeter wave radars that are required to have long range coverage, therefore, may have Doppler Nyquist ranges that are much smaller than the relative speeds of objects they are designed to detect, such as an aircraft.
In Doppler radar operation, the maximum unambiguous range is determined by the radar transmission pulse repetition time, T, as r=CT/2, where C is the speed of light, r is the range of the target. The radar maximum unambiguous Doppler velocity, vn, is also related to T as vn=λ/4T, where λ is the radar signal wavelength. Therefore, r and vn are interrelated as r*vn=Cλ/8. For a fixed T, r and vn have to be chosen with compromise to keep within this restriction. Therefore, the Doppler-Range ambiguity dilemma exists for the traditional pulse mode Doppler radars. Different methods, such as “Staggered” Pulse Repetition Time (PRT) and Polarization Diversity Pulse Pair (PDPP), have been used for extending the Doppler measurement range.
The “staggered” PRT is a multi-rate sampling method that determines the Doppler velocity based on the ratio of Doppler velocities measured by 2 or more PRTs. A weakness of this approach is that the resulting Doppler velocity estimates have increased sensitivity to noise, relative to the usage of a single PRT. The Polarization Diverse Pulse-Pair (PDPP) method utilizes two pulses with orthogonal polarizations. These two pulses can be transmitted with a shorter time interval to extend the Doppler Nyquist range, while the PRT can be adjusted independently for range coverage. Polarization isolation between these two orthogonal polarized signals enables the avoidance of the radar range ambiguity.
However, these methods have their intrinsic limits, such as increased phase error (“Staggered PRT) and increased complexity in radar hardware for the transmission and reception of dual polarized signals (PDPP). In addition, finite polarization isolation of the antenna and the radar hardware may result in cross talk between the receiver channels for strong echo signals in PDPP operation.
The aspects of the disclosed embodiments are directed to a pin alignment fixture. The advantages provided by aspects of the disclosed embodiments are achieved by the subject matter of the independent claims. Further advantageous modifications can be found in the dependent claims.
In one embodiment, a method for operating a radar system. The method includes transmitting at least two pairs of pulses, each pair including center frequencies f1 and f2 and such that the order the pulses f1 and f2 are transmitted is reversed every alternate pair transmission; receiving a corresponding reflection signal as a reception signal after the at least two pairs of pulses are reflected by a point scatterer; and determining the mean radial velocity vr of the point scatterer from a transmitted waveform and received signal of frequency f1 followed by a transmitted waveform and a received signal of frequency f2 in a first pulse pair and a transmitted waveform and received signal of frequency f2 followed by a transmitted waveform and a received signal of frequency f1 in a second pulse pair where the first and second pulse pairs are transmitted in succession.
In another embodiment, a method for operating a radar system. The method includes transmitting at least two pairs of pulses, each pair including center frequencies f1 and f2 and such that the order the pulses f1 and f2 are transmitted is reversed every alternate pair transmission; receiving a corresponding reflection signal as a reception signal after the at least two pairs of pulses are reflected by a point scatterer; and determining the mean radial velocity vr where vr=cΔΦ/(4π(f1+f2), c is the speed of light, ΔΦ=ΔΦorder1+ΔΦorder2, and ΔΦorder1 and ΔΦorder2 determined using pulse pair algorithm and measurements obtained by f1/f2 and f2/f1 pulse pairs as ΔΦorder1=arg(ERx,f1(t)E*Rx,f2(t+ΔT)), and ΔΦorder2=arg(ERx,f2(t)E*Rx,f1(t+ΔT)).
In another embodiment, a radar system. The radar system includes a transceiver, a radar antenna and a digital receiver/processor. The transceiver is configured to generate at least two pairs of pulses, each pair including center frequencies f1 and f2 and such that the order the pulses f1 and f2 are transmitted is reversed every alternate pair transmission. The radar antenna is connected to the transceiver to (1) receive from the transceiver and transmit the at least two pairs of pulses and (2) receive and transmit to the transceiver a corresponding reflection signal after the at least two pairs of pulses are reflected by a point scatterer. The digital receiver/processor is connected to the transceiver to receive the reflection signals and configured to determine the mean radial velocity vr of the point scatterer from a transmitted waveform and received signal of frequency f1 followed by a transmitted waveform and a received signal of frequency f2 in a first pulse pair and a transmitted waveform and received signal of frequency f2 followed by a transmitted waveform and a received signal of frequency f1 in a second pulse pair where the first and second pulse pairs are transmitted in succession.
These and other aspects, implementation forms, and advantages of the exemplary embodiments will become apparent from the embodiments described herein considered in conjunction with the accompanying drawings. It is to be understood, however, that the description and drawings are designed solely for purposes of illustration and not as a definition of the limits of the disclosed invention, for which reference should be made to the appended claims. Additional aspects and advantages of the invention will be set forth in the description that follows, and in part will be obvious from the description, or may be learned by practice of the invention. Moreover, the aspects and advantages of the invention may be realized and obtained by means of the instrumentalities and combinations particularly pointed out in the appended claims.
In the following, the invention will be explained in more detail with reference to the example embodiments shown in the drawings, in which:
The patent or application file contains at least one drawing executed in color. Copies of this patent or patent application publication with color drawing(s) will be provided by the Office upon request and payment of the necessary fee.
The present disclosure relates to utilizing a frequency diverse pulse-pair (FDPP) determination method and embodiments incorporating such technology, for example, using frequency diversity pulse-pairs for Doppler phase estimation to extend the Doppler Nyquist range or rate of millimeter radars without causing range ambiguity. This method and technology have potential application in many areas such as spaceborne, airborne and ground-based weather radar, air traffic control, commercial collision avoidance system and defense related high speed moving target detection.
To increase a radar unambiguous Doppler velocity range, the time interval between two adjacent pulses, ΔT, has to be small. However, a small ΔT may result in a short unambiguous range. The FDPP determination method and technology of the embodiments described herein utilize digital waveform generation and digital receiver technologies whereby, for example, a pair of pulses is transmitted with slightly shifted center frequencies in each pulse repetition period. More than two pulses, each with different shifted center frequencies, may also be utilized in the embodiments described herein. Radar return signals from these pulses can be separated by digital filters implemented in a digital receiver. As a result, since ΔT can be as small as needed, Doppler Nyquist then can be increased to a desired range without causing range ambiguity. However, since the frequency separation between the pulse pair is preferably at least several MHz so that the digital filter can separate them, the radar return signals from these pulses are likely decorrelated, which makes Doppler phase estimate unpractical.
The present disclosure includes embodiments involving systems and methods to minimize this effect. In one embodiment, the Frequency Diversity Pulse Pair (FDPP) method described herein is implemented by, for example, alternating the order of the pulse pair transmitted or order of the group of multiple pulses transmitted, the pulses differentiated based on the center frequency of each transmitted pulse. For example, where a pair of transmitted pulses have center frequencies f1 and f2, the pulses are transmitted in pairs such that the first pair may be f1 followed by f2 and the second pair may be a different order, such as f2 followed by f1. To elaborate further on the embodiment, two pulses at center frequencies of f1 and f2 are transmitted separated by a lag time of ΔT. While retaining ΔT, the order of the pulses is reversed every alternate transmission. From the receive channels at f1 and f2, the pulse-pair phase estimate of the two sequences are individually accumulated and stored as Δϕorder1 and Δϕorder2. Finally Doppler velocity is estimated from the sum of an equal number of the two individual pulse-pair phase estimates (denoted as Δϕ).
The use of two closely spaced radar frequencies can introduces two sources of error, however, two mechanisms of error cancellation are utilized in the embodiments disclosed herein. First, a “beat” phase that scales as a function of range can be introduced. However, this term vanishes when the phases of the f1/f2 pair and f2/f1 pairs are added together, thus, canceling out the expected value sense. Second, since there is little correlation between the f1 and f2 pulses, the variances of the f1/f2 phase estimates may be large. However, since the f1/f2 and f2/f1 phase estimates are highly anticorrelated, the sum of the two phase estimates has a much smaller variance than the individual phase estimates. As a result of the FDPP determination and integration of the phase estimates of f1/f2 pulse pair and f2/f1 pulse pair in equal numbers, the phase shift as a function of range between two pulses is canceled to enable the retrieval of Doppler phase.
In at least one aspect of the disclosed embodiments, the methods disclosed herein may be executed using a radar system 200, for example, shown in
In at least one aspect of the disclosed embodiments, the systems and methods disclosed herein may be executed by one or more computers or processor-based components under the control of one or more programs stored on computer readable medium, such as a non-transitory computer readable medium.
The computing apparatus 300 may include computer readable program code or machine readable executable instructions (such as, for example, the Frequency Diversity Pulse-Pair (FDPP) determination, integration of the phase estimates of f1/f2 pulse pair and f2/f1 pulse pair and other data analysis utilized in the embodiments disclosed herein) stored on at least one computer readable medium 302, which when executed, are configured to carry out and execute the processes and methods described herein, including all or part of the embodiments of the present disclosure. The computer readable medium 302 may be a memory of the computing apparatus 300. In alternate aspects, the computer readable program code may be stored in a memory external to, or remote from, the apparatus 300. The memory may include magnetic media, semiconductor media, optical media, or any media which may be readable and executable by a computer. Computing apparatus 300 may also include a processor 304 for executing the computer readable program code (such as, for example, the Frequency Diversity Pulse-Pair (FDPP) determination, integration of the phase estimates of f1/f2 pulse pair and f2/f1 pulse pair and other data analysis utilized in the embodiments disclosed herein) stored on the at least one computer readable medium 302. In at least one aspect, computing apparatus 300 may include one or more input or output devices to allow communication among the components of the exemplary radar system, including, for example, what may be generally referred to as a user interface 306, such as, the operator workstation described above, which may operate the other components included in the Doppler radar system or to provide input or output from the computing apparatus 300 to or from other components of the Doppler radar system. User interface 306 may include display unit 208 included in the embodiment of
In another embodiment of the FDPP determination, the transmitted waveform at frequency f1 is denoted as ETx,f1(t)=E0,f1 cos [2π f1t+ΨTx,f1], where E0,f1 is the amplitude of the transmitted signal, the phase of the transmitted signal is ΨTx,f1 and t denotes time. The received signal ERx,f1 at time t, where c is the speed of light, fD1 is the Doppler shift of f1, Af1 is the backscatter ratio of f1 and R the range to a point scatterer, can be written as ERx,f1(t)=Af1E0,f1 cos [2π f1(t+2R/c)+2π fD1(t+R/c)+ΨTx,f1].
Similarly, Tx and Rx signals at frequency f2 and transmitted at time t+ΔT can be written as follows, where the range to the point-scatter is R+vrΔT where vr is the radial velocity of the point-scatter, fD2 is the Doppler shift of f2, and Af2 is the backscatter ratio of f2, ETx,f2(t)=E0,f2 cos [2π f2t+ΨTx,f2] and ERx,f2(t+ΔT)=Af2E0,f2 cos [2π f2(t+2(R+vrΔT)/c)+2π fD2(t+(R+vrΔT)/c)+ΨTx,f2].
Assuming Af1=Af2 (where A is the backscatter ratio), E0,f1=E0,f2, f1>>fD1 (where fD1 is the Doppler shift for f1) and f2>>fD2 (where fD2 is the Doppler shift for f2), the echo phase change ϕRx−ϕTx for frequencies f1 and f2 are denoted as Φf1 and Φf2, respectively are
Φf1=2πf1(t+2R/c)+2πfD1(t+R/c)+ΨTx,f1−2πf1t−ΨTx,f1
Φf1=2πf1(2R/c)+2πfD1(t+R/c)
Φf2=2πf2 [t+2(R+vrΔT)/c]+2πfD2 [t+(R+vrΔT)/c]+ΨTx,f2−2πf2 t−ΨTx,f2
Φf2=2πf2(2(R+vrΔT)/c)+2πfD2 [t+(R+vrΔT)/c]
The FDPP determination is based on 2 quantities ΔΦorder1 and ΔΦorder2. Here ΔΦorder1=Φf2−Φf1 and ΔΦorder2=Φf1−Φf2.
ΔΦorder1=4π(f2−f1)R/C+2π(2f2+fD2)vrΔT/c
Similarly,
ΔΦorder2=4π(f1−f2)R/C+2π(2f1+fD2)vrΔT/c
ΔΦ=ΔΦorder1+ΔΦorder2
Since fD1≈fD2, using fD to replace fD1 and fD2, ΔΦ=4π(f1+f2+fD)vrΔT/c
Since f1>>fD and f2>>fD, ΔΦ=4π(f1+f2)vrΔT/c
vr=cΔΦ/(4π(f1+f2)ΔT)
Since all values other than vr are solely system dependent, the radial component of target (point scatterer) mean radial velocity vr of the point scatterer can be obtained once the ensemble-averaged ΔΦ is determined. In one embodiment, a “Pulse Pair (PP)” method can be used to calculate the phase change between the return signals of an f1/f2 pulse pair and an f2/f1 pulse pair. The calculated phase change can then be used to determine ΔΦ. PP is a method for weather radar Doppler phase estimate and D. S. Zrnic, Spectral Moment Estimates from Correlated Pulse Pairs, IEEE Transactions on Aerospace and Electronic Systems, Vol. AES-13, No. 4, 344-354, July 1977 related thereto is hereby incorporated by reference.
In Pulse Pair processing, phase change ΔΦorder1 of pulse pair f1/f2 and ΔΦorder2 of pulse pair f2/f1 are calculated using autocorrelation function of the radar return signals as ΔΦorder1=arg(Rf1,f2(ΔT)) and ΔΦorder2=arg(Rf2,f1(ΔT)), where Rf1,f2(ΔT)=ERx,f1(t)E*Rx,f2(t+ΔT) and Rf2,f1(ΔT)=ERx,f2(t)E*Rx,f1(t+ΔT) are the autocorrelation functions of pulse pair f1/f2 and f2/f1, respectively (note: ERx,f1(t), ERx,f2(t+ΔT), ERx,f2(t), and ERx,f1(t+ΔT) are radar received signals defined in previous sections.).
ΔΦ=ΔΦorder1+ΔΦorder2=arg(ERx,f1(t)E*Rx,f2(t+ΔT))+arg(ERx,f2(t)E*Rx,f1(t+ΔT))
In an embodiment disclosed herein, frequencies f1 and f2 may be in the range of, for example, from about 75 GHz to about 110 GHz preferably about 95 GHz (W-band); from about 26.5 GHz to about 40 GHz preferably about 35.5 GHz (Ka-band); from about 12 GHz to about 18 GHz preferably about 13.6 GHz (Ku-band); and from about 8 GHz to about 12 GHz preferably about 9.6 GHz (X-band). Radars that may be utilized to implement the embodiments disclosed herein may be built to measure the backscattering signal from cloud particles and rain drops. They may include narrow band sensors (for example, instantaneous bandwidth<10 MHz in order to maximize the signal to noise ratio) and the operational frequencies (for example, 95 GHz/35.5 GHz/13.6 GHz/9.6 GHz) at each band may chosen at the atmospheric absorption window (to minimize the atmospheric attenuation).
In the embodiment disclosed herein, the frequency separation between the center frequencies of f1 and f2 (the value of Δf=f1−f2) may be from about 2 MHz to about 10 MHz preferably about 6 MHz. If Δf is too small, then the digital filter implemented in the digital receiver may limit the separation of the returns from the f1 pulse and the f2 pulse. If Δf is too large, the return signals at f1 and f2 may be decorrelated and difficult to estimate the Doppler phase using the FDPP method. In the embodiment disclosed herein, ΔT can be preferably between about 10 microseconds and about 100 microseconds preferably about 30 microseconds, airborne radar may be closer to the lower end of the range and land based radar may be closer to the higher end of the range. In the embodiment disclosed herein, pulse repetition time may be in the range of from about 3 KHz to about 6 KHz preferably about 5 KHz, airborne radar may be closer to the higher end of the range and land based radar may be closer to the lower end of the range.
In order to determine the confidence in the above calculations, σ denoting variance and ρ denoting the correlation operators were determined.
σ(ΔΦ)=σ(ΔΦOrder1)+σ(ΔΦOrder2)+2Cov(ΔΦorder1,ΔΦorder2)
Next, the covariance term is decomposed as
Cov(ΔΦOrder1,ΔΦOrder2)=ρ(ΔΦOrder1,σ(ΔΦOrder2)·√{square root over (ΔΦOrder1·σ(ΔΦOrder2)})
Where σ(ΔΦOrder1)=σ(ΔΦOrder2)
Cov(ΔΦOrder1,ΔΦOrder2)=ρ(ΔΦOrder1,σ(ΔΦOrder2)·σ(ΔΦOrder1)
From the above relationships
σ(ΔΦ)=2σ(ΔΦOrder1)+2ρ(ΔΦOrder1,σ(ΔΦOrder2)·σ(ΔΦOrder1)
After rearranging terms in the above equation
σ(ΔΦ)=2σ(ΔΦOrder1)[1+ρ(ΔΦOrder1,σ(ΔΦOrder2)]
Therefore, the underlying premise of the FDPP determination is that as ρ(ΔΦOrder1,σ(ΔΦOrder2)→−1, the variance of the phase composite estimate σ(ΔΦ)→0.
Monte-Carlo simulations were qualitatively compared with data-analysis results (all at W-band).
System benefits and improvements of the embodiments of the present disclosure include the following: (1) enabling Doppler radar to detect target velocity and range with extended ambiguity range; (2) lower-cost and lighter weight Doppler velocity retrievals on air-borne millimeter wave radars; and (3) being a cost effective approach to mitigate the range-Doppler ambiguity limit without the need of significant investment in hardware. It has broad potential application in spaceborne, airborne and ground-based Doppler weather radar, spacecraft landing control, air traffic control, high speed moving target detection and collision avoidance, as well as weather radar, airport traffic control radar, aviation and auto collision avoidance system, and high speed moving target detection such as aircraft or missile. As a result of the above, the embodiments disclosed herein to extend the Doppler Nyquist range or rate of millimeter radars without causing range ambiguity and permit Doppler measurements from a rapid moving platform, such as spacecraft, or radars using fast scanning antennas that had been especially challenging due to spectrum broadening and rapid decorrelation between successive radar transmission pulses.
While there have been shown, described and pointed out, fundamental novel features of the invention as applied to the exemplary embodiments thereof, it will be understood that various omissions and substitutions and changes in the form and details of devices and methods illustrated, and in their operation, may be made by those skilled in the art without departing from the spirit or scope of the invention. Moreover, it is expressly intended that all combinations of those elements and/or method steps, which perform substantially the same function in substantially the same way to achieve the same results, are within the scope of the invention. Moreover, it should be recognized that structures and/or elements and/or method steps shown and/or described in connection with any disclosed form or embodiment of the invention may be incorporated in any other disclosed or described or suggested form or embodiment as a general matter of design choice. It is the intention, therefore, to be limited only as indicated by the scope of the claims appended hereto.
The invention described herein was made by an employee of the United States Government, and may be manufactured and used by or for the Government for governmental purposes without the payment of any royalties thereon or therefor.
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| Number | Date | Country | |
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| 20170363715 A1 | Dec 2017 | US |
