Patent Application
20200382128
An analog-to-digital (A/D) converter (ADC) may be used to generate digital codes which represent the level of an analog signal. An analog-to-digital converter may be configured to operate in an iterative, multi-stage fashion, using a successive-approximation register (SAR) and a digital-to-analog (D/A) converter (DAC). An analog-to-digital converter which uses a successive-approximation register to develop a sequence of digital codes is referred to herein as a successive-approximation register analog-to-digital converter.
A digital-to-analog converter for a successive-approximation register analog-to-digital converter may have one or more capacitor arrays, as illustrated, for example, in U.S. Pat. No. 6,894,627 and United States Patent Application Publication No. 2016/0336952. The one or more capacitor arrays may be operationally controlled by approximation codes that are developed by the successive-approximation register. The entire disclosures of U.S. Pat. No. 6,894,627 and United States Patent Application Publication No. 2016/0336952 are hereby fully incorporated herein by reference.
This disclosure relates to a system which has a digital-to-analog converter; a reference signal coupled to the digital-to-analog converter; a differential amplifier for applying gain, and for generating output signals as a function of sampled input signals, the reference signal, digital codes, and the gain applied by the differential amplifier coupled to the digital-to-analog converter; and a multi-bit successive-approximation register for determining the digital codes in successive stages coupled to the differential amplifier; and wherein the gain applied by the differential amplifier is corrected based on previously determined digital codes.
This disclosure also relates to a method of digitizing an analog signal. The method includes: (A) sampling a first input voltage; (B) using a capacitor array and an amplifier to generate first output voltages for successive stages as a function of (1) the first input voltage, (2) a reference voltage, and (3) first digital approximation codes, and using a multi-bit successive-approximation register analog-to-digital converter to determine the codes; (C) subsequently, sampling a second input voltage, and using the capacitor array and the amplifier to generate second output voltages for successive stages as a function of (1) the second input voltage, (2) a reference voltage, and (3) second digital approximation codes, and using the analog-to-digital converter to determine the second digital approximation codes; (D) correcting gains of the amplifier based on previously determined digital approximation codes; and (E) using the first and second digital approximation codes to generate digital output signals corresponding to the first and second input voltages, respectively.
Referring now to the drawings, where like reference numerals designate like elements and steps, there is shown in
The analog-to-digital converter system 10 has a first reference source 16, a digital-to-analog converter 18, a sample-and-hold amplifier (SHA) 20, a variable-gain differential amplifier 22 for generating output voltages, a multi-bit flash-type analog-to-digital converter 24, a second reference source 26, a first encoder 28, a logic and timing circuit 30, and a gain-error estimation circuit 32. The logic and timing circuit 30 has a digital combiner circuit 33. In the illustrated example, the gain-error estimation circuit 32 has a state machine 40 and logic circuitry 42. This disclosure is not limited, however, to the specific implementation shown in the drawings and described herein. In the illustrated example, the analog-to-digital converter 24, the first encoder 28, and the logic and timing circuit 30 are elements of a successive-approximation register 44.
All of the components of the analog-to-digital converter system 10 may be connected to or integrated into an integrated circuit (IC) and/or a chip 50 fabricated according to various semiconductor and/or other processes. One or more of the conductive lines 12, 14 and other devices and elements 18, 32, 44 of the analog-to-digital converter system 10 may be diffused or implanted into one or more layers of semiconductor material (not illustrated). The integrated devices and elements 18, 32, 44 include transistors, resistors, and other suitable electronic devices that are not shown in the drawings for the sake of clarity.
The digital to-analog converter 18 has a capacitor array 54 which is operated, in each of the N successive stages, in response to respective digital, thermometric approximation codes Ci, where i=0, 1 N−1. In each stage, according to timing determined by the logic and timing circuit 30, the capacitor array 54 generates a comparison voltage VDAC on a reference comparison line 56. As illustrated in
This disclosure may be implemented in a variety of configurations and is not limited to the details illustrated in the drawings or mentioned herein. For example, this disclosure is not necessarily limited to any particular capacitor array. Arrays having different numbers of capacitors of different characteristics, with or without capacitor arrays split by a coupling capacitor, and/or configured in different ways relative to the reference voltage Vref, may be employed with other elements to achieve advantages over the prior art in line with this disclosure. Moreover, by way of example, this disclosure may be implemented with a current-steering digital-to-analog converter, without any capacitor array.
Moreover, although the illustrated capacitor array 54 is operated under the control of the thermometric approximation codes Ci, where each code Ci includes M digital values (D1, D2 . . . DM) for M capacitors, an alternative system in accordance with this disclosure may have binary capacitors operated under the control of binary approximation codes. In general, for this disclosure, the manner in which the approximation codes are determined by a successive-approximation register system may be implemented in various configurations.
In operation, before a first stage begins, and according to timing controlled by the logic and timing circuit 30, the sample and hold amplifier 20 samples the voltage Vin on the input line 14 (Step 100,
The comparison voltage VDAC is applied to the second input of the differential amplifier 22 via the reference comparison line 56, and the differential amplifier 22 outputs a differential voltage Vdiff on a differential line 64. As illustrated in
The predetermined gains, which are different for each stage, are stored in the logic and timing circuit 30. The correction gain Ei is determined by the gain-error estimation circuit 32 for each stage as described in more detail below. For each stage, the predetermined gain for that stage is applied to the amplifier 22 by the logic and timing circuit 30 by a suitable device (not shown), and the correction gain Ei is applied to the amplifier 22 by a suitable input device 66 which receives signals from the gain-error estimation circuit 32.
For each stage of the illustrated system 10, the differential voltage Vdiff is applied to the flash-type analog-to-digital converter 24 via the differential line 64. In the illustrated example, the analog-to-digital converter 24 has cascading high-speed comparators (not illustrated), and, via the first encoder 28, generates the thermometric approximation codes Ci that are applied to the capacitors of the array 54. The differential voltage Vdiff is an example of a “differential voltage signal” in accordance with this disclosure.
A reference REFi that is applied to the converter 24 may vary from stage to stage under the control and timing of the logic and timing circuit 30. In general, the reference REFi to the flash-type analog-to-digital converter 24 is scaled for successive stages, under the control of the logic and timing circuit 30. In each stage, the first encoder 28 receives the output of the converter 24 (that is, a flash code via a line 25) and uses the output to generate the respective approximation code Ci (Step 110). The converter 24 is an example of a “flash device” in accordance with this disclosure.
For any i-th cycle, the actual value X of the differential voltage Vdiff is unknown, such that the effective residue Ri in the system 10 (Ri=X−Σk=0iCk) is also unknown. In general, however, the effective residue Ri is the portion of the differential voltage Vdiff that is not resolved by the successive-approximation register 44 till the i-th stage. The unresolved portion (the effective residue Ri) is resolved (within the limits of the ultimate resolution of the system 10) during the stages that follow the i-th stage. Thus, for any i-th stage, the effective residue Ri (=X−Σk=0iCk) can be assumed to be equal to the sum of the approximation codes Ck for the stages that follow the i-th stage (that is, where k=i+1, i+2 . . . N−1), as follows:
The expected performance of an ideal successive-approximation register analog-to-digital converter system is represented by a dashed line in
Moreover, the expected performance for the ideal converter is statistically symmetrical, such that the average effective residue Ri equals zero. When Ci=0, the effective residue values Ri are between −Y and Y, and the average of those values is zero. Likewise, when Ci=−1, the effective residue values Ri are between −Y and Y, and the average of those values is zero. The slope of a straight line that is statistically fitted to the symmetrical accumulated performance curve of the ideal converter (the dashed line in
The accumulated performance of a less-than-ideal successive-approximation register analog-to-digital converter system is represented by a solid line in
Further, the accumulated performance for the less-than-ideal converter is not statistically symmetrical; the average effective residue Ri does not equal zero. When Ci=0, the effective residue values Ri are between MIN[0] (<−Y) and MAX[0] (>Y), but the average of those values is still zero. As the absolute value of Ci increases, the midpoint between the corresponding maximum and minimum effective residues Ri diverges more and more from the respective ideal value. For example, when Ci=−1, the corresponding effective residue values Ri are between MIN[−1] (<−Y) and MAX[−1] (>Y), and the average of those values is less than zero (for a positive gain error). Therefore, the difference between the less-than-ideal converter system and the ideal system (that is, the gain-error value) can be quantified most clearly in connection with approximation codes Ci having the greatest absolute values.
For each stage of the less-than-ideal converter system, the slope of a straight line that is statistically fitted to the asymmetrical accumulated performance curve is a non-zero value which is proportional to an estimated gain error Êi, and can be statistically determined as follows: First, statistical values MAX, MIN, and MID are collected and defined as follows:
The slope of the line which passes through MID[k] is proportional to and thereby can be used to determine an estimated value of the gain error Êi, as follows:
Ê
i
∝{MID[p]−MID[−p]}MID[k]=(MAX[k]+MIN[k+1])/2
where Êi is the statistically-determined gain error for the i-th cycle, and p denotes the maximum approximation code Ci for the i-th cycle. As indicated above, as the absolute value of Ci increases, the midpoint between the corresponding maximum and minimum effective residues Ri diverges more and more from the ideal value. Therefore, the difference between the less-than-ideal converter system and the ideal system (that is, the gain-error value) can be quantified most clearly in connection with approximation codes Ci that have the greatest absolute values.
Then, the accuracy of the estimated gain error Êi is refined by a suitable gradient descent algorithm, such as the following:
Ê
i[n+1]=Êi[n]+∝*{MID[p]−MID[−p]}
where Êi[n] is the gain error estimated in a previous iteration, Êi[n+1] is the gain error estimated in a next iteration, α is a suitable learning rate, and {MID[p]−MID[−p]} is the slope of the line which passes through MID[k] as mentioned above.
Referring now again to
Then, continuing in the first stage (Step 108), a new approximation code C1 is determined by the successive-approximation register 44, based on the differential voltage Vdiff (Step 110). The new approximation code C1 is communicated to the logic and timing and gain-error estimation circuits 30, 32 and the capacitor array 54, and is stored within the state machine 40 and logic circuitry 42 of the gain-error estimation circuit 32 (Step 112).
Subsequently, the process proceeds through the second through N-th stages (repeating “No” from Step 114, followed by Steps 104, 106, 108, 110, 112 for each stage) until N approximation codes Ci have been determined by the successive-approximation register 44. At that point (“Yes” from Step 114), an output code Da, Db . . . is determined by a second encoder 70 (
Each approximation code Ci determined each time a sampled voltage Vin is digitized is stored by the gain-error estimation circuit 32. The approximation codes Ci provide statistical data that are used to estimate gain errors Ei within the system 10, as explained above. For each stage of each analog-to-digital conversion performed by the system 10, the statistically determined gain error Ei is applied (66) to the amplifier 22 such that the gain error Ei is effectively subtracted from the predetermined gain for the respective stage, so that the operation of the system 10 approximates that of an ideal analog-to-digital converter system. In other words, the digital approximation codes Ci are used to generate gain-error correction values Ei for controlling the amplifier 22.
The gain Gi of the differential amplifier 22 is advantageously adjusted before each multi-level comparison (that is, for each stage), but the gain correction process is performed in the background as represented by the broken line in
The system 10 is well adapted for low-power applications, and because the system 10 converts multiple bits in each cycle, it can operate at high speed. The system described herein may be used, if desired, to perform low power 18/16 bit, 65 mega-samples per second (MSPS), analog-to-digital conversions, but may be scaled to lower sampling rates. The system 10 may also be used, if desired, to support 14 bit, 125 mega-samples per second, analog-to-digital conversions. To determine the gain errors Ei, approximation codes Ci for a series of analog-to-digital conversions are accumulated in the gain-error estimation circuit 32 for a suitable period of time, such as, for example, one micro-second. This disclosure should not be limited, however, to the particular features of the examples described herein.
According to this disclosure, an ideal amplifier gain is optimally predetermined for each stage of the conversion cycle, and any deviation in the actual gain from the designed or ideal value may lead to less than optimal performance. Therefore, it is desirable to correct for any such gain error. Although the gain error potentially could be addressed by using an over range and allocating error correction cycles, those approaches may not be fully satisfactory. Including one or more extra stages may result in higher power consumption and slower operation, and therefore may be less than satisfactory. Factory trimming of gain values for the amplifier may be unsatisfactory because it does not provide for tracking of gain variations over time, and hence disadvantageous over-design may be required if factory trimming is the only approach employed.
This disclosure advantageously utilizes a process that operates in the background (within the gain-error estimation circuit 32) to correct for gain errors. Although the gain-error estimation circuit 32 may rely on timing signals from the logic and timing circuit 30, the variable input data for the process (if desired, essentially the only data input into and processed by the gain-error estimation circuit 32) are the sequentially generated approximation codes Ci. Due to the background nature of the process, voltage and temperature variations and other gain errors can be corrected for in real time, if desired, while the system 10 is operating.
As noted above, this disclosure is not limited to the details of the illustrated system 10 shown in the drawings. In an alternative system constructed in accordance with this disclosure, for example, the logic and timing and gain-error estimation circuits 30, 32 may have one or more suitable processors, memories, and/or clock generators. Also, for example, other gradient descent algorithms besides the one described specifically above, and other processing steps and elements, may be used to calculate gain-error values Ei for correcting the gain of the amplifier 22 in real time.
What have been described above are examples. This disclosure is intended to embrace alterations, modifications, and variations to the subject matter described herein that fall within the scope of this application, including the appended claims. As used herein, the term “includes” means including but not limited to. The term “based on” means based at least in part on. Additionally, where the disclosure or claims recite “a,” “an,” “a first,” or “another” element, or the equivalent thereof, it should be interpreted to include one or more than one such element, neither requiring nor excluding two or more such elements.
| Number | Date | Country | Kind |
|---|---|---|---|
| 201841041188 | Oct 2018 | IN | national |
This continuation application claims priority to U.S. patent application Ser. No. 16/221,323, filed Dec. 14, 2018, which application claims priority to India provisional Patent Application No. 201841041188, filed Oct. 31, 2018, both of which are incorporated herein by reference in their entirety.
| Number | Date | Country | |
|---|---|---|---|
| Parent | 16221323 | Dec 2018 | US |
| Child | 16997975 | US |
