Half-bridge resonant converter

Abstract

A half-bridge resonant converter includes: a primary winding; a secondary winding having a first and a second end and a central point; a first electronic switch; a second electronic switch; a first power-storage element; a second power-storage element; and a load having a first and a second end. Wherein, the first end of said the secondary winding serially connects with said the first electronic switch and the first power-storage element, and the second end of said the secondary winding serially connects with said the second electronic switch and the second power-storage element, and the first end of said the load connects simultaneously with said the first power-storage element and the second power-storage element, and the second end connects with the central point of the secondary winding.

Description

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is the circuit diagram of the half-bridge resonant converter of the present invention;

FIG. 2 is the circuit diagram of the conventional half-bridge resonant converter;

FIG. 3 is the current waveform of the conventional half-bridge resonant converter;

FIG. 4 is the current conduction circuit of the conventional half-bridge resonant converter in mode 1;

FIG. 5 is the current conduction circuit of the conventional half-bridge resonant converter in mode 2;

FIG. 6 is the current conduction circuit of the conventional half-bridge resonant converter in mode 3;

FIG. 7 is the conduction waveform of the element of the half-bridge resonant converter of the present invention and the conventional half-bridge resonant converter;

FIG. 8 is the loss current waveform as use of the Schottky Diode as the electronic switch diodes D₊ and D₋;

FIG. 9 is the loss current waveform as use of the MOSFET as the electronic switch diodes D₊ and D₋;

FIG. 10 is the loss current waveform of the half-bridge resonant converter of the present invention.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

While the invention has been described by way of examples and in terms of preferred embodiments, it is to be understood that those who are familiar with the subject art can carry out various modifications and similar arrangements and procedures described in the present invention and also achieve the effectiveness of the present invention. Hence, it is to be understood that the description of the present invention should be accorded with the broadest interpretation to those who are familiar with the subject art, and the invention is not limited thereto.

FIG. 1 shows the circuit diagram of the half-bridge resonant converter of the present invention, wherein the circuit includes: a primary winding; a secondary winding N2 having a first and a second end and a central point; a first electronic switch having a first and a second end, wherein the first end connects with the first end of the secondary winding; a second electronic switch having a first and a second end, wherein the first end connects with the second end of the secondary winding; a first power-storage element having a first and a second end, wherein the first end connects with the second end of the first electronic switch; a second power-storage element having a first and a second end, wherein the first end connects with the second end of the second electronic switch; and a load having a first and a second end, wherein the first end connects simultaneously with the second end of the first power-storage element and the second end of the second power-storage element, and the second end connects with the central point of the secondary winding.

The first electronic switch of the secondary winding is a combination of a MOSFET Q₊ and a winding N₃, and the second electronic switch is a combination of a MOSFET Q₋ and a winding N₃, so that the secondary winding can reach the objective of synchronizing rectification. The action of the first electronic switch is in the reverse of the second electronic switch, and causes the first end or the second end of the secondary winding to conduct the load alternatively to reach the objective of half-wave rectification. Moreover, the winding N₃ of the first electronic switch and the winding N₃ of the second electronic switch are the same windings of equal coil number.

The filter inductances L₊, L₋, are connected separately to the first electronic switch and the second electronic switch for rectifying the output current from the first electronic switch and the second electronic switch. The filter inductance L₊ is combined in parallel with a series of a diode D₊ and a resistance R₊ to form a first power-storage element, and he filter inductance L₋ is combined in parallel with a series of a diode D₋ and a resistance R₋ to form a second power-storage element for providing an power release route for the filter inductance L₊, L₋ to overcome the voltage drop between the output voltage V₀ of the load and the first end voltage V′ of the secondary winding.

The present invention is to improve the power loss and the conversion efficiency of a conventional half-bridge converter. FIG. 2 shows the circuit structure diagram of a conventional half-bridge converter. The converter includes a primary side circuit and a secondary side circuit, wherein V_d is the input voltage, and V₀ is the output measuring voltage, and coil number ratio between the primary winding N₁, and the secondary winding N₂ is N=N₁/N₂.

FIG. 3 shows the functioning waveform of the half-bridge converter. Because the first half-cycle and the second half-cycle of the converter are symmetrical working modes, therefore the half-bridge converter can be divided into the follows modes by the first half-cycle:

Mode 1: (t₀˜t₁)

In the state of mode 1, the transistor Q_H and Q_L are all not conducted. The initial current of the resonant inductance L_r and the magnetized inductance L_m is initialized as I₀, and the initial voltage of the resonant capacitance C_r is initialized as V₀. Because I₀ is smaller than 0, the current conductance waveform of I₀ is shown in FIG. 4. And because the bridged voltage of the magnetized inductance L_m is fixed as nV_o, therefore the magnetized inductance L_m can be deemed as a constant DC voltage source, as well as the resonant current I_Lr and the resonant capacitance C_r current are equal, thus obtains:

$\left\{\begin{array}{c}{L}_r\frac{i_{L_r}\left(t\right)}{t}+v_{C_r}\left(t\right)=\frac{V_d}{2}-{\mathrm{nV}}_o\\ i_{\mathrm{Lr}}\left(t_0\right)=I_0\\ V_c\left(t_0\right)=V_0\end{array}\right\}\hspace{1em}$

Further the current and the voltage equations of the resonant inductance L_r and the resonant capacitance C_r can be obtained as follows:

$\{\begin{array}{c}{i}_{L_r}\left(t\right)=\left\{I_0\cos \left[{\omega }_{r1}\left(t-t_0\right)\right]+\frac{\frac{V_d}{2}-{\mathrm{nV}}_o-V_0}{Z_{01}}\sin \left[{\omega }_{r1}\left(t-t_0\right)\right]\right\}u\left(t-t_0\right)\\ \begin{array}{c}{v}_{C_r}\left(t\right)=\{\frac{V_d}{2}-{\mathrm{nV}}_o-\left(\frac{V_d}{2}-{\mathrm{nV}}_o-V_0\right)\cos \left[{\omega }_{r1}\left(t-t_0\right)\right]+\\ I_0Z_{01}\sin \left[{\omega }_{r1}\left(t-t_0\right)\right]\}u\left(t-t_0\right)\end{array}\end{array}\hspace{1em}$

According to the above equation, the current and the voltage waveform of the resonant inductance L_r and the resonant capacitance C_r from t₀ to t₁ can be obtained as shown in FIG. 3.

Wherein, as the magnetized inductance L_m can be deemed as a constant DC voltage source, therefore the resonant inductance L_r and the resonant capacitance C_r of the primary side may be deemed as resonant, and the resonant frequency is as follows:

${\omega }_{r1}=\frac{1}{\sqrt{\mathrm{LrCr}}}$

and a characteristic resistance Z₀₁ can be obtained as follows:

$Z_{01}=\sqrt{\frac{\mathrm{Lr}}{\mathrm{Cr}}}$

while the current equation of the magnetized inductance L_m can be derived as follows:

$\begin{array}{c}\{\begin{array}{c}{L}_m\frac{i_{L_m}\left(t\right)}{t}={\mathrm{nV}}_o\\ i_{L_m}\left(t_0\right)=I_0\end{array}\Rightarrow i_{L_m}\left(t\right)=i_{L_m}\left(t_0\right)+{\int }_0^t\frac{{\mathrm{nV}}_o}{L_m}\tau \\ =I_0+\frac{{\mathrm{nV}}_o}{L_m}\left(t-t_0\right)\end{array}$

and the current slope of the magnetized inductance L_m can be expressed as:

$\mathrm{Slope}\left(I_{\mathrm{Lm}}\right)=\frac{n·\mathrm{Vo}}{\mathrm{Lm}}$

According to the above equation, the current waveform of the magnetized inductance L_m from t₀ to t₁ can be obtained as shown in FIG. 3.

When the resonant current I_Lr is larger than 0, the current direction of the resonant current I_Lr is reversed, therefore the diode D_H will be terminated, and then the working mode of the half-bridge converter will get into mode 2.

Mode 2: (t₁˜t₂)

In the state of mode 2, as the resonant current I_Lr is in reverse direction, therefore the transistor Q_H is conducted and the current conductance waveform is shown in FIG. 5. The same with mode 1, the magnetized inductance L_m can be deemed as a constant DC voltage source, as well as the resonant current I_Lr and the resonant capacitance C_r current are equal, thus obtains:

$\left\{\begin{array}{c}{L}_r\frac{i_{L_r}\left(t\right)}{t}+v_{C_r}\left(t\right)=\frac{V_d}{2}-{\mathrm{nV}}_o\\ i_{\mathrm{Lr}}\left(t_1\right)=0\\ V_c\left(t_1\right)=V_1\end{array}\right\}\hspace{1em}$

Further the current and the voltage equations of the resonant inductance L_r and the resonant capacitance C_r can be obtained as follows:

$\{\begin{array}{c}i_{L_r}\left(t\right)=\frac{\frac{V_d}{2}-{\mathrm{nV}}_o-V_0}{Z_{01}}\sin \left[{\omega }_{r1}\left(t-t_1\right)\right]u\left(t-t_1\right)\\ v_{C_r}\left(t\right)=\left\{\frac{V_d}{2}-{\mathrm{nV}}_o-\left(\frac{V_d}{2}-{\mathrm{nV}}_o-V_1\right)\cos \left[{\omega }_{r1}\left(t-t_1\right)\right]\right\}u\left(t-t_1\right)\end{array}\hspace{1em}$

According to the above equation, the current and the voltage waveform of the resonant inductance L_r and the resonant capacitance C_r from t₁ to t₂ can be obtained as shown in FIG. 3.

As well as the resonant frequency and characteristic resistance are the same as which in the state of mode 1:

${\omega }_{r1}=\frac{1}{\sqrt{\mathrm{LrCr}}},Z_{01}=\sqrt{\frac{\mathrm{Lr}}{\mathrm{Cr}}}$

and the equation of the magnetized inductance L_m can be derived as follows:

$\begin{array}{c}\{\begin{array}{c}{L}_m\frac{i_{L_m}\left(t\right)}{t}={\mathrm{nV}}_o\\ i_{L_m}\left(t_0\right)=I_1\end{array}\Rightarrow i_{L_m}\left(t\right)=i_{L_m}\left(t_1\right)+{\int }_0^t\frac{{\mathrm{nV}}_o}{L_m}\tau \\ =I_1+\frac{{\mathrm{nV}}_o}{L_m}\left(t-t_1\right)\end{array}$

and the slop of the magnetized inductance L_m can be expressed as:

$\mathrm{Slope}\left(I_{\mathrm{Lm}}\right)=\frac{n·\mathrm{Vo}}{\mathrm{Lm}}$

According to the above equation, the current waveform of the magnetized inductance L_m from t₁ to t₂ can be obtained as shown in FIG. 3.

The current I₂ of the secondary side can not be reversed from the transistor polarity of the secondary side circuit, therefore the relative current I₁ of primary side shall not be smaller than 0. Hence the half-bridge converter will be get into mode 3 if the resonant current I_Lr and the magnetized current I_Lm are the same.

Mode 3: (t₂˜t₃)

In the state of mode 3, as the resonant current I_Lr and the magnetized current I_Lm are equal, the primary side current I₁ will be 0 and the current waveform is shown in FIG. 6. The resonant inductance L_r and the magnetized inductance L_m are in series and resonant with the resonant capacitance C_r and a current relation form can be obtained as follows:

$\left\{\begin{array}{c}\left(L_r+L_m\right)\frac{i_{L_r}\left(t\right)}{t}+v_{C_r}\left(t\right)=\frac{V_d}{2}\\ i_{\mathrm{Lr}}\left(t_2\right)=I_2\\ V_c\left(t_2\right)=V_2\end{array}\right\}\hspace{1em}$

Further the current and the voltage equations of the resonant inductance L_r and the resonant capacitance C_r can be obtained as follows:

$\{\begin{array}{c}{i}_{L_r}\left(t\right)=\left\{I_2\cos \left[{\omega }_{r2}\left(t-t_2\right)\right]+\frac{\frac{V_d}{2}-V_2}{Z_{02}}\sin \left[{\omega }_{r2}\left(t-t_2\right)\right]\right\}u\left(t-t_2\right)\\ v_{C_r}\left(t\right)=\left\{\frac{V_d}{2}-\left(\frac{V_d}{2}-V_2\right)\cos \left[{\omega }_{r2}\left(t-t_2\right)\right]+I_2Z_{02}\sin \left[{\omega }_{r2}\left(t-t_2\right)\right]\right\}u\left(t-t_2\right)\end{array}\hspace{1em}$

According to the above equation, the current and the voltage waveforms of the resonant inductance L_r resonant capacitance C_r from t₀ to t₁ can be obtained as shown in FIG. 3.

Wherein, as the resonant inductance L_r and the magnetized inductance L_m are in series and resonant with the resonant capacitance C_r, therefore a resonant frequency can be obtained as follows:

${\omega }_{r2}=\frac{1}{\sqrt{\left(\mathrm{Lr}+\mathrm{Lm}\right)\mathrm{Cr}}}$

and a characteristic resistance Z₀₂ can be obtained as follows:

$Z_{02}=\sqrt{\frac{\mathrm{Lr}+\mathrm{Lm}}{\mathrm{Cr}}}$

Hence the current equation of the magnetized inductance L_m can be derived as:

i _Lm(t)=i_Lr(t)

Further, according to the above equation, the current waveform of the magnetized inductance L_m from t₂ to t₃ can be obtained as shown in FIG. 3, and the primary side current I₁ will be 0. As the secondary side current I₂ and the primary side current I₁ are in direct ratio, a relation can be obtained as follows:

$I_2=\frac{N_1}{N_2}I_1=0$

As such, in the state of mode 3, the current of the resonant inductance L_r and the magnetized inductance L_m are all equal, and the slop is smaller than which in the state of mode 1 and mode 2:

$\mathrm{Slope}\left(I_{\mathrm{Lm}}\right)=\frac{n·\mathrm{Vo}}{L_c+\mathrm{Lm}}$

When the transistor Q_H of primary side is turned off, mode 3 will be terminated.

FIG. 7 shows the conducting status of the half-bridge resonant converter of the present invention and the conventional half-bridge resonant converter element. The diode D₊ and D₋ used in the circuit of the secondary winding of the conventional half-bridge resonant converter as the electronic switch will resulted in a considerable power loss. FIG. 8 shows the loss current waveform as use of the Schottky Diode as the electronic switch diodes D₊ and D₋, take the current of 16 A as an example, wherein if the Schottky Diode is a conventional Schottky Diode and the forward voltage drop is about 0.5V, therefore the power loss is about:

P _d =V _F ×I _O=0.5×16=8 W

and wherein if the Schottky Diode is a low voltage drop type Schottky Diode and the forward voltage drop is about 0.3V, therefore the power loss is about:

P _d V _F ×I _O=0.3×16=4.8 W

If the diodes D₊ and the D₋ are replaced by the MOSFET Q₊ and Q₋, the power loss will be reduced substantially. FIG. 9 shows the loss current waveform when using MOSFET to substitute the electronic switch diode D₊ and D₋, wherein the forward voltage drop when the MOSFET conducted is about 0.07V, and the voltage drop when the Body Diode conducted is about 0.6V, while the conduction time of the MOSFET is about twice than the conduction time of the Body Diode, therefore the power loss is about:

$P_d=V_F\times I_O=0.07\times 16\times \frac{2}{3}+0.6\times 16\times \frac{1}{3}=3.9W$

Using the MOSFET to substitute the diodes as electronic switched will incur the problem of a reverse bias. In the state of mode 3, the secondary side current I₂ is 0, and V′ and the V₀ have a voltage drop, thus may possibly cause the MOSFET generating a reverse bias, and then the reverse bias will cause the Body Diode to be conducted and incur the loss of electrical power raising substantially. Therefore, the present invention is to connect an power-storage element in series behind the electronic switch, and to use the filter inductance L₊ and L₋ to eliminate the voltage difference existing between V′ and the V₀:

$\frac{i_2\left(t\right)}{t}=\frac{V_o-V^{\prime }}{L_{+}}$

The resistance connected with the diode in series, D₊ serially connected with R₊ and D₋ serially connected with R₋, in the power-storage element forming a power release route of the filter inductance L₊ and the L₋. When mode 3 is terminated, the filter inductance L₊ and L₋ will release the power stored in the inductance via the power release route of D₊ serially connected with R₊ and D₋ serially connected with R₋. Thereby the half-bridge resonant converter of the present invention can overcome the problem that the reverse bias incurs the Body Diode conducted. FIG. 10 shows the loss current waveform of the half-bridge resonant converter of the present invention, wherein the forward voltage drop during the MOSFET conducted is about 0.07V, therefore its power loss is about:

P _d =V _F ×I _O=0.07×16=1.12 W

According to the abovementioned comparison of the values of power loss, the half-bridge resonant converter of the present invention can reach the objective of lowest power loss.

While the invention has been described by way of examples and in terms of preferred embodiments, it is to be understood that the invention is not limited thereto. To the contrary, those who are familiar with the subject art can carry out various modifications and similar arrangements and procedures, under the scope of appended claims and broadest interpretation.

Claims

1. A half-bridge resonant converter, comprising: a primary winding;a secondary winding having a first and a second end, and a central point;a first electronic switch having a first and a second end, wherein said first end connects with the first end of said secondary winding;a second electronic switch having a first and a second end, wherein said first end connects with the second end of said secondary winding;a first power-storage element having a first and a second end, wherein said first end connects with the second end of said first electronic switch;a second power-storage element having a first and a second end, wherein said first end connects with the second end of said second electronic switch; anda load having a first and a second end, wherein said first end connects simultaneously with the second end of said first power-storage element and the second end of the second power-storage element, and said second end connects with the central point of said secondary winding.

2. The half-bridge resonant converter according to claim 1, wherein the action of said first electronic switch is in the reverse of said second electronic switch, and causes the first end or the second end of said secondary winding to conduct said load alternatively.

3. The half-bridge resonant converter according to claim 1, wherein said first electronic switch comprises a MOSFET and a third winding bridged with the Gate (G) and the Source (S) of said MOSFET, further the first end of said first electronic switch is the Source (S) of said MOSFET, and the second end is the Drain (D) of said MOSFET.

4. The half-bridge resonant converter according to claim 1, wherein said second electronic switch comprises a MOSFET and a third winding bridged with the Gate (G) and the Source (S) of said MOSFET, further the first end of the second electronic switch is the Source (S) of said MOSFET, and the second end is the Drain (D) of said MOSFET.

5. The half-bridge resonant converter according to claim 3, wherein the third winding of said first electronic switch and the third winding of said second electronic switch are the same windings with equal coil number.

6. The half-bridge resonant converter according to claim 1, wherein said first power-storage element is a combination of an inductance in parallel with a series connection of a diode and a resistance, so that the power of said inductance can be released through the series connection of said diode and said resistance, moreover the first end of said first power-storage element is the positive end of said diode, and the second end of said first power-storage element is the end of the combination of said resistance and said inductance.

7. The half-bridge resonant converter according to claim 1, wherein the second power-storage element is a combination of an inductance in parallel with a series of a diode and a resistance, so that the power of said inductance can be released through the series connection of said diode and said resistance, moreover the first end of said second power-storage element is the positive and of said diode, and the second end of said second power-storage element is the end of the combination of said resistance and said inductance.