Patent Application
20100080319
A modulator is arranged in a device for signal transmission, such as used in systems for wireless or wireline communication. One of the functions of a modulator is to modulate a useful signal representing an information to be transmitted onto a carrier frequency signal so to provide a transmission signal. The transmission signal is amplified before being provided to a transmission channel. In case of a base-station of a mobile communication system, the amplifier has to provide for a high amplification gain.
Generally, in mobile communication systems, diverse modulation schemes allowing for a high bandwidth are provided, such as EDGE (Enhanced Data Rates for GSM Evolution), UMTS (Universal Mobile Telecommunication System), etc. These modulation schemes usually provide a non-constant envelope of the transmission signal. This may be due to an amplitude modulation of the transmission signal. The amplitude modulation carries part of the information transmitted. To allow for a correct demodulation and reassembly of the information, certain linearity requirements have to be fulfilled by the modulator. E.g., the linearity requirements are set by a standard specification. In consequence, the power amplifier of a base-station has to provide a linear amplification over a wide range of signal level. This is usually achieved by providing for a “quasi-linear” amplifier running with a high back-off to ascertain the linearity requirements with respect to the transmission signal. This mode of operation leads to a low efficiency in power and usually to higher costs in production of an according modulator.
For these and other reasons, there is a need for the present invention.
The detailed description is described with reference to the accompanying figures. In the figures, the left-most digit(s) of a reference number identifies the figure in which the reference number first appears. The same numbers are used throughout the drawings to reference like features and components.
This disclosure is directed to techniques for modulation in a transmission device. More particularly, the techniques involve implementation of a transmission device and of a modulator. The disclosed apparatus' can be implemented in a variety of communication devices or systems. For example, a transmission device or a modulator can be implemented in mobile phones, base stations, etc. The following systems and methods are described with reference to a mobile communication system; however, it will be appreciated that the disclosed transmission devices and modulators can be implemented generally in any electronic communication system.
Mobile communication systems include components such as mobile communication devices, base stations, etc., that can receive input signals . . . .
The disclosed technique for implementing a modulation involves a combination of a switched output stage and an error estimation path.
The transmission device includes an input 102. A baseband signal comprised of information to be transmitted may be received at the input 102. The baseband signal is passed to a Modulation Path 104. The Modulation Path 104 performs a modulation of the baseband signal to produce an output signal. A modulation may include any of the following signal processing: coding; up-converting the baseband signal on a carrier signal; amplifying; filtering; etc. The different kind of signal processing may be performed at various sequence or times.
The Modulation Path 104 includes one or several switched output stages not shown in
To achieve an according bandwidth, the Modulation Path 100 may include a plurality of switched output stages in parallel. In another embodiment, it may include a single switched output stage. In one embodiment, it includes at least one serial circuit of switched output stages to achieve a required power level of the output signal.
In one embodiment, the Modulation Path 104 further includes a coding unit that performs a coding the time depending phase component and the time depending envelope component of the input signal received at input 102. The coding unit may thus be arranged to provide a high bandwidth modulation scheme, such as 8-PSK, used in EDGE, UMTS, etc. One elementary function of the coding unit is to provide a rectangular pulse train derived from the baseband signal provided at the input 102. Exemplary implementations of the coding unit make use of a Delta-Sigma Modulation, Pulse-Width Modulation or any other method to transform an envelope signal into pulses. One aspect of using a coding unit is that the spectrum of the rectangular pulse train concentrates at multiples of the reference frequency used in the coding unit. Thereby, gaps in the spectrum may be used to improve the SNR in the high frequency domain, e.g. after up-converting the rectangular pulse train on a carrier frequency. Another aspect is that the frequency of switching at the output stage is reduced in comparison to a known band-pass modulator. Thus, a high efficiency is achieved while spectral components concentrate around the carrier frequency, which implies a smaller bandwidth of the output stage needed.
The transmission device 100 further includes an Error Computation Unit 106. The function of the Error Computation Unit 106 is to estimate an according Inband Error of the output signal produced by the Modulation Path 104. This is achieved by producing an error signal based on information on error contributions, such as unwanted image components, quantization effects, non-linearity present in the output signal provided by the Modulation Path 104. Information on error contributions may be retrieved by a feedback or feed forward kind of correction, i.e. by using at least one of:
The Error computation unit 106 includes a linear output stage not shown in
The transmission device 100 further includes an Error Correction Unit 108. The Error Correction Unit 108 receives the output signal provided by the Modulation Path 104 and the error signal provided by the Error Computation Unit 106. Its function is to correct the output signal by the error signal to produce a transmission signal. The Error Correction Unit 108 may include any of the following apparatus': an adder; a power combiner; a filter, such as a band-pass filter, etc. The transmission signal is provided at an output 110 comprised by the transmission device 100. The output 110 may couple to a feed to feeds the transmission signal into a transmission channel, such as an antenna or a plug.
The Coding Unit 202 performs a coding the time depending phase component and the time depending envelope component of the input signal 102. The Coding Unit 202 may thus be arranged to provide a high bandwidth modulation scheme, such as 8-PSK, used in EDGE, UMTS, etc. One elementary function of the Coding Unit 202 is to provide a pulse train derived from the input signal 102. Exemplary implementations of the Coding Unit 202 make use of a Delta-Sigma Modulation or of a Pulse-Width Modulation or any other method to transform an envelope signal into pulses. In general, the Coding Unit 202 produces a first intermediate signal 204, being a rectangular pulse train.
An Up-Converter 206 receives the first intermediate signal 204. It further receives a carrier signal 208, i.e. a high frequency signal. The Up-Converter 206 performs an up-conversion of the first intermediate signal 204 by the carrier signal 208 to produce a second intermediate signal 210. This corresponds to shifting the center frequency of the first intermediate signal 204, i.e. baseband, to the center frequency of the carrier signal 208, i.e. the carrier frequency.
In many communication systems, the carrier frequency is more than 1 GHz. Yet, in other communication systems, the carrier frequency may be lower than 1 GHz, e.g. in GSM it is 800-900 MHz.
The Up-Converter 206 may include a mixing device, such as a multiplier. A local oscillator, not shown in
A Switched Output Stage 212 receives the second intermediate signal 210. It performs an amplification of the second intermediate signal 210 to provide an output signal 214. The Switched Output Stage 212 may include a switched amplifier, i.e. an amplifier of any of Class-F; Inverse Class F; Class D, Class S, etc. To achieve an according bandwidth/outputpower, Switched Output Stage 212 may include a plurality of switched amplifiers in parallel. In another embodiment, it may include a single switched amplifier. In one embodiment, it includes at least one serial circuit of switched amplifiers to achieve a required power level of the output signal 214.
The system 200 further includes an Error Estimator 216. The Error Estimator 216 estimates an Inband Error of the output signal 214. This is achieved by producing an error signal based on information on error contributions, such as unwanted image components, quantization effects, non-linearity present in the output signal 214. Information on error contributions may be retrieved by a feedback or feed forward kind of correction, i.e. by using at least one of:
A second Up-Converter 220 receives the first intermediate error signal 218. It further receives a second carrier signal 222, i.e. a high frequency signal. The second carrier signal 222 may be identical to the carrier signal 208. The second Up-Converter 220 may be of a structure similar to Up-Converter 206. It performs an up-conversion of the first intermediate error signal 218 by the carrier signal 222 to produce a second intermediate error signal 224. This corresponds to shifting the center frequency of the first intermediate error signal 218, e.g baseband, to the center frequency of the carrier signal 222, i.e. the carrier frequency.
A Linear Output Stage 226 receives the second intermediate error signal 224. It performs an amplification of the second intermediate error signal 224 to provide an error signal 228. It includes a linear amplifier, such as a Class A, Class B, Class AB, etc. The inventors have recognized that the dynamic range of the error signal 228 is much smaller than the dynamic range of the output signal 214. It is therefore sufficient to perform amplification of the error signal by a simple linear amplification structure.
The system 200 further includes a Combiner 230. The Combiner 230 receives the output signal 214 and the error signal 228. It performs a combination of the output signal 214 and the error signal to provide a transmission signal 232. This may be achieved by subtracting or adding the error signal 228 to the output signal 214. The Combiner 230 may also perform a band-pass filtering of the result of combination. The transmission signal 232 may be transferred to a transmission channel by means of an according feed, such as an antenna or a plug.
The Coding Unit 202, the Up-Converter 206, and the Switched Output stage 212 form a Modulation Path that provides the output signal 214, derived from input signal 102 by a modulation of the later. The modulation of the input signal 102 is performed by making use of a switched amplification. There is a large variety of coding an enevelope section and/or a phase section of a baseband signal to achieve an according input to the switched amplifier. This includes techniques, such as an Envelope Modulation, which is described in further detail with respect to
A Pulse Width Modulator (PWM) 306 receives the first input signal 302. It performs a coding of the first input signal 302 to a rectangular pulse stream. The PMW 306 operates at a reference frequency chosen with regard to the bandwidth of the envelope component and to the required SNR. It produces a first intermediate signal 204 in form of a pulse stream. The width of a pulse is in linear relation to a value of an amplitude at a sampling time determined by the reference frequency.
A Phase Modulator 308 receives the second input signal 304. The Phase Modulator 308 provides a high-frequency and phase modulated LO signal 310 having a center frequency at a carrier frequency of the system 300. It may include any structure of a frequency synthesizer, such as a Digitally Controlled Oscillator (DCO), a Voltage Controlled Oscillator (VCO); a Phase-Locked Loop (PLL), a Digital PLL (DPLL), a Ring Oscillator; etc. The LO signal 310 thus carries the phase information of the input signal, shifted to the carrier frequency. In many embodiments, the LO signal 310 is a rectangular (bit-like) pulse stream.
A First Mixer 312 receives the first intermediate signal 204 and the LO signal 310. It mixes or performs of a multiplication of the two to provide a second intermediate signal 210. In real implementation, a knot may replace the First Mixer 312. A Switched Output Stage 212, as described with respect to
A Filter 314 receives the first intermediate signal 204. It performs a filtering of the first intermediate signal 204 to estimate an Inband Error. The Filter 314 may include a highpass type of filter. It provides a first intermediate error signal 218.
A Second Mixer 316 receives the first intermediate error signal 218 and the LO signal 310. It mixes or performs of a multiplication of the two to provide a second intermediate error signal 224. In real implementation, a knot may replace the Second Mixer 316. A Linear Output Stage 226, as described with respect to
An Adder/Power-Combiner 318 receives the output signal 214 and the error signal 228. It subtracts the error signal 228 from the output signal 214. The resulting signal is passed to a Band-Pass Filter (BPF) 320. The BPF 320 performs a band-pass filtering to provide a transmission signal 112. The transmission signal 232 may be transferred to a transmission channel by means of an according feed, such as an antenna or a plug.
A First PWM 406 receives the first input signal 402. It performs a coding of the first input signal 402 to a rectangular pulse stream. The First PMW 406 operates at a reference frequency chosen with regard to the signal bandwidth and to the required SNR. It produces a first intermediate signal 408 in form of a rectangular pulse stream. Since the Inphase component takes positive, as well as negative values, a 3 level pulse-width modulation may be applied. A pulse thus may take one of the values (+A, 0, −A), A being the maximal amplitude of the rectangular pulse stream. Transition from +A to −A may always pass through 0 to avoid phase shifts by 180° (i.e. by Pi). The width of a pulse is in linear relation to a value of amplitude of the first input signal 402 at a sampling time determined by the reference frequency.
A Second PWM 410 receives the second input signal 404. It performs a coding of the second input signal 404 to a rectangular pulse stream. The Second PMW 408 operates at same reference frequency as the First PWM 406. It produces a second intermediate signal 412 in form of a rectangular pulse stream. Since the Quadrature component takes positive, as well as negative values, a 3 level pulse-width modulation may be applied. A pulse thus may take one of the values (+A, 0, −A), A being the maximal amplitude of the rectangular pulse stream. Transition from +A to −A may always pass through 0 to avoid phase shifts by 180° (i.e. by Pi). The width of a pulse is in linear relation to a value of amplitude of the second input signal 404 at a sampling time determined by the reference frequency.
A First Mixer 414 receives the first intermediate signal 408 and a LO signal 416. It mixes or performs of a multiplication of the two to provide a third intermediate signal 418. In real implementation, a knot may replace the First Mixer 414. The LO signal 416 may be provided by a frequency synthesizer or Local Oscillator, such as, such as a Digitally Controlled Oscillator (DCO), a Voltage Controlled Oscillator (VCO); a Phase-Locked Loop (PLL), a Digital PLL (DPLL), a Ring Oscillator; etc. A First Switched output stage 420, as described with respect to
A Second Mixer 424 receives the first intermediate signal 412 and a phase-shifted LO signal 426. It mixes or performs of a multiplication of the two to provide a fourth intermediate signal 428. In real implementation, a knot may replace the Second Mixer 424. The phase-shifted LO signal 426 corresponds to the LO signal 416 by a phase-shift of 90° (i.e. Pi/2). The phase shifted LO signal 426 may be derived from the LO signal 416, e.g by a phase shifting device, a delay line, etc. A frequency synthesizer, the same or a different to the frequency synthesizer that generates LO signal 416 may generate the phase-shifted LO signal 426. A Second Switched Output Stage 430, as described with respect to
The signal paths including the First Mixer 414, the First Switched Output Stage 420, the second Mixer 424 and the Second Switched Output Stage 430 form an IQ-Modulation Path.
A First Filter 434 receives the first intermediate signal 408. It performs a filtering of the first intermediate signal 408 to estimate an Inphase component of an Inband Error. The First Filter 434 may include a highpass type of filter. It provides a first intermediate error signal 436. A Second Filter 438 receives the second intermediate signal 412. It performs a filtering of the second intermediate signal 412 to estimate an Quadrature component of an Inband Error. The Second Filter 438 may include a highpass type of filter. It provides a second intermediate error signal 440.
A Third Mixer 442 receives the first intermediate error signal 436 and the LO signal 416. It mixes or performs of a multiplication of the two to provide a third intermediate error signal 444. A Fourth Mixer 446 receives the second intermediate error signal 440 and the phase-shifted LO signal 426. It mixes or performs of a multiplication of the two to provide a fourth intermediate error signal 448. An Adder 450 receives the third intermediate error signal 444 and the fourth intermediate error signal 448 to provide a combined error signal 452. A Linear Output Stage 226, as described with respect to
The system 400 further includes a combiner 454 that receives the Inphase output signal 422 and the Quadrature output signal 432 to perform a phase-correct combination of both. It further receives the error signal to subtract the Inband Error contribution. It thus provides a transmission signal 456 that may be passed via a BPF to a transmission channel, e.g. by means of a plug or an antenna.
In general, any high-frequency signal may be written in the subsequent representation, if being phase- and amplitude-modulated:
x(t)=a(t) cos(ωct+φ(t)). (1)
The amplitude a(t) of the signal depends on time t, like the phase φ(t) is time-depending. Time-dependency of both components is used to code information on the signal. The carrier frequency fc is considered by ωc=2*Pi*fc. Because of the time-dependancy of the amplitude a(t), this signal may not be used for controlling a switched amplifier. In the described technique, the envelope contribution a(t) is coded, e.g. using a pulse-width modulation.
The resulting coded signal is a rectangular pulse signal having a constant frequency, which depends on the reference frequency of the applied coding, such as delta-sigma coding, pulse-width modulation, etc.. In pulse-width modulation, the pulse length is in linear dependency of the amplitude a(t). The reference frequency is chosen in dependency of bandwidth of the envelope contribution, i.e. the amplitude. It will also be chosen with respect to a required SNR: In general, it is smaller than a frequency used in known techniques, such as a Band-pass Modulation. The time domain of an envelope signal modulated by a pulse-width modulation is shown in
Usually, a pulse-width modulated signal has a larger bandwidth than the carrier frequency signal. In consequence, the spectral parts of a baseband signal overlap, if up-converted by a carrier frequency fc.
In some embodiments, a modulation frequency fPWM of some pulse-width modulated signals in a systems using the described technique is adjusted according to the following relation:
f
PWM*(n−0.5)=2*fc, (2)
n being an integer number. This results in a maximal SNR, in relation of the complex baseband signal a(t)*ej*φ(t) to noise caused by image frequency.
The described technique makes use of further reducing the SNR requirements. Mirror contributions of the PWM modulated signal at ±2*fc are extracted, e.g. by high pass or bandpass filtering. The mirror contributions a mixed (e.g. by multiplication) with a phase modulated carrier signal to the desired carrier frequency fc. The spectrum of such a signal is shown in
As shown in
It has to be noted that mirror components are deterministic. Mirror contributions depend on the envelope, the modulation frequency fPWM used for PWM modulation, the modulation method, etc. The contributions may therefore e calculated be means of a digital signal processor instead of using a filter as shown with respect to the above-described examples.
The method introduced may, but need not, be implemented at least partially in architecture(s) such as shown in
At block 1402, a useful signal is received as input. The useful signal may comprise information to be transmitted, such as audio, video, mail, etc. At block 1404, the useful signal is coded to provide a first intermediate signal. Coding may be based on a methods such as Sigma-Delta Modulation or Pulse Width Modulation. At block 1406, the first intermediate signal is up-converted to provide a second intermediate signal. Up-converting refers to any shifting of center frequency of a spectrum. It is performed using a RF carrier signal, such as a carrier signal of a wireline or wireless communication system. The RF carrier signal may comprise a single carrier frequency or a frequency band. At block 1408, the second intermediate signal is amplified. Amplification may involve a switched amplification, thereby achieving a high efficiency. An output signal is provided.
At block 1410, a first intermediate error signal is provided. The first intermediate error signal may be derived from the first intermediate signal, e.g. involving a step of filtering. Since error contributions are deterministic, the first intermediate error signal may be provided by other means, such as a digital signal processing. At block 1412, the first intermediate error signal is up-converted to provide a second intermediate error signal. As above, up-converting refers to any shifting of center frequency of a spectrum. It is performed by using the RF carrier signal. At block 1414, the second intermediate error signal is amplified. Amplification may involve a linear amplification, thereby making use of a simple implementation. An error signal is provided.
At block 1416, a output signal and the error signal are combined to provide a modulated signal. This step may include a bandpass filtering of the modulated signal.
Although embodiments for power amplifier with output power control have been described in language specific to structural features and/or methods, it is to be understood that the appended claims are not necessarily limited to the specific features or methods described. Rather, the specific features and methods are disclosed as exemplary implementations for power amplifier with output power control.
