1. Field of the Invention
The present invention relates to analog to digital converters (ADC's), and more particularly, to reducing nonlinearities and inter-symbol interference in high-speed analog to digital converters.
2. Related Art
A subranging analog to digital converter (ADC) architecture is suitable for implementing high-performance ADC's (i.e. high speed, low power, low area, high resolution).
Modern flash, folding and subranging analog to digital converters (ADC's) often use averaging techniques for reducing offset and noise of amplifiers used in the ADC. One aspect of averaging is the topology that is used to accomplish averaging, i.e., which amplifier outputs in which arrays of amplifiers are averaged together.
In general, flash, folding and subranging ADC's use cascades of distributed amplifiers to amplify the residue signals before they are applied to the comparators. These residue signals are obtained by subtracting different DC reference voltages from an input signal Vin. The DC reference voltages are generated by the resistive ladder (reference ladder) 104 biased at a certain DC current.
High-resolution ADC's often use auto-zero techniques, also called offset compensation techniques, to suppress amplifier offset voltages. In general, autozeroing requires two clock phases (φ1 and φ2). During the auto-zero phase, the amplifier offset is stored on one or more capacitors, and during the amplify phase, the amplifier is used for the actual signal amplification.
Two different auto-zero techniques can be distinguished, which are illustrated in
The second technique, shown in
Unfortunately, the performance of cascaded arrays of amplifiers degrades significantly at high clock and input signal frequencies. The cause of this degradation is illustrated in
When the amplifier 201 is in the auto-zero phase φ1, the input capacitors C1a, C1b are charged to the voltage Vsample that is provided by the track-and-hold amplifier 101. As a result, a current IC will flow through the input capacitors C1a, C1b and an input switch (not shown). Due to the finite on-resistance RSW of the input switch (see FIG. 4), an input voltage is generated, which will settle exponentially towards zero. This input voltage is amplified by the amplifier 201 and results in an output voltage that also slowly settles towards zero (assuming the amplifier 201 has zero offset).
Essentially, the auto-zero amplifier 201 is in a “reset” mode one-half the time, and in an “amplify” mode the other one-half the time. When in reset mode, the capacitors C1a, C1b are charged to the track-and-hold 101 voltage, and the current IC flows through the capacitors C1a, C1b and the reset switches, so as to charge the capacitors C1a, C1b.
When the ADC has to run at high sampling rates, there is not enough time for the amplifier 201 output voltage to settle completely to zero during the reset phase. As a result, an error voltage is sampled at the output capacitors C2a, C2b that is dependent on the voltage Vsample. This translates into non-linearity of the ADC, and often causes inter-symbol interference (ISI).
The problem of ISI occurs in most, if not all, ADC architectures and various approaches exist for attacking the problem. The most straightforward approach is to decrease the settling time constants. However, the resulting increase in power consumption is a major disadvantage.
Another approach is to increase the time allowed for settling, by using interleaved ADC architectures. However, this increases required layout area. Furthermore, mismatches between the interleaved channels cause spurious tones. The ISI errors can also be decreased by resetting all cascaded amplifiers during the same clock phase. Unfortunately, this is not optimal for high speed operation either.
The present invention is directed to an analog to digital converter topology that substantially obviates one or more of the problems and disadvantages of the related art.
There is provided an analog to digital converter including a reference ladder, a clock having phases φ1 and φ2, and a track-and-hold amplifier tracking an input signal with its output signal during the phase φ1 and holding a sampled value during the phase φ2. A plurality of coarse amplifiers each input a corresponding tap from the reference ladder and the output sign. A plurality of fine amplifiers input corresponding taps from the reference ladder and a signal corresponding to the output signal, the taps selected based on outputs of the coarse amplifiers. A circuit responsive to the clock receives the signal corresponding to the output signal, the circuit substantially passing the signal corresponding to the output signal and the corresponding taps to the fine amplifiers during the phase φ2 and substantially rejecting the signal corresponding to the output signal during the phase φ2. An encoder converts outputs of the coarse and fine amplifiers to an N-bit digital signal representing the input signal.
In another aspect of the present invention there is provided an analog to digital converter including a reference ladder and a two-phase clock having phases φ1 and φ2. A track-and-hold amplifier tracking an input signal with its output signal during the phase φ1 and holding a sampled value during the phase φ2. A plurality of coarse amplifiers each inputting a signal corresponding to the output signal and a corresponding tap from the reference ladder. A switching circuit that receives the signal corresponding to the output signal and has a differential mode transfer function of approximately 1 on the phase φ2 and approximately 0 on the phase φ1. A plurality of fine amplifiers inputting corresponding taps from the reference ladder and the signal corresponding to the output signal through the switching circuit, the taps selected based on outputs of the coarse amplifiers. An encoder converts outputs of the coarse and fine amplifiers to an N-bit digital signal representing the input signal.
In another aspect of the present invention there is provided an analog to digital converter including a reference ladder and a multi-phase clock. A track-and-hold amplifier tracking an input signal with its output signal during one phase of the multi-phase clock and holding a sampled value during another phase of the multi-phase clock. A plurality of coarse amplifiers each inputting a signal corresponding to the output signal and a corresponding tap from the reference ladder. Switching means that receives the signal corresponding to the output signal and responsive to the multi-phase clock, the means substantially passing the signal corresponding to the output signal to the fine amplifiers during the one phase and substantially rejecting the signal corresponding to the output signal during the another phase. A plurality of fine amplifiers inputting, through the switching means, corresponding taps from the reference ladder and the output signal, the taps selected based on outputs of the coarse amplifiers. An encoder converts outputs of the coarse and fine amplifiers to an N-bit digital signal representing the input signal.
In another aspect, an input stage includes a track-and-hold amplifier whose output signal tracks an input signal during one clock phase, and holds a sampled value during another clock phase. A coarse amplifier inputting the output signal and a coarse tap. A transfer matrix that substantially passes a signal corresponding to the output signal during the one clock phase and substantially blocks the signal corresponding to the output signal during the another clock phase. A fine amplifier inputting a fine tap and the output signal through the transfer matrix, the fine tap selected based on an output of the coarse amplifier.
In another aspect, an input stage includes a differential coarse amplifier inputting a signal corresponding to an input signal and a coarse tap during one clock phase, and a sampled value during another clock phase. A plurality of cross-coupled transistors that substantially pass the first signal and a fine tap during the one clock phase and substantially block the first signal and the fine tap during the another clock phase, the fine tap selected based on a signal from the differential coarse amplifier. A differential fine amplifier inputting an output of the plurality of cross-coupled transistors.
Additional features and advantages of the invention will be set forth in the description which follows, and in part will be apparent from the description, or may be learned by practice of the invention. The advantages of the invention will be realized and attained by the structure particularly pointed out in the written description and claims hereof as well as the appended drawings.
It is to be understood that both the foregoing general description and the following detailed description are exemplary and explanatory and are intended to provide further explanation of the invention as claimed.
The accompanying drawings, which are included to provide a further understanding of the invention and are incorporated in and constitute a part of this specification, illustrate embodiments of the invention and together with the description serve to explain the principles of the invention. In the drawings:
Reference will now be made in detail to the preferred embodiments of the present invention, examples of which are illustrated in the accompanying drawings.
Recently, a technique to address the nonlinearity was published by Miyazaki et al., “A 16 mW 30 M Sample/s pipelined A/D converter using a pseudo-differential architecture,” ISSCC Digest of Tech. Papers, pp. 174-175 (2002), see particularly FIG. 10.5.2 therein. The technique applies only to amplifiers that use the auto-zero technique of FIG. 2.
In Miyazaki, four extra switches and two extra capacitors are required. The resulting circuit topology has a common-mode transfer function of “1” and a differential-mode transfer function of “0” during the reset clock phase.
However, an important disadvantage of the circuit shown in Miyazaki is that it requires twice the amount of capacitance. This has a serious impact on the ADC layout area. Furthermore, the capacitive loading of the track-and-hold 101 doubles, which significantly slows down the charging of the capacitors C1a, C1b (roughly by a factor of two).
The problem of ISI can be solved in a very elegant way by complementing the reset switches shown in
The transfer circuit shown in the dashed box 510 has a transfer function of “1” for common-mode signals at all times, so that the common mode transfer function is HCM (φ1)=1, HCM (φ2)=1. However, the transfer function varies for differential signals depending on the clock phase (φ1 or φ2). More specifically, the transfer function for differential signals is HDM(φ1)=0, and HDM (φ2)=1. Hence, a differential voltage created across nodes 1 and 2 (due to the charging of the input capacitors C1a, C1b) is not transferred to input nodes 3 and 4 of the amplifier 201 during φ1. Therefore, the output voltage of the amplifier 201 is not affected by Vsample in any way, reducing the occurrence of ISI. The input capacitors C1a, C1b subtract track-and-hold amplifier 101 voltage from a reference ladder 104 voltage.
The technique presented herein can find application in various types of ADC architectures that use auto-zero techniques for combating amplifier offsets.
Thus, the circuit within the dashed box 510 may be referred to as a transfer matrix that has a property such that its differential mode transfer function H(φ1)=0, H(φ2)=1. This is different from a conventional approach, where the transfer function may be thought of as being H=1 for both φ1 and φ2.
It will be appreciated that while the overall transfer function of the transfer matrix 510 is HDM (φ1)=0, HDM (φ2)=1, HCM (φ1)=1, HCM (φ2)=1, this is primarily due to the switches M1-M4, which essentially pass the differential voltage of nodes 1 and 2 through to nodes 3 and 4 respectively, on φ2. However, the gain factor need not be exactly 1, but may be some other value. The important thing is that it be substantially 0 on φ1.
Note that either PMOS or NMOS transistors may be used as switches in the present invention. Note further that given the use of the FET transistors as switches (rather than the amplifiers), the drain and the source function equivalently.
Note further that in the event of using a plurality of cascaded amplifier stages for a pipeline architecture (designated A, B, C, D), if the A and B stage switches are driven by the phase φ1, and the C and D stages are driven by φ2, the transfer matrix 510 is only needed for the A stage and the C stage. On the other hand, if the switches of the stages A, B, C and D are driven by alternating clock phases (i.e., φ1, φ2, φ1, φ2), each stage will need its own transfer matrix 510.
It will be appreciated that the various aspects of the invention as further disclosed in related application Ser. No. 10/158,595, Filed: May 31, 2002, Titled: HIGH SPEED ANALOG TO DIGITAL CONVERTER, Inventor: Jan Mulder; application Ser. No. 10/153,709, Filed: May 24, 2002, Titled: Distributed Averaging Analog To Digital Converter Topology, Inventors: Mulder et al.; application Ser. No. 10/158,773, filed on May 31, 2002, Titled: Subranging Analog To Digital Converter With Multi-Phase Clock Timing; application Ser. No. 10/158,774, Filed: May 31, 2002; Titled: Analog To Digital Converter With Interpolation of Reference Ladder, Inventors: Mulder et al.; and application Ser. No. 10/158,193, Filed: May 31, 2002, Inventor: Jan Mulder; Titled: CLASS AB DIGITAL TO ANALOG CONVERTER/LINE DRIVER, Inventors: Jan Mulder et al., all of which are incorporated by reference herein, may be combined in various ways, or be integrated into a single integrated circuit or product.
It will be understood by those skilled in the art that various changes in form and details may be made therein without departing from the spirit and scope of the invention as defined in the appended claims. Thus, the breadth and scope of the present invention should not be limited by any of the above-described exemplary embodiments, but should be defined only in accordance with the following claims and their equivalents.
This application is a Continuation of application Ser. No. 10/688,921, Filed: Oct. 21, 2003, now U.S. Pat. No. 6,788,238 Titled: HIGH SPEED ANALOG TO DIGITAL CONVERTER, Inventor: Jan MULDER, which is a Continuation of application Ser. No. 10/349,073, Filed: Jan. 23, 2003, now U.S. Pat. No. 6,674,388 issued Jan. 6, 2004, Titled: HIGH SPEED ANALOG TO DIGITAL CONVERTER, Inventor: Jan MULDER, which is a Continuation of application Ser. No. 10/158,595, Filed: May 31, 2002, now U.S. Pat. No. 6,573,853 Titled: HIGH SPEED ANALOG TO DIGITAL CONVERTER, Inventor: Jan MULDER; which is a Continuation-in-Part of application Ser. No. 10/153,709, Filed: May 24, 2002, now U.S. Pat. No. 6,628,224 Titled: DISTRIBUTED AVERAGING ANALOG TO DIGITAL CONVERTER TOPOLOGY, Inventors: MULDER et al.; is a continuation of application Ser. No. 10/158,773, filed on May 31, 2002, Titled: SUBRANGING ANALOG TO DIGITAL CONVERTER WITH MULTI-PHASE CLOCK TIMING; application Ser. No. 10/158,774, Filed: May 31, 2002; Titled: ANALOG TO DIGITAL CONVERTER WITH INTERPOLATION OF REFERENCE LADDER, Inventors: MULDER et al.; and application Ser. No. 10/158,193, Filed: May 31, 2002, Inventor: Jan MULDER; Titled: CLASS AB DIGITAL TO ANALOG CONVERTER/LINE DRIVER, Inventors: Jan MULDER et al., all of which are incorporated by reference herein.
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Number | Date | Country | |
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20040257255 A1 | Dec 2004 | US |
Number | Date | Country | |
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Parent | 10688921 | Oct 2003 | US |
Child | 10893999 | US | |
Parent | 10349073 | Jan 2003 | US |
Child | 10688921 | US | |
Parent | 10158595 | May 2002 | US |
Child | 10349073 | US | |
Parent | 10158773 | May 2002 | US |
Child | 10153709 | US | |
Parent | 10158774 | May 2002 | US |
Child | 10158773 | US | |
Parent | 10158193 | May 2002 | US |
Child | 10158774 | US |
Number | Date | Country | |
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Parent | 10153709 | May 2002 | US |
Child | 10158595 | US |