High stability gain structure and filter realization with less than 50 ppm/° c. temperature variation with ultra-low power consumption using switched-capacitor and sub-threshold biasing

Abstract

An ultra-low power sub-threshold gm stage is disclosed where transconductance is very stable with process, temperature, and voltage variations. This technique can be implemented in a differential amplifier with constant gain and a second order biquad filter with constant cut off frequency. The amplifier gain can achieve a small temperature coefficient of 48.6 ppm/° C. and exhibits small sigma of 75 mdB with process. The second order biquad can achieve temperature stability of 69 ppm/° C. and a voltage coefficient of only 49 ppm/mV.

Description

BACKGROUND

Internet-of-Things (IoT) envisions a large scale deployment of ultra-low power (ULP) electronic devices integrated in our environment to perform meaningful sensing and communication. However, existing IoT devices have lower available power and can perform only a limited amount of sensing, processing, and communication, of which communication has the highest power consumption. To reduce the power consumption needed for communication, the amount of data must be reduced, which can be done by incorporating computing at the edge in IoT [1].

A few recently published IoT devices achieve micro-watt level power consumption [2]. However, system power consumption and area remains high for applications needing miniaturized devices while operating from harvested energy. Analog computing and processing can achieve both lower power and lower area. Gain structures such as differential amplifiers, op-amps, filters, etc. are the fundamental building blocks of an analog computer. However, these circuit elements suffer from inaccuracies and drifts due to process, voltage, and temperature (PVT) variations.

Transconductance, g_m of a current biased transistor or a differential pair is a key parameter that impacts the gain, bandwidth, and other critical parameters in analog circuits. A transistor biased with a current source is typically used to generate g_m. A constant g_m can be generated by using the current biases generated using the circuits shown in FIG. 1A and FIG. 1B. A transistor is biased such that its g_m is inversely proportional to a precise resistor R_X as shown in FIG. 1A ([3], [4]). However, this approach requires precise off-chip resistors to make g_m immune to process variations, which leads to large area and cost overhead. Also, these circuits are based on the square law model of MOSFETs, which is not accurate for modern-day short channel devices. In [5], authors introduce a novel circuit to address the supply variation, but require an off-chip resistor. Off-chip implementation of resistance gives rise to large parasitic capacitance C_P in parallel with R_X. This forms a pole-zero pair at low frequencies which can cause the system to become unstable [6]. Hence, a large compensation capacitor is needed to ensure stability of the constant g_m circuit.

Several on-chip solutions have been presented in recent years where the resistor in the conventional circuit is replaced with a transistor operating in triode region [7]-[10]. In [7], the MOSFET is biased in the linear region with current and voltage references to track the g_m of a transistor. However, the circuit suffers from high power consumption in the μWs range. The work in [8] also uses a voltage reference to bias the linear region transistor but has a high temperature coefficient. [10] uses a PT invariant current source to bias a transistor at the voltage where its on-resistance is invariant to temperature. [9] uses a master-slave topology to tune the on-chip resistor to its desired value. A switched capacitor network is used as the reference resistor which controls the biasing of the MOS in triode region via a feedback loop. The on-chip resistor can also be tuned via a PLL locked to a frequency reference. In [11], a precise crystal oscillator for precise tuning of the resistor is used to realize a constant g_m circuit. In [12], the author presents a differential pair in which the tail current is adjusted by a feedback loop such that g_m of the input transistors is proportional to 1/R. Although this circuit does not depend on the square law of the MOSFET, it requires an external resistor. [13] also proposes using an external resistor to obtain constant g_m bias based on analog computation technique. [14] describes a complementary constant g_m biasing scheme of Nauta transconductors for g_m—C filter applications. Here, the conventional beta-multiplier circuit is modified to include a PMOS transistor to provide a voltage bias. The resistor is trimmed to account for process and temperature variations. Also, the beta multiplier circuits work on the square law assumption, which is inaccurate for short channel transistors or for ultra-low power sub-threshold operation.

Various embodiments disclosed herein relate to a constant g_m bias circuit operating in the sub-threshold region for low power consumption. Although sub-threshold operation can achieve very low power consumption, it is traditionally associated with PVT variations. However, stable and precise outputs for various fundamental analog circuits can be obtained using sub-V_th operation ([15], [16]). In the present application, the resistor in a traditional current source is replaced with a switched capacitor resistor (SCR). This biasing scheme is then used for the biasing of a differential amplifier and a second order biquad filter to demonstrate proof of concept. This biasing configuration was initially theorized for biasing saturation region MOS transistors in [17], but was considered impractical by [18] due to ripple and other second order effects. However, this technique has merit in sub-threshold operation where low frequency operation is desired for ULP operation. This application discloses sub-threshold biasing and shows that the ripple of this circuit can be minimized and a very precise realization of filter and gain structure can be achieved whose stability can be compared to conventional bandgap reference voltage. Simulation results show that we can achieve a temperature stability of 48.6 ppm/° C. for gain of a single stage differential amplifier realization. Similar biasing circuit is also used to implement a second order filter circuit, which achieved a temperature stability of 69 ppm/° C.

BRIEF SUMMARY OF THE DISCLOSURE

Various embodiments disclosed herein relate to an ultra-low power sub-threshold g_m stage where transconductance is very stable with process, temperature, and voltage variations. This technique can be implemented, e.g., in a differential amplifier with constant gain and a second order biquad filter with constant cut off frequency. The amplifier gain can achieve a small temperature coefficient of 48.6 ppm/° C. and exhibits small sigma of 75 mdB with process. The second order biquad can achieve temperature stability of 69 ppm/° C. and a voltage coefficient of only 49 ppm/mV.

A method in accordance with one or more embodiments comprises generating a biasing current (I₀) using a constant gm bias circuit operating in the subthreshold region for ultra-low power consumption. The circuit includes a switched capacitor resistor having a first capacitor, wherein the biasing current (I₀) is given by: I₀=V_T ln(M)/R₁, where V_T is the thermal voltage of the constant g_m bias circuit, where the thermal voltage is kT/q, where k is the Boltzman's constant, T is the temperature in Kelvin, and q is the unit of charge, M is a multiplication factor relating two transistors in the constant gm bias circuit, R₁=1/f₁C₁, where C₁ is a capacitance of the first capacitor, and f₁ is a frequency of a reference clock of the constant gm bias circuit.

A differential amplifier circuit in accordance with one or more embodiments includes a constant gm bias circuit operating in the subthreshold region for ultra-low power consumption generating a biasing current (I₀), said constant gm bias circuit including a switched capacitor resistor having a first capacitor, wherein the biasing current (I₀) is given by: I₀=V_T ln(M)/R₁, where V_T is the thermal voltage of the constant g_m bias circuit, where the thermal voltage is kT/q, where k is the Boltzman's constant, T is the temperature in Kelvin, and q is the unit of charge, M is a multiplication factor relating two transistors in the constant gm bias circuit, R₁=1/f₁C₁, C₁ is a capacitance of the first capacitor, and f₁ is a frequency of a reference clock of the constant gm bias circuit. A differential amplifier is biased with the biasing current, the differential amplifier including a load switched capacitor resistor having a second capacitor, wherein the differential amplifier has constant gain (A_v) given by: A_v=ln(M)f₁C₁/ηf₃C₃, where η is a process constant value for the differential amplifier, C₃ is a capacitance of the second capacitor, and f₃ is a frequency of the load switched capacitor resistor of the differential amplifier.

A second-order biquad filter in accordance with one or more embodiments includes a constant gm bias circuit operating in the subthreshold region for ultra-low power consumption generating a biasing current (I₀). The constant gm bias circuit includes a switched capacitor resistor having a first capacitor, wherein the biasing current (I₀) is given by: I₀=V_T ln(M)/R₁, where V_T is the thermal voltage of the constant g_m bias circuit, where the thermal voltage is kT/q, where k is the Boltzman's constant, T is the temperature in Kelvin, and q is the unit of charge, M is a multiplication factor relating two transistors in the constant gm bias circuit, R₁=1/f₁C₁, where C₁ is a capacitance of the first capacitor, and f₁ is a frequency of a reference clock of the constant gm bias circuit. A second order biquad filter is biased with the biasing current to obtain a stable cut-off frequency.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1A shows an exemplary conventional MOS-based current source for constant gm biasing.

FIG. 1B shows an exemplary BJT-based current source for constant gm biasing.

FIG. 2A shows an exemplary RC filter for ripple suppression implemented using transistors.

FIG. 2B shows an exemplary differential amplifier with an SCR and decoupling capacitors in accordance with one or more embodiments.

FIG. 2C shows an exemplary operational transconductance amplifier (OTA) in accordance with one or more embodiments.

FIG. 2D shows an exemplary second order biquad filter in accordance with one or more embodiments.

FIG. 3 shows amplifier output before and after filtering with an ideal filter.

FIGS. 4A and 4B show differential amplifier gain variance with temperature and voltage, respectively.

FIG. 5A shows process variation of differential amplifier gain at 50□, and FIG. 5B shows process variation of cut off frequency of a biquad filter.

FIGS. 6A and 6B show variation of cut off frequency of a biquad LPF with temperature and supply voltage, respectively.

FIGS. 7A and 7B show variation of unity gain frequency of OTA with temperature and process, respectively.

FIG. 8 is a table showing performance comparisons between known constant gm bias circuits and exemplary circuits in accordance with various embodiments.

DETAILED DESCRIPTION

Precision Analog Gain Structure and Filter Implementation

The transconductance, g_m of a transistor in sub-threshold is given by

$\begin{array}{cc}{g}_m=\frac{I_D}{\eta V_T}& \left(1\right)\end{array}$

where I_D is the bias current, η is a process constant, and V_T is the thermal voltage. The transconductance is inversely proportional to temperature. If the transistor is biased with a current source that is proportional to absolute temperature (PTAT), then a constant g_m can be obtained as

$\begin{array}{cc}{g}_m=\frac{I_{\mathrm{PTAT}}}{\eta V_T}& \left(2\right)\end{array}$

PTAT current is obtained using the conventional BJT based current source with a resistor R₁ shown in FIG. 1B, where Q₁ has M times the area of Q₂.I₀=V_T ln(M)/R₁  (3)

• • where V_T is the thermal voltage of the constant g_m bias circuit, where the thermal voltage is kT/q, where k is the Boltzman's constant, T is the temperature in Kelvin, and q is the unit of charge,
  • M is a multiplication factor relating two transistors in the constant gm bias circuit,R₁=1/f₁C₁,
  • C₁ is a capacitance of the first capacitor, and
  • f₁ is a frequency of a reference clock of the constant gm bias circuit.

When I_o is used to bias a transistor in sub-threshold, its transconductance can be calculated using Eq. (2) and (3) asg_m=ln(M)/ηR₁  (4)

• • which has a lower variation as it only depends on R₁.

The conventional MOS-based current source shown in FIG. 1A can also be biased in the sub-threshold region to obtain a PTAT current. This is because the current through a MOS biased in sub-threshold region is,

$\begin{array}{cc}{I}_{\mathrm{DS}}=I_{S0}\exp \left(\frac{V_{\mathrm{GS}}-V_{\mathrm{th}}-V_{\mathrm{off}}}{\eta V_T}\right)\left(1-\exp \left(-\frac{V_{\mathrm{DS}}}{V_T}\right)\right)& \left(5\right)\end{array}$

• • where I_S0 is the sub-threshold saturation current and V_off is the offset voltage as discussed in [19] and [20]. The current characteristic is exponential, similar to that of a BJT. However, we use BJTs for a more precise operation. Assuming V_DS>3 V_T in a circuit configuration shown in FIG. 1A which is biased in sub-threshold, the offset V_off in equation (5) gives rise to the following current

$\begin{array}{cc}{I}_o=\frac{V_T\ln \left(K\right)}{R}+\mathrm{const}& \left(6\right)\end{array}$

The constant term, const, is independent of temperature, which makes transconductance vary with temperature. A BJT based PTAT current source does not suffer from this issue and provides better stability.

Resistor implementation is susceptible to process and temperature variations. Also, for small current generation in the nAs range, a very large resistor is needed, which can be difficult to implement. These challenges are overcome by replacing the resistor with a switched capacitor implementation. The SCR implementation is achieved by using MIM cap and a reference frequency clock. MIM caps have a small temperature coefficient of 35 ppm/° C. [21] and small process variations. Also, the stable frequency reference which is used for SCR network is a fundamental component in most systems. It has a stability of less than 3-5 ppm/° C. and can operate with a power consumption of 1-2 nW [22]. Since large resistors in the range of Mega-Ω s are required to generate small current for sub-threshold biasing, the area of the switched capacitor design can be much lower than the resistor based design while providing high degree of accuracy. Also, since the same bias circuit is used to bias several other circuits on the chip, the overhead is not large.

The resistance of the SCR is given by

$\begin{array}{cc}R=\frac{1}{f_1{C}_1}& \left(7\right)\end{array}$ where f₁ is the frequency of the reference clock and C₁ is the capacitor.

From Eq. (4) and (7), transconductance is now given byg_m=ln(M)f₁C₁/η  (8)

Eq. 8 shows that g_m is now a function of capacitor and reference frequency. Decoupling capacitors (C_d) of 8 pF are added at nodes A and B in FIG. 1B to suppress the ripple originating from the SCR and also to ensure that the resistance of the network remains constant with temperature. Since nodes V_X and V_Y are used to bias other circuits, RC filters are used to filter out the ripple. The filter needed for ripple suppression can be implemented using transistors (FIG. 2A) or with active filters to save area. The op-amp in the current source also helps in filtering out ripple at V_X as it is designed to have small bandwidth.

One source of temperature instability in an SCR based circuit can manifest from the charge injection through the gates of SCR circuits. We use small size switches with dummy switches for charge injection cancellation for better stability with temperature and voltage.

Differential Amplifier

An exemplary single stage differential amplifier with resistive loads in accordance with one or more embodiments depicted in FIG. 2B is biased with the described constant g_m biasing scheme. For low power consumption, the amplifier is designed in the sub-threshold region and is biased with 50 nA current. The gain A_V of the amplifier is given by,A_v=g_mR_D  (9)

where R_D is the load resistor. A gain of 20 dB with this single stage amplifier requires large resistors of ˜18 MΩ at the load to obtain an output common mode level of 500 mV. Here, the large resistors are replaced with switched capacitor resistors to achieve the same gain (FIG. 2B). The differential amplifier was designed with both a poly-resistor and SCR with 600 fF flying capacitor, and 3 pF decoupling capacitors to estimate area. The area of the poly-resistor amplifier was 1.3× higher area than the SCR implementation. Even lower area and lower power solution can be envisioned with SCR. The gain with SCR load is given byA_v=ln(M)f₁C₁/ηf₃C₃  (10)

• • where f₃ is the frequency of the load SCR of the amplifier.

Small flying capacitor and high frequency of 90 kHz is used to emulate the load resistor. This is to ensure that the ripple frequency from the load SCR falls outside the bandwidth of the amplifier. The small ripple introduced by the SCR can be filtered out using a filter (FIG. 3). Also, the frequency for the load SCR can be generated from the reference clock on the chip. From eq. (10), it is clear that gain is constant and the only process dependent parameter is η. Capacitors show little variation with process, and are in a ratio that is reliable even with variations. FIGS. 4A and 4B show the temperature and voltage variation of the amplifier gain, and FIG. 5A shows variation with process. The amplifier gain (in V/V) has a temperature coefficient of less than 50 ppm/° C., and varies by 1.27% for a 25% increase in supply.

Second Order Filter

An exemplary second order biquad filter in accordance with one or more embodiments shown in FIG. 2D is biased with the constant g_m biasing circuit to obtain stable cut-off frequency. The filter includes four identical operational transconductance amplifiers (OTAs) (FIG. 2C) biased with sub-threshold PTAT current, whose unity gain frequency is given by

$\begin{array}{cc}UGF=\frac{g_m}{C_2}& \left(11\right)\end{array}$

From Eq (8) and (11)UGF=ln(M)f₁C₁/ηC₂  (12)

Eq. (12) shows that the UGF is now completely dependent on the ratio of C₁ and C₂ which implies that it is only dependent on reference clock frequency and η to achieve a very high stability. Variation of UGF with temperature is shown in FIG. 7A, which is less than 0.7%. The cut-off frequency of the biquad filter shown in FIG. 2D is given by ([23]),

$\begin{array}{cc}{\left({\omega }_o\right)}^2=\frac{g_{m3}{g}_{m4}}{{\left(C_4\right)}^2}& \left(13\right)\end{array}$

Hence, replacing transconductance by Eq. (8), and assuming all transconductances are equal,Wo=ln(M)f₁C₁/ηC₄  (14)

The cut off frequency is independent of temperature and supply as shown in FIG. 6A and FIG. 6B. The temperature and voltage coefficients of the filter are found to be 69 ppm/° C. and 49 ppm/mV respectively. Table I shown in FIG. 8 compares our work with prior-works. The work in [16] shows better stability than our work but relies on an external voltage source to fine-tune the PTAT current source. The work in [15] also uses sub-threshold biasing to realize high stability but can only achieve lower bandwidth of 12 Hz. Our work achieves higher performance and reduces the power consume when compared to a more related recent prior art [9]. We also report a very wide temperature range of operation of 0-120° C. when compared to [14] and [10].

In summary, the present application discloses a methodology for constant g_m biasing where the circuits are biased in sub-threshold region with a proportional to absolute temperature current to obtain constant transconductance which is highly stable with temperature, supply and process variations. A differential amplifier with constant gain, and a second order biquad filter with constant cut off frequency disclosed demonstrate proof of concept. The differential amplifier achieves 49 ppm/° C. temperature coefficient for the gain, which varies by 1.27% with a 25% increase in supply. Simulations also show that the with a temperature coefficient of 69 ppm/° C.

Having thus described several illustrative embodiments, it is to be appreciated that various alterations, modifications, and improvements will readily occur to those skilled in the art. Such alterations, modifications, and improvements are intended to form a part of this disclosure, and are intended to be within the spirit and scope of this disclosure. While some examples presented herein involve specific combinations of functions or structural elements, it should be understood that those functions and elements may be combined in other ways according to the present disclosure to accomplish the same or different objectives. In particular, acts, elements, and features discussed in connection with one embodiment are not intended to be excluded from similar or other roles in other embodiments. Additionally, elements and components described herein may be further divided into additional components or joined together to form fewer components for performing the same functions.

Accordingly, the foregoing description and attached drawings are by way of example only, and are not intended to be limiting.

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Claims

1. A method comprising: generating a biasing current (I0) using a constant gm bias circuit operating in the subthreshold region for ultra-low power consumption, said circuit including a switched capacitor resistor having a first capacitor, wherein the biasing current (I0) is given by: I0=VT ln(M)/R1,where VT is the thermal voltage of the constant gm bias circuit, where the thermal voltage is kT/q, where k is the Boltzman's constant, T is the temperature in Kelvin, and q is the unit of charge,M is a multiplication factor relating two transistors in the constant gm bias circuit, R1=1/f1C1,C1 is a capacitance of the first capacitor, andf1 is a frequency of a reference clock of the constant gm bias circuit; andbiasing a differential amplifier with the biasing current, the differential amplifier including a load switched capacitor resistor having a second capacitor, wherein the differential amplifier has constant gain (Av), independent of manufacturing process variation or temperature variation, given by: Av=ln(M)f1C1/ηf3C3,where η is a process constant value for the differential amplifier,C3 is a capacitance of the second capacitor, andf3 is a frequency of the load switched capacitor resistor of the differential amplifier.

2. The method of claim 1, wherein the method is implemented in a ultra-low power and lower area Internet of things (IoT) device.

3. A differential amplifier circuit, comprising: (a) a constant gm bias circuit operating in the subthreshold region for ultra-low power consumption generating a biasing current (I0), said constant gm bias circuit including a switched capacitor resistor having a first capacitor, wherein the biasing current (I0) is given by: I0=VT ln(M)/R1,where VT is the thermal voltage of the constant gm bias circuit, where the thermal voltage is kT/q, where k is the Boltzman's constant, T is the temperature in Kelvin, and q is the unit of charge,M is a multiplication factor relating two transistors in the constant gm bias circuit, R1=1/f1C1,C1 is a capacitance of the first capacitor, andf1 is a frequency of a reference clock of the constant gm bias circuit; and(b) a differential amplifier biased with the biasing current, the differential amplifier including a load switched capacitor resistor having a second capacitor, wherein the differential amplifier has constant gain (Av) given by: Av=ln(M)f1C1/ηf3C3 where η is a process constant value for the differential amplifier;C3 is a capacitance of the second capacitor, andf3 is a frequency of the load switched capacitor resistor of the differential amplifier.

4. The differential amplifier circuit of claim 3, wherein the differential amplifier circuit is implemented in a ultra-low power and lower area Internet of things (IoT) device.

5. The differential amplifier circuit of claim 3, wherein the constant gm bias circuit includes one or more decoupling capacitors for suppressing ripple originating from the switched capacitor resistor in the constant gm bias circuit.

6. A second-order biquad filter, comprising: (a) a constant gm bias circuit operating in the subthreshold region for ultra-low power consumption generating a biasing current (I0), said constant gm bias circuit including a switched capacitor resistor having a first capacitor, wherein the biasing current (I0) is given by: I0=VT ln(M)/R1,where VT is the thermal voltage of the constant gm bias circuit, where the thermal voltage is kT/q, where k is the Boltzman's constant, T is the temperature in Kelvin, and q is the unit of charge,M is a multiplication factor relating two transistors in the constant gm bias circuit, R1=1/f1C1,C1 is a capacitance of the first capacitor, andf1 is a frequency of a reference clock of the constant gm bias circuit; and(b) a second order biquad filter biased with the biasing current to obtain a stable cut-off frequency.

7. The second-order biquad filter of claim 6, wherein the second order biquad filter comprises four operational transconductance amplifiers, each transconductance amplifier having a second capacitor, wherein the unity gain frequency (UGF) of the second order biquad filter given by: UGF=ln(M)f1C1/ηC2 where η is a constant value for the amplifiers, andC2 is a capacitance of the second capacitor.

8. The second-order biquad filter of claim 6, wherein the second order biquad filter includes a plurality of third capacitors, and wherein the cut-off frequency of the second order biquad filter is given by Wo=ln(M)f1C1/ηC4,where η is a constant value for the amplifiers, andwhere C4 is a capacitance of the second capacitor.

9. The second-order biquad filter of claim 6, wherein second-order biquad filter is implemented in a ultra-low power and lower area Internet of things (IoT) device.

10. The second-order biquad filter of claim 6, wherein the constant gm bias circuit includes one or more decoupling capacitors for suppressing ripple originating from the switched capacitor resistor in the constant gm bias circuit.

11. A method comprising: generating a biasing current (I0) using a constant gm bias circuit operating in the subthreshold region for ultra-low power consumption, said circuit including a switched capacitor resistor having a first capacitor, wherein the biasing current (I0) is given by: I0=VT ln(M)/R1,where VT is the thermal voltage of the constant gm bias circuit, where the thermal voltage is kT/q, where k is the Boltzman's constant, T is the temperature in Kelvin, and q is the unit of charge,M is a multiplication factor relating two transistors in the constant gm bias circuit, R1=1/f1C1,C1 is a capacitance of the first capacitor, andf1 is a frequency of a reference clock of the constant gm bias circuit, andwherein generating a biasing current (I0) includes suppressing ripple originating from the switched capacitor resistor in the constant gm bias circuit using one or more decoupling capacitors.

12. A method comprising: generating a biasing current (I0) using a constant gm bias circuit operating in the subthreshold region for ultra-low power consumption, said circuit including a switched capacitor resistor having a first capacitor, wherein the biasing current (I0) is given by: I0=VT ln(M)/R1,where VT is the thermal voltage of the constant gm bias circuit, where the thermal voltage is kT/q, where k is the Boltzman's constant, T is the temperature in Kelvin, and q is the unit of charge,M is a multiplication factor relating two transistors in the constant gm bias circuit, R1=1/f1C1,C1 is a capacitance of the first capacitor, andf1 is a frequency of a reference clock of the constant gm bias circuit; andbiasing a second order biquad filter with the biasing current to obtain a stable cut-off frequency.

13. The method of claim 12, wherein the second order biquad filter comprises four operational transconductance amplifiers, each transconductance amplifier having a second capacitor, wherein the unity gain frequency (UGF) of the second order biquad filter given by: UGF=ln(M)f1C1/ηC2 where η is a constant value for the amplifiers, andC2 is a capacitance of the second capacitor.

14. The method of claim 13, wherein the second order biquad filter includes a plurality of third capacitors, and wherein the cut-off frequency of the second order biquad filter is given by Wo=ln(M)f1C1/ηC4,where η is a constant value for the amplifiers, andwhere C4 is a capacitance of the second capacitor.

15. The method of claim 12, wherein the method is implemented in a ultra-low power and lower area Internet of things (IoT) device.