The non-isolated switching DC-to-DC converters can be broadly divided into three basic categories based on their input to output DC voltage conversion characteristics: a) step-down only (buck converter), step-up only (boost converter) and step-down/step-up such as flyback, SEPIC, and Ćuk converters (1,2). This invention relates to the step-down class of switching DC-to-DC power converters such as buck converter.
Many Point of Load Applications (POL) and Voltage Regulator Modules (VRM's) require a rather large step-down conversion ratios, such as 12:1 or even 24:1 to convert the standard 12V input voltage to 1V or 0.5V output regulated voltage required by the modern microprocessors and other electronic loads. This invention also relates to this particular subset of the step-down converters. However, it is equally applicable to a broader class of other moderate to high step-down voltage conversions.
Classifications of currently known switching converters can also be made based on the type of the voltage and current waveforms exhibited by the switches into three broad categories:
The present invention creates an entirely different new fourth category of the hybrid-switching converters consisting of a resonant inductor and a resonant capacitor forming a resonant circuit for a part of a switching period and a hybrid transformer obeying square-wave switching laws over the entire switching period. Because of the mixed use of the square-wave switching and unique resonant inductor switching a term hybrid-switching method is proposed for this new switching power conversion method. The resonant capacitor takes a dual role, as it forms a resonant circuit during OFF-time interval with the resonant inductor, while during ON-time interval operates like a capacitive energy storage and transfer device such as, for example, in the Ćuk converter (1,2), but here being charged by the magnetizing inductance of the hybrid transformer and not input and output PWM inductors as in Ćuk converter.
Another classification can be made with respect to number of switches used, such as two, four, six etc. The present Square-Wave Switching or Pulse Width Modulated (PWM) switched-mode power conversion theory (and their resonant modifications described above) a-priori excludes the converter topologies with the odd number of switches, such as 3 switches, 5 switches (5). The PWM switching method is based on the classical square-wave switching characterized by square-wave like current and voltage waveforms of its switches over the entire switching period. The direct consequence is that switches come in complementary pairs: when one switch is closed its complementary switch is open and vice versa. Thus when half of the switches are ON their complementary switches are OFF and vice versa for second OFF-time interval. Thus, the converters are characterized by two distinct switching intervals (ON-time interval and OFF-time interval) and even number of switches, such as 2, 4, 6, and cannot have an odd number of switches, such as 3, 5, etc.
The present invention breaks the new ground by introducing the switching converters featuring three switches, which results in hybrid switched-mode power conversion method. Despite the clear use of the resonant capacitor discharge during the OFF-time interval and half-sinusoidal resonant discharge current, the output DC voltage is controlled by a simple duty ratio control and NOT by use of the conventional resonant simple control of output DC voltage by duty ratio D control and not by conventional resonant control methods. Furthermore, unlike in the conventional resonant converters, the output DC voltage is to the first order independent of the DC load current and dependent on duty ratio D only as in conventional PWM converters.
The present invention also breaks another new ground by using a hybrid transformer in a dual role of transferring inductive and capacitive energy storage through it. Present modifications of the buck converter such as tapped-inductor buck use tapped inductor to transfer inductive energy storage only to the output but do not have capacitive energy storage.
The main objective is to provide an alternative to the present buck converter and tapped inductor buck converter to provide the converter with large step-down conversion ratios needed, such as 24:1 and achieve that with much improved efficiency while providing simultaneously magnetic size reductions and a fast transient response. This is achieved by providing step-down converter with a hybrid transformer, which in addition to inductive energy transfer of tapped-inductor buck converter, provides a simultaneous transfer of the resonant capacitor discharge current to the load via the same two winding magnetic structure but now operating as a true ac transformer during the OFF-time interval, hence the proposed name hybrid transformer. Both energy transfer mechanisms provide the increased total current to the load at any duty ratio thereby increasing voltage step-down significantly.
Although one of the main applications of the present invention is for the large step-down (12:1) or higher and low output voltages such as 1V or lower, the same advantages described are also applicable to other output voltages and moderate step-down conversion ratios such as 48V to 12V and 15V to 3.3V conversion.
The following notation is consistently used throughout this text in order to facilitate easier delineation between various quantities:
a illustrates a prior-art buck converter,
a shows input current of the buck converter in
a shows the prior-art tapped-inductor buck converter,
a shows the family of the DC voltage gains for different turns ratio n of tapped-inductor buck converter in
a illustrates the input current of the converter in
a shows a first embodiment of the present invention and
a illustrates an all MOSFET implementation for the three switches of the converter in
a illustrates a linear switched network for converter of
a illustrates one equivalent linear circuit model for linear switched network of
a illustrates simplified circuit model of
a illustrates a voltage waveform across the N turns of the hybrid transformer of the converter in
a shows the family of DC voltage gain characteristics for different hybrid transformer turns ratio's n in the converter of
a illustrates converter circuit of
a illustrates one embodiment of converter in
a illustrates one embodiment of converter in
a illustrates the salient waveforms for the prototype of the converter in
a illustrates the salient waveforms for the prototype of the converter in
a illustrates the salient waveforms for the prototype of the converter in
a illustrates the salient waveforms for the prototype of the converter in
a shows the two-phase extension of the present invention in which two modules are operated in parallel but phase shifted for half a period in order to obtained the reduced output ripple voltage and
a shows the output current waveforms of the two modules in
a shows the output current waveforms of the two modules in
a illustrates another embodiment of present invention, and
a illustrates the converter circuit for OFF-time interval for the converter in
a illustrates a voltage waveform across N turns of hybrid transformer of the converter in
a shows the family of the DC voltage gain plots obtained for different hybrid transformer turns ratio's n for the converter of
a shows the converter circuit of
Prior-art Buck Converter
The non-isolated prior-art Pulse Width Modulated (PWM) buck switching converter shown in
The minimum implementation of semiconductor switches in buck converter is shown on
V/Vg=D (1)
There are three fundamental problems associated with the buck converter when it is required to operate at a large step-down conversion ratios such as 12:1 and 24:1 as needed for modern microprocessors requiring 1V or 0.5V voltage from a 12V input source:
Some of these problems are alleviated in the prior-art tapped-inductor buck converter, but new problems seriously effecting efficiency are introduced as described next.
Prior-art Tapped-Inductor Buck Converter
In order to solve the problem of the prior-art buck converter which must operate at 4% duty ratio to achieve the large 24:1 step-down conversion needed, a prior-art tapped-inductor buck converter of
The following definition of the tapped-inductor in
Turns N1 and N2 of the tapped-inductor and their dot connections are made with reference to their designations in
N=N1+N2 (2)
wherein N is an integer number for primary number of turns of the tapped-inductor and N2 is another integer number for number of turns of the secondary of the tapped-inductor. Note that this makes N2 turns common to both primary and secondary windings. Note also when switch S is turned-OFF there is no current in N1 turns and the inductive energy stored in the tapped-inductor magnetizing inductance during ON-time interval is released to the load during OFF-time interval.
In special applications requiring large step-down and low output voltage, the small size of tapped-inductor indicates that the secondary winding turns N2 can be reduced just to one turn:
N2=1 (3)
so that the turns ratio n can now be defined as:
n=N/N2=N (4)
We will use this turns ratio n as parameters in subsequent analysis and comparisons. However, the turns N1 and N2 will also be invoked at some instances, where the reference is needed to particular windings to refer to the current flowing through them or voltage across them.
The voltage waveform on the primary side of the tapped-inductor with N turns as defined in
VgDTS=Vn(1−D)TS (5)
M=V/Vg=D/(n−nD+D) (6)
where M is a DC voltage gain as a function of the duty ratio D and the turns ratio “n”. The family of the DC voltage gains for increasing values of integer value “n” from 1, 2, 3, 4 etc. is shown by graphs in
However, all of these advantages are far outweighed by the fundamental problem associated with this converter topology. The practical tapped-inductor does have associated with it a leakage inductance L1, which is illustrated in the tapped-inductor model of
E=½L1Ip2 (7)
P1=EfS (8)
Where Ip is the primary current at the instant of the turn-OFF of the main switch S and P1 is corresponding power loss, which is proportional to switching frequency. This will clearly reduce efficiency significantly and prevent increase of the switching frequency in order to reduce the size of the tapped-inductor. As the present buck converters designed for Voltage Regulator Modules for 12V to 1V conversion already operate at 1 MHz switching frequency, this limitation is the main reason that tapped-inductor buck is rarely if at all used.
In addition to above power loss, the additional problem is also seen in
The tapped-inductor therefore does provide an additional step-down in voltage conversion ratio from primary to secondary winding as per (6), but it also produces at a transition point an unwanted jump in instantaneous current during the transition form ON-time interval to OFF-time interval as seen in waveforms of the input current (
It is also obvious that adding a separate external inductor Lr in series with the primary of the tapped-inductor would magnify the power loss problem by an even larger magnitude directly proportional to the size of the external inductor.
Basic Operation of the Hybrid-Switching Step-down Converter with a Hybrid Transformer
The present invention, however, does exactly that, as seen in
Furthermore, this discharge is not abrupt but smooth, as the resonant capacitor Cr is provided in series with the resonant inductor to provide such a smooth and continuous discharge path. Even a small leakage inductance was a big problem in the prior-art tapped-inductor buck converter. The large resonant inductor at the same position in the present invention is, however, a part of the solution, which does not dissipate the energy stored on this inductance, but instead sends it to the load in a non-dissipative way as will be explained in the later section.
The present invention is, therefore, shown in
We now define another turn's ratio m, the current conversion ratio of the hybrid transformer as:
m=N1/N2 (9)
which has an additional role of amplifying the capacitor resonant discharge current by this turns ratio and deliver it via hybrid transformer secondary turns N2 to the load during the OFF-time interval.
For low voltage applications an all n-channel MOSFET implementation shown in
The other two switches S1 and S2 will in comparison have much-reduced rms currents. Note also their desirable connection, so that as seen in
The switching topology of
Such a configuration with three switches is not possible in conventional square-wave PWM and conventional true resonant switching converters (1,2,5). However, here it is essential for its operation and is made possible by the new hybrid-switching method, which uses a unique combination of the square-wave switching and resonant switching.
The switching topology of
However, crucial to the present invention of
Two Windings Coupled on the Common Magnetic Core
Although it appears that the two windings coupled on the common magnetic core could have one and only one interpretation, this is not the case as the following analysis of the presently known two winding magnetic structures are reviewed. This will also serve as the definition of the terms, which will be from here on used in describing the magnetic structure used in the present invention.
Transformer and Autotransformers in General
Faraday discovered in 1831 a principle of magnetic induction of two windings and was therefore also the inventor of the transformer used today commonly in utility AC line power transmission. The transformer, as discovered by Faraday, is a magnetic device, which does not store energy, except for the very small fraction of the input current (1% or less) circulating in transformer magnetizing inductance which is needed to establish the magnetic flux in the core and enable instantaneous transfer of the input ac power to output ac power. As there is no energy stored, the magnetic core coupling the two windings is made of high permeability magnetic material and has no air-gap thus resulting in high magnetizing inductance and low magnetizing current.
Such transformer is also capable via winding turns ratio to step-up or step-down the input ac voltage. It also provides a galvanic isolation between primary and secondary windings important for safety protection from the high voltage primary potential. An autotransformer connection can be used when galvanic isolation is not needed in which case the primary and secondary winding have one common terminal. The other terminal of the secondary winding is then provided as a tap on the primary winding. Note that we will for this case reserve the autotransformer name to indicate a magnetic structure with no energy storage.
Therefore, these true ac transformers and autotransformers operate with bi-directional magnetic flux and corresponding bi-directional magnetic flux density B as shown in
Transformers as used in Switching Converters
Ćuk-type Transformer and Bridge-type Transformers
In switching converters, the transformers with such bi-directional flux capabilities and BH loop also exist, such as the transformer in the Ćuk converter (single ended transformer) which is designated as new converter in the
Forward Converter Transformer Type
Another transformer utilized in the well known forward converter also has no DC bias and no stored energy but falls short of the above described ac transformer, as it utilizes only one half of the core flux capability as illustrated in
Flyback Transformer Type
Unfortunately, in switching converters, the magnetic structure used in the flyback converter is also commonly called a transformer, even though it does not meet the fundamental feature of the transformer of not storing the energy. To the contrary, this type of magnetic structure actually stores the inductive energy in the in magnetizing inductance of the transformer during ON-time interval and then releases the stored inductive energy during the subsequent OFF-time interval. Therefore, the magnetic core must have an air-gap to store that energy and prevent the saturation of the core flux due to the DC-bias of the core, as illustrated in
Tapped-Inductor Type
We have already seen this tapped-inductor structure in the tapped-inductor buck converter. The tapped-inductor, is in-fact, just a variant of the flyback “transformer” as it also stores all the inductive energy in the magnetizing inductance during ON-time interval and releases it to the load during the OFF-time interval with the only difference being that it lacks the isolation feature since part of the winding is common to both primary and secondary windings. Thus, tapped-inductor could also be designated as a flyback “autotransformer”, to signify the lack of isolation feature.
Coupled-Inductor Magnetic Structure
In some switching converters, such as Ćuk converter (1), the separate inductors have identical AC voltage excitation, so that the inductors could be coupled on the common magnetic core (1) resulting in two switching converter variants: one with the separate inductors and another with coupled-inductors with either converters being operational but with coupled-inductors bringing additional performance benefits. Note, therefore, the key difference with tapped-inductor magnetic structure as used in switching converters. For example, the tapped-inductor buck converter of
In most current applications the coupled-inductor structure results in the DC storage of two separate inductors added together resulting in the need for a gapped core. However, it is also possible to find the coupled-inductor structures in which DC ampere turns excitations of the two inductors cancel after magnetic coupling resulting in no DC energy storage and hence in a true ac transformer-like structure with no air-gap needed for storage. Such a transformer despite the DC bias in each separate inductor could be described through coupled inductor equations modeling the ac transformer.
Hybrid Transformer
In the switching converters it is possible to have a two-winding magnetic structure such as the one in the converter of
This is a consequence of the fact that the converter of
Clearly, this combined inductive and capacitive energy storage and transfer ultimately result in the energy storage of the hybrid transformer of
An alternative way to calculate the net DC bias is to observe that the primary winding N1 is DC blocked by resonant capacitor Cr whose charge balance demands that the net DC current flowing into N1 winding is zero, hence no DC bias is generated from the primary N1 winding. Thus, all the DC bias is coming from the secondary N2 turn winding and the respective total current in that winding during the OFF-time interval.
Because the two winding structure operates partly as a tapped-inductor (for inductive current flow) and partly as a transformer (for capacitive discharge resonant current) this two winding structure is designated as a hybrid transformer.
Combined Capacitive and Inductive Storage and Transfer
The converter of
The energy in previous ON-time interval is during this OFF-time interval being released to the load through two different charge transfer paths as described below.
By the principle of linear superposition, the equivalent circuit model for discharge interval of
From
From
This results in the first basic relationship of the present invention, that the output current i0 is the sum of the resonant inductor current ir and the hybrid transformer secondary current iS, which are designated in
i0=iS2+iS (10)
Therefore, the load current is being supplied with the current during both parts of the switching interval, the ON-time interval and OFF-time interval. The conventional tapped-inductor buck converter supplies the load with the inductive energy storage and transfer only, since there is no capacitive energy storage and transfer. The present invention, on the other hand, supplies to the load an additional current based on the capacitive energy storage and transfer via hybrid transformer action. This results in a fundamentally much more effective power transfer based on combined inductive and capacitive energy storage and transfer working together and in synchronism during two switching subintervals. It is important that the resonant capacitor charge current is never wasted, as it is delivered to the load during the ON-time interval, and via hybrid transformer also discharged to the load. In the process, the energy stored on the resonant inductor is never lost but, just the opposite, completely delivered to the load, as shown in later section on experimental verification.
The load current during the OFF-time interval TOFF (
a) Inductive energy discharge through secondary winding of hybrid transformer.
b) Resonant discharge current of the resonant capacitor amplified by transformer turns ratio m and delivered to the load via hybrid transformer secondary. Note that this part was missing in the tapped-inductor buck converter.
c) Direct contribution of the resonant inductor current to the load. Note that this part is also missing in the tapped-inductor buck converter.
We now analyze a series of equivalent circuit models in
a) Steady-state DC voltage V, on the resonant capacitor Cr which will, in turn, lead to determination of the DC voltage gain M and
b) Provide explanation for a unique one-half cycle resonant current flow to the load.
From the circuit model in
∫VCrdt=Vr−2V=0 (11)
since the DC voltage across the resonant inductor voltage must be zero, as the resonant inductor cannot support any DC voltage across it and must be fully flux-balanced during this OFF-time interval. From (11) the summation of DC voltages around the loop in
C>>Cr (12)
Finally, the switches are replaced with ideal short circuits to result in the final simple series resonant circuit model of
Note that the series connection of the active switch S2 and current rectifier CR is left in the circuit model of
No Current Jump in the Hybrid Transformer
The placement of an external resonant inductor in series with the primary of the hybrid transformer in the present invention dictates that the primary inductor current must be continuous at the transition from the ON-time interval to the OFF-time interval, since the continuity of the external resonant inductor current dictates so. Note that this is in complete contrast to prior art tapped-inductor case in which operation of the tapped inductor caused the jump in the primary current waveform. The next experiment was designed to verify the elimination of the jump in the primary current of the hybrid transformer and the proof that the resonant inductor energy is not dissipated but instead delivered to the load.
Experimental Verification
The prototype to verify basic qualitative and quantitative operation of the present invention was built with a hybrid transformer turns ratio n=2 and m=1. The input voltage was 24V resulting in 5:1 voltage step-down at 50% duty ratio. Hence expected DC output voltage is 4.8V. The converter was also operated at 3 A DC load current for a 15 W output.
The resonant inductor Lr and resonant capacitor Cr were selected to operate with a resonant period of Tr=50 μsec or 20 kHz switching frequency for duty ratio D=0.5. Furthermore, the OFF-time interval is chosen to be fixed and equal to:
TOFF=0.5Tr, fr=1/Tr (13)
This was chosen in order to insure that the half of the resonant period coincides with the OFF-time interval. The variation of the ON-time interval or effectively duty ratio, while keeping the OFF-time interval constant as per (13), is now controlling the output DC voltage.
The measurements were first made when the duty ratio is adjusted to 50% or D=0.5. The experimental waveforms shown in
1) gate drive of the first switch S1.
2) current of the second switch iS2
3) secondary current iS of the hybrid transformer
4) output current i0, of the converter hybrid transformer.
The new hybrid-switching method can now be explained with the reference to
Finally, the half-sinusoidal resonant inductor current i0 is shown in
Note that the resonant capacitor Cr plays a dual role as the energy storage and energy transfer capacitor as in regular PWM square-wave converters, such as the Ćuk (1,2) and the SEPIC converters (2). However, here capacitor discharge interval is not liner but resonant. For example, during the ON-time interval the resonant capacitor Cr displays the characteristic linearly increasing ac ripple voltage as displayed in
Another characteristic of this hybrid-switching method not present in any other resonant methods is that despite the clear presence of the resonance, the usual dependence of the DC voltage gain M on resonant component values Lr and Cr as well as on the load current is completely absent and the conversion gain M is dependent on duty ratio D only. From the above it is obvious how in this new hybrid-switching conversion method both capacitive and inductive energy storage and transfers are taking place simultaneously in transferring power from the source to the load using both resonant current and square-wave current switching.
Note the marked difference with respect to the energy transfer in the conventional buck converter of
Evaluation of DC Voltage Gain
We now turn to evaluation of the DC voltage gain first. We assume a duty ratio control D of the main switch S1.
Flux Balance on Two Magnetic Components
First the flux balance on the resonant inductor Lr obtained previously for n=2 case can be now generalized for an arbitrary turns ratio n to:
∫VCrdt=Vr−nV=0 (14)
We then apply the second flux balance criteria, the flux balance on the winding N (equality of the shaded areas in
VgD−(n+1)VD=nV(1−D) (15)
M=D/(n+D) (16)
Note a remarkable result (16). Despite the presence of the resonance, owing to the hybrid-switching method described above, the DC voltage gain M is only a function of the duty ratio D and the hybrid transformer turns ratio n and is NOT a function of resonant component values nor the load current I. All other prior-art switching methods employing one or more resonant inductors resulted in the heavy dependence on the resonant component values as well as the DC load current. Therefore, the output voltage of the converter in
Up until now, the resonant converters were intrinsically tied to the control and regulation via changing switching frequency relative to the fixed resonant frequency (which spanned the entire switching cycle) so the conventional resonant converters were a-priori excluded from the regulation via PWM duty ratio control. The present invention actually confirms that PWM duty ratio control is not only possible but also advantageous in this new type of hybrid switching converters employing the resonant currents flowing only during a switching subinterval, such as OFF-time interval and not during the entire switching interval as in conventional resonant converters.
Resonant Circuit Analysis
Resonance Equations for OFF-time Interval
In
We now undertake to solve the pertinent resonance equations, which will describe analytically such time domain solutions. The derived analytical results could then be used to calculate the component values needed for optimum operation of the converter.
From the resonant circuit model of
Lrdir/dt=vr (17)
Crdvr/dt=−ir (18)
whose solutions are:
ir(t)=Im sin ωrt (19)
vr(t)=RNIm cos ωrt (20)
where RN is characteristic impedance, ωr is radial resonant frequency, fr resonant frequency and Tr resonant period given by:
RN=√Lr/Cr (21)
ωr=1/√LrCr (22)
Tr=1/fr=2π√LrCr (23)
Note the importance of the quantity Tr. From the equivalent circuit model in
Resonant Inductor Size
Note from the equivalent circuit model of
As seen from
Resonant Capacitor Size
Resonant capacitor size is also rather small and typically comparable to the size of the resonant inductor. This comes as a result of two facts:
The DC voltage gain M (15) can also be expressed in the following form:
M=D/n(1+D/n)≦D/n (24)
The conversion gain of Mi of a fully isolated transformer (not autotransformer) converter type would be expected to result in conversion gain Mi given by:
Mi=D/n (25)
Thus, the voltage gain M of the present invention with hybrid transformer and step-down ratio n results in higher step-down conversion ratio than could be expected of the isolated converter types, such as the conventional forward converter type for example, with conversion gain (25). This is clearly attributed to the presence of the capacitive energy transfer, hybrid resonant switching, and the resonant current contribution to the load as per (9).
The family of the DC voltage gains M with turn's ratio n as a parameter are displayed in the graphs of
From the comparison of two families of curves it is also clear that the present invention provides for the same duty ratios the significantly larger step-down conversion ratios than tapped-inductor buck. For example, for D=0.5 and n=2 the tapped-inductor conversion ratio is 3 while for present invention conversion ratio is 5. At duty ratio D=2/3 and for n=2, the present invention results in four times reduction of the input voltage compared to two times reduction of the tapped-inductor buck, thus a factor of two higher reduction at the same duty ratio and for same turns ratio n=2. Hence 12V input would be reduced to quite low 3V output voltage with present invention while it would result in 6V output with tapped-inductor buck converter. This is clearly attributed to the presence of the capacitive energy transfer and resonance via the hybrid-switching method. Comparison with the ordinary buck converter leads to even larger reduction factor of 8/3=2.67 so 12V would result in 8V output voltage in ordinary buck converter operated at 2/3-duty ratio. Note that 8/3 higher conversion ratios over the buck converter is achieved by addition of only a single turn to make a two winding hybrid transformer compared to a single turn inductor in buck converter. The DC-bias of the buck converter with single turn is actually higher than the DC bias of the hybrid transformer.
Note also that the operation at higher duty ratios is desirable as it leads directly to the reduction of the ac flux and magnetic size reduction as per graph in
It is now also instructive to compare the operation of the two converter types having the same DC voltage gain at same DC operating duty ratio D point but using the appropriate turns ratios for each case. For example, the present invention with n=2 will result in 5:1 step-down conversion ratio while the tapped-inductor buck converter at duty ratio D=0.5 would need to operate with n=4 as seen at the intersection of the two curves displayed in
Voltage Stresses of the Three Switches
From the derived DC currents in all branches one can also derive analytical expressions for the rms currents in various branches so that the conduction losses of the three switches could be calculated. What remains is to determine the voltage stresses of all three switches so that the proper rated switching devices could be selected. From the circuit diagram for OFF-time interval in
S1: VS1=Vg−V (26)
S2: VS2=Vg−V (27)
CR: VCR=(Vg−V)/n (28)
Both active switches have lower voltage stresses than the comparable buck converter. However, note in particular large voltage stress reduction for the rectifier switch CR that conducts most of the power for the large step-down. For example, for 12V to 1V conversion and n=4, the blocking voltage of the rectifier switch is VCR=11/4 V=2.75V. This is to be compared with the blocking voltage of 12V for comparable buck converter or a factor of 4.4 reductions in voltage stress of the switch, which processes by far the most of the power to the load for high step-down conversion and is critical for overall efficiency.
Other Switch Implementations
Hybrid transformer can be replaced by a transformer with two separate windings to result in two extensions illustrated in
Large Voltage Step-down Operation
We not turn to describe a very unique performance of the present invention when the large voltage-step-down conversion such as 12:1 or 24:1 is desired, such as for 12V to 1V and 12V to 0.5V conversion. It will be described bellow and experimentally confirmed that the larger the step-down voltage conversion the better performance is of the present invention.
The same experimental prototype used previously is now employed to investigate the performance at low duty ratios, or for large voltage step-downs. The following were the chosen operating conditions: Vg=24V, TOFF=20 μsec is constant and hybrid transformers with turns ratios n=2 and m=1 was used again. The duty ratio is then adjusted to 0.15 to obtain output voltage of V=1V for an effective 24 to 1 voltage step-down. The experimental waveforms where then recorded for the load current changes from 8 A to 1 A in decrements of 1 A, to observe the behaviors over the wide load current change. These measurements are illustrated in
Note a rather remarkable qualitative difference in the wave-shape of the output current as illustrated in
i0(t)=I+I sin(ωrt) (29)
where I is the average DC load current and I is also a peak of the full sinusoidal resonant output current waveform. The measurement taken for reduced load currents of 6 A, 5 A, 4 A, 3 A, 2 A and 1 A shown in
This is very important result, confirming that for large and sudden changes in the load current, the converter does not need to go through many cycles to settle on a new steady-state, as the same steady state is reached every cycle. With DC load current reduced the peak magnitude of the output sinusoidal current is likewise adjusted to the new value in a single cycle.
This unique property can also be fully taken advantage of by paralleling the two identical converters as described in next section.
Two-Phase Extension
The common method to reduce the output voltage ripple in the buck converter is to use a multi-phase buck converter with several buck converters (typically four phases) which are operated in parallel but phase shifted by a quarter of period to result in reduced ripple current and reduced output ripple voltage.
The present invention, however, makes possible the near ideal cancellation of the output ripple current with just two phases as illustrated in the two-phase converter implementation shown in
i01(t)=I+I sin(ωrt) (30)
i02(t)=I−I sin(ωrt) (31)
Therefore, the total output current i0 of the two modules is then:
i0(t)=i01(t)+i01(t)=2I (32)
since the DC currents of each phase-module add together while their time varying sinusoidal currents ideally cancel.
The actual current waveform of each of the two-phase modules are shown in
Another example is illustrated for the transformer turns ratio n=4 and for duty ratio D=0.25, with output current waveforms in
Reduction of Turn-OFF Losses of the Main Input Switch
As explained in introduction with reference to
Other Embodiments
An alternative converter topology could be obtained by connecting the second switch S2 of the basic converter in
Those skilled in the art could also find other beneficial placements of the resonant inductor, which would also employ above combined inductive and capacitive energy storage and transfer which is the main feature of the present invention.
Protection of the Load
The converter extension of
In the buck converter, shorting of the main switch will cause that the input 12V voltage will be directly applied to low 1V output and result in damage to the expensive loads such as microprocessors.
This cannot happen in this extension of the present invention, since shorting of the input switch will not cause the damage to the load. After a small transient spike the output voltage will be reduced to near zero output voltage as the resonant capacitor and output capacitor serve as an effective capacitive divider. Since the output capacitor value is many times (at least ten times) higher in value than the resonant capacitor, the output voltage will be limited to 1/10 of input voltage or 1.2V.
Equally important, a single-point failure of the resonant capacitor (its shorting) will not result in the catastrophic destruction either as the present invention of
Modeling and Analysis
Equivalent circuit model analysis of converter in
Evaluation of DC Voltage Gain
We now turn to evaluation of the DC voltage gain for the converter topology in
Flux Balance on Two Magnetic Components
First the flux balance on the resonant inductor Lr can be now shown for an arbitrary turns ratio n to:
∫vrdt=Vr−(n−1)V=0 (33)
We than apply the second flux balance criteria, the flux balance on the hybrid transformer (equality of the shaded areas in
VgD−nVD=nV(1−D) (34)
M=D/n (35)
The family of the DC conversion gains as a function of duty ratio for different turns ratios n is shown in graphs in
Resonant Circuit Analysis
The same resonant circuit model is obtained for this case (
Voltage Stresses of the Three Switches
Let us now evaluate the voltage stresses in the converter of
S1: VS1=Vg (36)
S2: VS2=Vg (37)
S3: VS3=Vg/n (38)
Both active switches have voltage stresses equal to the input voltage as in a buck converter. However, note in particular large voltage stress reduction for the synchronous rectifier switch S3 that conducts most of the power for the large step-down. For example, for 12V to 1V conversion and n=4, the blocking voltage of the synchronous rectifier switch is VS3=12/4 V=3V. This is to be compared with the blocking voltage of 12V for comparable buck converter or a factor of 4 times reduction in voltage stress of the switch.
Voltage Regulation via Duty Ratio Control
The converters of present invention in
All other converters based on resonance have a DC voltage gain not only dependent on the resonant component values, but also of not being suitable for the duty ratio control. In these resonant converters the output voltage is controlled in a resonant circuit fashion by changing the ratio of switching frequency to the resonant frequency, which is not capable to regulate the output voltage over even the modest change in DC load currents due to high dependence on the resonance Q factor. However, the present invention employs the very simple duty ratio control of the output voltage and is independent on the load current and resonant component values.
The optimal control method introduced so far is constant OFF-time, variable ON-time control which ultimately means also a variable switching frequency. However, for the practical step-down conversion ratios, such as 10:1 and higher as used in experimental examples, the change of the ON-time interval is relatively small from the nominal value, so that even though a variable switching frequency is employed, the change of switching frequency is also small. However, if so desired, a constant switching frequency and variable duty ratio could be employed at the minor sacrifice in efficiency due to presence of zero coasting intervals and somewhat increased values of rms currents.
Conclusion
A three-switch step-down converter with a resonant inductor, a resonant capacitor and a hybrid transformer provides efficiency, size, cost and other performance advantages over the conventional buck converter and tapped-inductor buck converter.
The resonant inductor is connected to the primary of the isolation transformer. This insures the continuity of the primary and secondary currents of the hybrid transformer and therefore eliminates the current jump at the transition from ON-time interval to OFF-time interval present in conventional tapped-inductor buck converter thereby eliminating associated losses.
Despite the presence of the resonant inductor current during the OFF-time interval, the output voltage is neither dependent on resonant component values nor on the load current as in conventional resonant converters but dependent on duty ratio D and hybrid transformer turn ratio n. Hence a simple regulation of output voltage is achieved using duty ratio control only.
The dual inductive and capacitive energy storage and transfer together with lower voltage stresses on the switches results in increased efficiency and reduced size and cost compared to buck converter and tapped-inductor buck converters. Much reduced voltage stress on the synchronous rectifier switch also results in proportionally reduced size of the silicon needed for the switch implementation. Hence for 12V to 1V converter total cost of the silicon used for switches could be reduced fourfold.
The present invention also introduces a new hybrid switching method, which implements for the first time a use of odd number of switches, such as three in this case, which is strictly excluded from use in conventional Square-wave, Resonant and Quasi-resonant switching converters, which all require an even number of switches (2, 4, 6 etc.), operating as complementary pairs.
Finally, two identical converters can be used in two-phase extension In a such a way to reduce the output ripple current by a tenfold.
Number | Name | Date | Kind |
---|---|---|---|
6486642 | Qian | Nov 2002 | B1 |
6989997 | Xu et al. | Jan 2006 | B2 |
7321224 | Iwamoto et al. | Jan 2008 | B2 |
7915874 | Cuk | Mar 2011 | B1 |
8207717 | Uruno et al. | Jun 2012 | B2 |
20010024373 | Cuk | Sep 2001 | A1 |
20020118000 | Xu et al. | Aug 2002 | A1 |
Entry |
---|
Slobodan Cuk, “Modelling, Analysis and Design of Switching Converters”, PhD thesis, Nov. 1976, California Institute of Technology, Pasadena, California, USA;www.caltech.edu. |
Vatche Vorperian, “Resonant Converters”, PhD thesis, California Institute of Technology, Pasadena, California, USA; www.caltech.edu. |
Steve Freeland, “A Unified Analysis of Converters with Resonant Switches; II Input Current Shaping ” PhD thesis, Oct. 20, 1987,Caltech, Pasadena, California; www.caltech.edu. |
Slobodan Cuk, “Modeling, Analysis and Design of Switching Converters”, PhD thesis, Nov. 1976, California Institute of Technology, Pasadena, California, USA; www.caltech.edu. |
Vatche Vorperian, “Resonant Converters”, PhD thesis, May 1984, California Institute of Technology, Pasadena, California, USA; www.caltech.edu. |
Steve Freelan, “A Unified Analysis of Converters with Resonant Switches; Input Current Shaping for Single-Phase Ac-Dc Power Converters”, PhD thesis, Oct. 20, 1987, Caltech, Pasadena, California; www.caltech.edu. |
Dragan Maksimovic, “Synthesis of PWM and Quasi-Resonant DC-to-DC Power Converters”, PhD thesis, Jan. 12, 1989, Caltech, Pasadena, California; www.caltech.edu. |
Number | Date | Country | |
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20120249102 A1 | Oct 2012 | US |