This application claims priority under 35 U.S.C. 119(b) to co-pending and commonly owned Indian Patent Application No. 201841031723 entitled “IN-RUSH CURRENT PROTECTION FOR LINEAR REGULATORS” filed on Aug. 24, 2018, the entirety of which is incorporated by reference herein.
The present embodiments relate generally to linear regulators, and specifically to in-rush current protection for linear regulators.
A linear regulator may be configured to receive an input voltage and produce a relatively stable output voltage that is resistant to ripple or other small voltage variations in the input voltage. For example, the linear regulator may maintain the output voltage by varying its effective resistance (e.g., corresponding to the resistance of a pass element) in response to changes in an amount of current flowing through a load element coupled to its output. Accordingly, the output voltage of the linear regulator should be lower than the input voltage by at least a saturation or overdrive voltage to maintain the pass element in saturation. However, in some instances, the pass element may not remain in saturation, which may push the output voltage closer to the input voltage. The minimum voltage difference between the output voltage and the input voltage is referred to as “dropout voltage.” A low-dropout (LDO) regulator is a linear regulator with a very small dropout voltage (e.g., <2V). LDO regulators are often used for producing supply voltages within low-voltage microprocessors or other integrated circuit (IC) devices.
LDO regulators with very low dropout voltages, high load currents, and high external capacitances tend to draw large amounts of in-rush current during power-up or initialization. In some instances, a sufficiently large surge of in-rush current may collapse or burn the wiring of the power supply grid on which the LDO regulator operates, resulting in permanent damage to the device. Thus, it may be desirable to control the flow of in-rush current when powering on the LDO regulator.
This Summary is provided to introduce in a simplified form a selection of concepts that are further described below in the Detailed Description. This Summary is not intended to identify key features or essential features of the claims subject matter, nor is it intended to limit the scope of the claimed subject matter.
A linear voltage regulator including a pass element, an error amplifier, and in-rush current protection (ICP) circuitry is disclosed. The pass element is configured to produce an output voltage across a capacitor based on a received input voltage. The error amplifier is configured to output a control voltage based at least in part on the output voltage and a reference voltage. The control voltage is used to control a flow of output current through the pass element. The ICP circuitry is coupled to the pass element and the error amplifier and configured to maintain the output current below a threshold level while charging the capacitor from a discharged state to the output voltage. For example, the ICP circuitry may be configured to charge the capacitor when powering on the linear voltage regulator.
In some implementations, the ICP circuitry may include a pre-charge circuit configured to charge the capacitor to a first threshold voltage while preventing the pass element from providing the output current to the capacitor. For example, the pre-charge circuit may include switching circuitry configured to selectively suppress the reference voltage at an input of the error amplifier, and a transistor coupled in parallel with the pass element and configured to provide a pre-charge current to the capacitor when the reference voltage is suppressed. In some aspects, a low-pass filter may be coupled to the input of the error amplifier to filter the reference voltage. The gate of the transistor may be switchably coupled to the output of the error amplifier based at least in part on the charge on the capacitor. For example, the gate may be coupled to the output of the error amplifier when the capacitor is charged to the first threshold voltage. More specifically, the transistor may be configured to provide at least a portion of the output current when the gate is coupled to the output of the error amplifier.
In some other implementations, the ICP circuitry may include a voltage clamp configured to throttle the output current through the pass element when the charge on the capacitor is below a second threshold voltage. For example, the voltage clamp may include one or more transistors coupled between the output of the error amplifier and a first voltage potential. In some aspects, the one or more transistors may be configured to pull the control voltage toward the first voltage potential when the charge on the capacitor is below the second threshold voltage. For example, an impedance across the one or more transistors may be lower than an output impedance of the error amplifier. In some aspects, the first voltage potential may correspond to the input voltage. Moreover, at least one of the one or more transistors may be turned off when the charge on the capacitor reaches the second threshold voltage.
The present embodiments are illustrated by way of example and are not intended to be limited by the figures of the accompanying drawings.
In the following description, numerous specific details are set forth such as examples of specific components, circuits, and processes to provide a thorough understanding of the present disclosure. The term “coupled” as used herein means connected directly to or connected through one or more intervening components or circuits. Also, in the following description and for purposes of explanation, specific nomenclature is set forth to provide a thorough understanding of the aspects of the disclosure. However, it will be apparent to one skilled in the art that these specific details may not be required to practice the example embodiments. In other instances, well-known circuits and devices are shown in block diagram form to avoid obscuring the present disclosure. Some portions of the detailed descriptions which follow are presented in terms of procedures, logic blocks, processing and other symbolic representations of operations on data bits within a computer memory. The interconnection between circuit elements or software blocks may be shown as buses or as single signal lines. Each of the buses may alternatively be a single signal line, and each of the single signal lines may alternatively be buses, and a single line or bus may represent any one or more of a myriad of physical or logical mechanisms for communication between components.
Unless specifically stated otherwise as apparent from the following discussions, it is appreciated that throughout the present application, discussions utilizing the terms such as “accessing,” “receiving,” “sending,” “using,” “selecting,” “determining,” “normalizing,” “multiplying,” “averaging,” “monitoring,” “comparing,” “applying,” “updating,” “measuring,” “deriving” or the like, refer to the actions and processes of a computer system, or similar electronic computing device, that manipulates and transforms data represented as physical (electronic) quantities within the computer system's registers and memories into other data similarly represented as physical quantities within the computer system memories or registers or other such information storage, transmission or display devices.
The techniques described herein may be implemented in hardware, software, firmware, or any combination thereof, unless specifically described as being implemented in a specific manner. Any features described as modules or components may also be implemented together in an integrated logic device or separately as discrete but interoperable logic devices. If implemented in software, the techniques may be realized at least in part by a non-transitory computer-readable storage medium comprising instructions that, when executed, performs one or more of the methods described above. The non-transitory computer-readable storage medium may form part of a computer program product, which may include packaging materials.
The non-transitory processor-readable storage medium may comprise random access memory (RAM) such as synchronous dynamic random access memory (SDRAM), read only memory (ROM), non-volatile random access memory (NVRAM), electrically erasable programmable read-only memory (EEPROM), FLASH memory, other known storage media, and the like. The techniques additionally, or alternatively, may be realized at least in part by a processor-readable communication medium that carries or communicates code in the form of instructions or data structures and that can be accessed, read, and/or executed by a computer or other processor.
The various illustrative logical blocks, modules, circuits and instructions described in connection with the embodiments disclosed herein may be executed by one or more processors. The term “processor,” as used herein may refer to any general purpose processor, conventional processor, controller, microcontroller, and/or state machine capable of executing scripts or instructions of one or more software programs stored in memory.
The LDO regulator 100 may include a pass element 110, an error amplifier 120, and in-rush current protection (ICP) circuitry 130. The pass element 110 may have a variable resistance (or capacitance) that is adjustable via a control voltage Vctrl. More specifically, the control voltage Vctrl may control a voltage drop across the pass element 110. For example, in some implementations, the pass element 110 may comprise one or more metal-oxide semiconductor field-effect transistors (MOSFETs), such as PMOS transistors and/or NMOS transistors. In some other implementations, the pass element 110 may comprise one or more bipolar junction transistors (BJTs), such as NPN transistors and/or PNP transistors.
The error amplifier 120 compares the output voltage Vout (or a fraction thereof) with a reference voltage Vref and adjusts the control voltage Vctrl based, at least in part, on a difference between the voltages. During operation, variations in the input voltage Vin and/or changes in resistance of the external load may cause changes in the output current Iout. Such changes in the output current Iout may, in turn, affect the output voltage Vout. The error amplifier 120 adjusts the voltage drop across the pass element 110 (e.g., by changing a resistance of the pass element 110), in response to changes in the output current Iout, to maintain the output voltage Vout at a relatively constant voltage. The error amplifier 120 may have a relatively high output impedance. For example, the error amplifier 120 may be implemented by a folded cascode error amplifier, a telescopic error amplifier, or any suitable two-stage error amplifier.
In the example of
In some embodiments, the ICP circuitry 130 may control the flow of in-rush current when powering on the LDO regulator 100. More specifically, the ICP circuitry 130 may throttle the output current Iout, during device power-up, by adjusting the resistance of the pass element 110. In some aspects, the ICP circuitry 130 may pre-charge the capacitor Cout to a predetermined level (e.g., by controlling the reference voltage Vref provided to the error amplifier 120 and the control voltage Vctrl provided to the pass element 110) to reduce the draw of output current Vout by the capacitor Cout. In some other aspects, the ICP circuitry 130 may maintain a resistance of the pass element 110 at a relatively high level (e.g., by clamping the control voltage Vctrl to a predetermined voltage) to limit the flow of output current through the pass element 110. After the output current Iout reaches a substantially steady-state level, the ICP circuitry 130 may be deactivated or bypassed, thus allowing the LDO regulator 100 to resume normal operation.
During normal operation, the ballast transistor MB may pass an output current Iout from the voltage supply rail VDD to the output of the LDO regulator 200. In the example of
The differential amplifier 210 may adjust the resistance (or impedance) of the ballast transistor MB to maintain the output voltage Vout at the desired voltage level. For example, the inverting (−) input of the differential amplifier 210 may be coupled to receive a reference voltage Vref and the non-inverting (+) input of the differential amplifier 210 may be coupled to receive a feedback voltage VFB that is representative of the output voltage Vout. In the example of
The gate voltage VG controls the resistance of the ballast transistor MB. For example, as the gate voltage VG decreases, the resistance across the transistor MB (e.g., from source to drain) also decreases. Similarly, as the gate voltage VG increases, the resistance across the transistor MB also increases. Accordingly, the differential amplifier 210 may adjust the gate voltage VG, based on feedback from the output of the LDO regulator 200 (e.g., in the form of the feedback voltage VFB), to maintain the output voltage Vout at the desired voltage level. More specifically, the differential amplifier 210 may adjust the gate voltage VG, in response to variations in the input voltage VDD and/or a load impedance (not shown for simplicity) coupled to the output of the LDO regulator 200, to maintain a constant and steady output voltage Vout.
For example, the differential amplifier 210 may increase the gate voltage VG when the feedback voltage VFB rises (e.g., corresponding to an increase in the output current Iout) to increase the resistance of the ballast transistor MB, thereby reducing the flow of current through the transistor MB. Similarly, the differential amplifier 210 may decrease the gate voltage VG when the feedback voltage VFB drops (e.g., corresponding to a decrease in the output current Iout) to decrease the resistance of the ballast transistor MB, thereby increasing the flow of current through the transistor MB.
As described above, a capacitor Cout may be coupled to the output of the LDO regulator 200 to filter and/or maintain the output voltage Vout. For example, when charged, the capacitor Cout may resist large instantaneous changes in the output current Iout. However, the capacitor Cout may be initially discharged (e.g., to 0V) prior to powering on the LDO regulator 200. Thus, during device power-up, the capacitor Cout may act as a shunt, drawing a substantial amount of in-rush current from the input voltage VDD. As described above, a sufficiently large surge of in-rush current may collapse the power supply grid on which the LDO regulator 200 operates.
In some embodiments, the pre-charge circuit 220 may pre-charge the capacitor Cout to a predetermined level prior to initiating normal operation of the LDO regulator 200. Specifically, the pre-charge circuit 220 may pre-charge the capacitor Cout to a threshold voltage potential that is just below the desired output voltage Vout for the LDO regulator 200 in order to reduce the voltage difference across the ballast transistor MB, and thus reduce the surge of in-rush current during device power-up. For example, the pre-charge circuit 220 may suppress or ground the reference voltage Vref provided to the inverting input of the differential amplifier 210 to prevent the output of the differential amplifier 210 from turning on the ballast transistor MB. While the ballast transistor MB is turned off (and is not affected by the feedback voltage VFB), the pre-charge circuit 220 may provide a pre-charge current IPC to the capacitor Cout until the voltage across the capacitor reaches the threshold voltage potential. In some aspects, the pre-charge current IPC may be supplied by the voltage supply rail VDD.
Still further, in some embodiments, the voltage clamp 230 may throttle the flow of current through the ballast transistor MB while charging the capacitor Cout to the desired output voltage Vout. Specifically, the voltage clamp 230 may maintain the resistance of the ballast transistor MB at a relatively high level until the capacitor Cout is charged to the desired output voltage Vout. For example, once the capacitor Cout has been pre-charged to the threshold voltage potential, the pre-charge circuit 220 may suppress the pre-charge current IPC and provide the normal reference voltage Vref to the inverting input of the differential amplifier 210. This allows the differential amplifier 210 to begin adjusting the gate voltage VG based on the feedback voltage VFB. However, in some embodiments, the voltage clamp 230 may ensure that the gate voltage VG does not drop below a predetermined voltage level. For example, the predetermined voltage level may prevent the ballast transistor MB from fully turning on (e.g., even when the voltages Vref and VFB are substantially equal), thus throttling the flow of current through the ballast transistor. In some aspects, the voltage clamp 230 may pull up the gate voltage VG using the voltage supply rail VDD.
When the capacitor Cout has been charged to the desired output voltage Vout, the voltage clamp 230 may be decoupled from the gate of the ballast transistor MB. Thereafter, the LDO regulator 200 may resume normal operation (e.g., by adjusting the resistance of the ballast transistor MB in response to changes in the feedback voltage VFB). In other words, control of the ballast transistor MB is returned to the differential amplifier 210. By pre-charging the capacitor Cout to a threshold voltage potential, the pre-charge circuit 220 may prevent the LDO regulator 200 from drawing a large surge of in-rush current during device power-up. Moreover, by clamping the gate voltage VG to a predetermined voltage level, the voltage clamp 230 may further limit the flow of output current Iout while charging the capacitor Cout to the desired output voltage Vout.
In the example of
and Vt1 represents a threshold voltage potential). Thus, when the feedback voltage VFB is less than the comparison voltage (e.g., VFB<Vref−Vt1), the comparator 312 outputs PRE_CH in a logic-high state (and
In some embodiments, the LDO pre-charge controller 310 may enable hysteresis once the feedback voltage VFB reaches or exceeds the comparison voltage. For example, when the feedback voltage VFB is greater than or equal to the comparison voltage (e.g., VFB≥Vref−Vt1), the second input of the comparator 312 may switch to the hysteresis voltage (e.g., Vref−Vh1, where Vh1>Vt1). The hysteresis voltage may be configured to prevent the output of the comparator 312 from oscillating in response to slight voltage fluctuations in the feedback voltage VFB when the feedback voltage VFB is substantially equal to the comparison voltage (e.g., VFB=Vref−Vt1).
In the example of
In the example of
and Vt2 represents a threshold voltage potential). Thus, when the reference voltage Vref is less than or equal to the comparison voltage (e.g., Vref≤VFB+Vt2), the comparator 332 outputs V_CLAMP in a logic-high state (e.g., after some delay). However, when the reference voltage Vref is greater than the comparison voltage (e.g., Vref>VFB+Vt2), the comparator 332 may output V_CLAMP in a logic-low state (e.g., after some delay).
In some embodiments, the LDO voltage clamp controller 330 may enable hysteresis once the reference voltage Vref reaches or exceeds the comparison voltage. For example, when the reference voltage Vref is less than or equal to the comparison voltage (e.g., Vref≤VFB+Vt2), the second input of the comparator 332 may switch to the hysteresis voltage (e.g., Vref−Vh2, where Vh2>0). The hysteresis voltage may be configured to prevent the output of the comparator 332 from oscillating in response to slight voltage fluctuations in the feedback voltage VFB when the comparison voltage is substantially equal to the reference voltage Vref (e.g., Vref=VFB+Vt2).
In some embodiments, the LDO regulator 400 may include an LDO enable transistor MEN which may be used to turn the LDO regulator 400 on and/or off. In the example of
During normal operation, the ballast transistors MB1 and MB2 may pass an output current Iout from the voltage supply rail VDD to the output of the LDO regulator 400. In the example of
The differential amplifier 410 may adjust the resistance (or impedance) of the ballast transistors MB1 and MB2 to maintain the output voltage Vout at the desired voltage level. In some embodiments, the inverting (−) input of the differential amplifier 410 is switchably coupled to receive a reference voltage Vref and the non-inverting (+) input of the differential amplifier 410 is coupled to receive a feedback voltage VFB that is representative of the output voltage Vout. In the example of
As described above, the gate voltage VG controls the resistance of the ballast transistors MB1 and MB2. Furthermore, the differential amplifier 410 may adjust the gate voltage VG, based on feedback from the output of the LDO regulator 400 (e.g., in the form of the feedback voltage VFB), to maintain the output voltage Vout at the desired voltage level. More specifically, the differential amplifier 410 may adjust the gate voltage VG, in response to variations in the input voltage VDD and/or a load impedance (not shown for simplicity) coupled to the output of the LDO regulator 400, to maintain a constant and steady output voltage Vout. For example, the differential amplifier 410 may increase the gate voltage VG when the feedback voltage VFB rises to increase the resistance of the ballast transistors MB1 and MB2, thereby reducing the flow of output current Iout. Similarly, the differential amplifier 410 may decrease the gate voltage VG when the feedback voltage VFB drops to decrease the resistance of the ballast transistors MB1 and MB2, thereby increasing the flow of output current Iout.
A capacitor Cout may be coupled to the output of the LDO regulator 400 to filter and/or maintain the output voltage Vout. For example, when charged, the capacitor Cout may resist large instantaneous changes in the output current Iout. However, the capacitor Cout may be initially discharged (e.g., to 0V) prior to powering on the LDO regulator 400. Thus, during device power-up, the capacitor Cout may act as a shunt, drawing a substantial amount of in-rush current from the input voltage VDD. As described above, a sufficiently large surge of in-rush current may collapse the power supply grid on which the LDO regulator 400 operates.
In some embodiments, the LDO regulator 400 may include pre-charge circuitry to pre-charge the capacitor Cout to a predetermined level prior to initiating normal operation of the LDP regulator 400. Specifically, the pre-charge circuitry may pre-charge the capacitor Cout to a threshold voltage potential that is just below the desired output voltage Vout for the LDO regulator 400 in order to reduce the voltage difference across the ballast transistor MB, and thus reduce the surge of in-rush current during device power-up. In the example of
When PRE_CH is in a logic-high state, PRE_EN is in a logic-low state and
In some embodiments, the LDO regulator 400 may further include voltage clamping circuitry to throttle the flow of current through the ballast transistors MB1 and MB2 while charging the capacitor Cout to the desired output voltage Vout. Specifically, the voltage clamping circuitry may maintain the resistance of the ballast transistors MB1 and MB2 at a relatively high level until the capacitor Cout is charged to the desired output voltage Vout. In the example of
As described above with respect to
When V_CLAMP is in a logic-low state, the transistor MC_EN is turned on to couple the voltage clamp transistor MC to the gates of the ballast transistors MB1 and MB2. More specifically, as the voltages at the inputs of the differential amplifier 410 approach one another, the voltage at the output of the differential amplifier 410 may drop, thus pulling down the gate voltage VG and turning on the ballast transistors MB1 and MB2. However, the voltage clamp transistors MC and MC_EN may ensure that the gate voltage VG does not drop too far below a threshold voltage level, for example, to prevent ballast transistors MB1 and MB2 from fully turning on. In some embodiments, the resistance across the voltage clamp transistors MC and MC_EN may be lower than the output impedance of the differential amplifier 410. Thus, the voltage clamp transistors MC and MC_EN may hold the gate voltage VG at the threshold voltage level (e.g., VDD−(VGS of MC)), preventing the voltage at the output of the differential amplifier 410 from dropping. Accordingly, the voltage clamp transistors MC and MC_EN may throttle the flow of output current Iout through the ballast transistors MB1 and MB2 while the differential amplifier 410 reaches a steady state and the capacitor Cout is charged to the desired output voltage Vout.
As described above with respect to
When V_CLAMP is in a logic-high state, the transistor MC_EN is turned off and the voltage clamp transistor MC is decoupled from the gates of the ballast transistors MB1 and MB2. As a result, control of the gate voltage VG is returned to the differential amplifier 410 (e.g., the voltage clamp transistors MC and MC_EN no longer have control of the gate voltage VG). Once control of the gate voltage VG has been returned to the differential amplifier 410, the LDO regulator 400 may resume normal operation (e.g., as described above).
It is noted that, while the voltage clamp transistors MC and MC_EN may throttle the flow of output current Iout via the ballast transistors MB1 and MB2, the output current Iout may jump when the voltage clamp transistors MC and MC_EN release control of the gate voltage VG back to the differential amplifier 410 (e.g., when V_CLAMP is asserted to a logic-high state). In some embodiments, to prevent sudden jumps in the output current Iout, control of the gate voltage VG may be gradually returned to the differential amplifier 410 via multiple successive stages of voltage clamp transistors. For example, each successive stage of voltage clamp transistors may release its control over the gate voltage VG in sequence (e.g., at different times), until the output current Iout is locked to the output of the differential amplifier 410.
In the example of
and Vt2 represents a threshold voltage potential). Thus, when the reference voltage Vref is greater than or equal to the comparison voltage (e.g., Vref≥VFB+Vt2), the comparator 532 outputs V_CLAMP1 and V_CLAMP2 in a logic-low state. However, when the reference voltage Vref is less than the comparison voltage (e.g., Vref<VFB+Vt2), the comparator 532 may output V_CLAMP1 in a logic-high state after some delay and may output V_CLAMP2 in a logic-high state after some further delay. Accordingly, the LDO voltage clamp controller 530 may stagger the transitions of V_CLAMP1 and V_CLAMP2.
In some embodiments, the LDO voltage clamp controller 530 may enable hysteresis once the reference voltage Vref reaches or exceeds the comparison voltage. For example, when the reference voltage Vref is greater than or equal to the comparison voltage (e.g., Vref≥VFB+Vt2), the second input of the comparator 532 may switch to the hysteresis voltage (e.g., Vref−Vh2, where Vh2>0). The hysteresis voltage may be configured to prevent the output of the comparator 532 from oscillating in response to slight voltage fluctuations in the feedback voltage VFB when the comparison voltage is substantially equal to the reference voltage Vref (e.g., Vref=VFB+Vt2).
In some embodiments, the LDO regulator 500 may include an LDO enable transistor MEN which may be used to turn the LDO regulator 500 on and/or off. In the example of
During normal operation, the ballast transistors MB1-MB3 may pass an output current Iout from the voltage supply rail VDD to the output of the LDO regulator 500. In the example of
The differential amplifier 510 may adjust the resistance (or impedance) of the ballast transistors MB1-MB3 to maintain the output voltage Vout at the desired voltage level. In some embodiments, the inverting (−) input of the differential amplifier 510 is switchably coupled to receive a reference voltage Vref and the non-inverting (+) input of the differential amplifier 510 is coupled to receive a feedback voltage VFB that is representative of the output voltage Vout. In the example of
As described above, the gate voltage VG controls the resistance of the ballast transistors MB1-MB3. Furthermore, the differential amplifier 510 may adjust the gate voltage VG, based on feedback from the output of the LDO regulator 500 (e.g., in the form of the feedback voltage VFB), to maintain the output voltage Vout at the desired voltage level. More specifically, the differential amplifier 510 may adjust the gate voltage VG, in response to variations in the input voltage VDD and/or a load impedance (not shown for simplicity) coupled to the output of the LDO regulator 500, to maintain a constant and steady output voltage Vout. For example, the differential amplifier 510 may increase the gate voltage VG when the feedback voltage VFB rises to increase the resistance of the ballast transistors MB1-MB3, thereby reducing the flow of output current Iout. Similarly, the differential amplifier 510 may decrease the gate voltage VG when the feedback voltage VFB drops to decrease the resistance of the ballast transistors MB1MB3, thereby increasing the flow of output current Iout.
A capacitor Cout may be coupled to the output of the LDO regulator 500 to filter and/or maintain the output voltage Vout. For example, when charged, the capacitor Cout may resist large instantaneous changes in the output current Iout. However, the capacitor Cout may be initially discharged (e.g., to 0V) prior to powering on the LDO regulator 500. Thus, during device power-up, the capacitor Cout may act as a shunt, drawing a substantial amount of in-rush current from the input voltage VDD. As described above, a sufficiently large surge of in-rush current may collapse the power supply grid on which the LDO regulator 500 operates.
In some embodiments, the LDO regulator 500 may include pre-charge circuitry to pre-charge the capacitor Cout to a predetermined level prior to initiating normal operation of the LDO regulator 500. Specifically, the pre-charge circuitry may pre-charge the capacitor Cout to a threshold voltage potential that is just below the desired output voltage Vout for the LDO regulator 500 in order to reduce the voltage difference across the ballast transistor MB, and thus reduce the surge of in-rush current during device power-up. In the example of
When PRE_CH is in a logic-high state, PRE_EN is in a logic-low state and
In some embodiments, the LDO regulator 500 may further include voltage clamping circuitry to throttle the flow of current through the ballast transistors MB1-MB3 while charging the capacitor Cout to the desired output voltage Vout. Specifically, the voltage clamping circuitry may maintain the resistance of the ballast transistors MB1-MB3 at a relatively high level until the capacitor Cout is charged to the desired output voltage Vout. In the example of
As described above with respect to
When V_CLAMP1 and V_CLAMP2 are in a logic-low state, the transistors MC1_EN and MC2_EN are turned on to couple respective voltage clamp transistors MC1 and MC2 to the gates of the ballast transistors MB1-MB3. More specifically, as the voltages at the inputs of the differential amplifier 510 approach one another, the voltage at the output of the differential amplifier 510 may drop, thus pulling down the gate voltage VG and turning on the ballast transistors MB1-MB3. However, the voltage clamp transistors MC1/MC1_EN and MC2/MC2_EN may ensure that the gate voltage VG does not drop too far below a threshold voltage level, for example, to prevent ballast transistors MB1-MB3 from fully turning on. In some embodiments, the resistance across the voltage clamp transistors MC1/MC1_EN and MC2/MC2_EN may be lower than the output impedance of the differential amplifier 510. Thus, the voltage clamp transistors MC1/MC1_EN and MC2/MC2_EN may hold the gate voltage VG at the threshold voltage level (e.g., VDD−(VGS of MC1 or MC2)), preventing the voltage at the output of the differential amplifier 510 from dropping. Accordingly, the voltage clamp transistors MC1/MC1_EN and MC2/MC2_EN may throttle the flow of output current Iout through the ballast transistors MB1MB3 while the differential amplifier 510 reaches a steady state and the capacitor Cout is charged to the desired output voltage Vout.
As described above with respect to
More specifically, when the voltage across the capacitor Cout reaches the second threshold voltage potential, V_CLAMP1 first transitions to the logic-high state and, after some delay, V_CLAMP2 subsequently transitions to the logic-high state. When V_CLAMP1 is in a logic-high state, the transistor MC1_EN is turned off and the voltage clamp transistor MC1 is decoupled from the gates of the ballast transistors MB1-MB3. This may reduce some amount of pull on the gate voltage VG that was previously exerted by the first transistor stage MC1 and MC1_EN. However, while V_CLAMP2 is still in the logic-low state, the second transistor stage MC2 and MC2_EN may still exert some pull on the gate voltage VG. Thus, when V_CLAMP1 initially transitions to the logic-high state (e.g., and V_CLAMP2 remains in the logic-low state), the gate voltage VG may drop slightly toward the output voltage of the differential amplifier 510.
When V_CLAMP2 is in a logic-high state, the transistor MC2_EN is turned off and the voltage clamp transistor MC2 is decoupled from the gates of the ballast transistors MB1-MB3. It is noted that, by the time V_CLAMP2 transitions to the logic-high state, V_CLAMP1 is already in the logic-high state. As a result, control of the gate voltage VG is returned to the differential amplifier 510 (e.g., the voltage clamp transistors MC1/MC1_EN and MC2/MC2_EN no longer have control of the gate voltage VG). Once control of the gate voltage VG has been returned to the differential amplifier 510, the LDO regulator 500 may resume normal operation (e.g., as described above).
The LDO regulator may produce an output voltage across a capacitor based on a received input voltage (610). With reference for example to
The LDO regulator may also generate a control voltage based at least in part on the output voltage and a reference voltage (620). With reference for example to
Further, the LDO regulator may maintain the output current below a threshold level while charging the capacitor from a discharged state to the output voltage (630). With reference for example to
Although specific embodiments have been described with respect to LDO regulators, those of skill in the art will appreciate that the in-rush current protection techniques disclosed herein may be applicable to other types of linear voltage regulators. Further, those of skill in the art will appreciate that information and signals may be represented using any of a variety of different technologies and techniques. For example, data, instructions, commands, information, signals, bits, symbols, and chips that may be referenced throughout the above description may be represented by voltages, currents, electromagnetic waves, magnetic fields or particles, optical fields or particles, or any combination thereof.
Further, those of skill in the art will appreciate that the various illustrative logical blocks, modules, circuits, and algorithm steps described in connection with the aspects disclosed herein may be implemented as electronic hardware, computer software, or combinations of both. To clearly illustrate this interchangeability of hardware and software, various illustrative components, blocks, modules, circuits, and steps have been described above generally in terms of their functionality. Whether such functionality is implemented as hardware or software depends upon the particular application and design constraints imposed on the overall system. Skilled artisans may implement the described functionality in varying ways for each particular application, but such implementation decisions should not be interpreted as causing a departure from the scope of the disclosure.
The methods, sequences or algorithms described in connection with the aspects disclosed herein may be embodied directly in hardware, in a software module executed by a processor, or in a combination of the two. A software module may reside in RAM memory, flash memory, ROM memory, EPROM memory, EEPROM memory, registers, hard disk, a removable disk, a CD-ROM, or any other form of storage medium known in the art. An exemplary storage medium is coupled to the processor such that the processor can read information from, and write information to, the storage medium. In the alternative, the storage medium may be integral to the processor.
In the foregoing specification, embodiments have been described with reference to specific examples thereof. It will, however, be evident that various modifications and changes may be made thereto without departing from the broader scope of the disclosure as set forth in the appended claims. The specification and drawings are, accordingly, to be regarded in an illustrative sense rather than a restrictive sense.
Number | Date | Country | Kind |
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201841031723 | Aug 2018 | IN | national |