Patent Application
20220404439
This application is based on and claims priority under 35 USC § 119 to Japanese Patent Application No. 2021-103331 filed Jun. 22, 2021, the disclosure is incorporated herein by reference in its entirety.
The present invention relates to an information processing device and a magnetic sensor system.
As a related art described in a gazette, in a magnetic field sensor with two CMOS inverters, amorphous wires connected in series with resistors, and a multi-vibrator oscillation circuit configured with a capacitance, there is a frequency-modulated magnetic field sensor in which a power supply voltage is set to not more than 3V and a series resistance of a resistor R and an amorphous wire is set to not more than 500Ω, and the duration during which the input voltage of the two CMOS inverters is held at the threshold voltage is changed by an externally applied magnetic field (refer to Japanese Patent Application Laid-Open Publication No. 2004-108778).
For example, in a magnetic sensor using a magnetic impedance effect element as a sensitive element that senses a magnetic field, an alternating current is supplied to the sensitive element to detect the magnetic field from change in impedance in the sensitive element. On this occasion, the change in impedance at the sensitive element is detected by the change in frequency of the sensitive element in some cases.
In the magnetic impedance effect element, the impedance changes sensitively in the external magnetic field due to the skin effect of a high magnetic permeability alloy magnetic material. In the skin effect, the depth of the surface layer through which the current flows (skin depth δ) is expressed as δ=√(2ρ/ωμ) (ρ: electrical resistivity, ω: angular frequency of flowing current, μ: maximum differential magnetic permeability perpendicular to flowing current), and therefore, the higher the frequency, the thinner the δ and the larger the change in the impedance. Consequently, as the frequency becomes higher, the sensitivity of the magnetic sensor using the magnetic impedance effect element is likely to increase.
However, if the frequency of the output signal outputted from the sensitive element increases, the sampling rate when measures the frequency is insufficient, and thereby the accuracy of the frequency measurement decreases in some cases.
In addition, in the case of detecting a minute magnetic field, the change in frequency to be obtained becomes smaller, and at a limited sampling rate, the frequency resolution is sometimes insufficient to detect the change in the magnetic field. Therefore, it is desirable to increase the frequency change rate by external magnetic fields to improve the detection limit in the frequency measurement.
The present invention provides an information processing device and a magnetic sensor system in which accuracy of frequency measurement is less likely to deteriorate even though the frequency of output signals outputted from the magnetic sensor increases, and which have detection limits for high frequency measurement even with a minute frequency change rate.
Thus, according to the present invention, there is provided an information processing device including: an obtaining part obtaining an output signal outputted by a magnetic sensor and oscillating at a frequency determined in response to strength of a magnetic field; a frequency determination part utilizing interference between the output signal and a reference signal with a reference frequency, which is a frequency used as a reference, to determine the frequency of the output signal; and a magnetic field calculation part calculating the strength of the magnetic field based on the determined frequency of the output signal.
Here, as a result of the interference, the frequency determination part may reduce the frequency of the output signal to a low frequency while maintaining an amount of change in the frequency to improve a rate of change in the frequency, and may use the low frequency to determine the frequency of the output signal.
In addition, as a result of the interference, the frequency determination part may determine the frequency of the output signal from a waveform obtained based on a phase difference between the output signal and the reference signal.
Further, the frequency determination part may determine the frequency of the output signal based on a triangular wave generated by averaging the waveform with respect to time.
Still further, the frequency determination part may cause the output signal having been integrated and averaged and the reference signal to interfere with each other to generate the triangular wave.
The frequency determination part may use plural frequencies as the reference frequency and may cause the output signal and the reference signal to interfere with each other to generate the triangular wave.
Moreover, the interference may be caused by at least one of a logic operation and a phase frequency comparator.
Further, the frequency determination part may cause the interference by mixing the output signal and the reference signal by a superheterodyne system.
Still further, the frequency determination part may utilize aliasing caused by interference between the output signal and the reference signal to determine the frequency of the output signal.
The frequency determination part may change a sampling frequency, as the reference signal, for the output signal to cause the interference.
In addition, the frequency determination part may use a D-flip-flop to cause the interference.
Moreover, the frequency determination part may use a sampling clock signal at a digital signal input port to cause the interference.
Moreover, according to the present invention, there is provided a magnetic sensor system including: a magnetic sensor including a delay generation part serially connecting a sensitive element sensing a magnetic field by a magnetic impedance effect and a capacitive element, and a potential supply part connected to the delay generation part and supplying a potential to cause an alternating current, whose frequency is set by the delay generation part, to flow to the delay generation part, the magnetic sensor outputting the alternating current as an output signal; a frequency determination part utilizing interference between the output signal and a reference signal with a reference frequency, which is a frequency used as a reference, to determine the frequency of the output signal; and a magnetic field calculation part calculating strength of a magnetic field based on the determined frequency of the output signal.
According to the present invention, it is possible to provide an information processing device and a magnetic sensor system in which accuracy of frequency measurement is less likely to deteriorate even though the frequency of output signals outputted from the magnetic sensor increases, and which have detection limits for high frequency measurement even with a minute frequency change rate.
Exemplary embodiments of the present invention will be described in detail based on the following figures, wherein:
Hereinafter, exemplary embodiments according to the present invention will be described with reference to attached drawings.
The magnetic sensor 110 is an oscillation circuit whose frequency changes due to changes in the characteristics of sensitive elements 1 due to an external magnetic field, and various types of oscillation circuits can be used. For example, an LC resonance circuit configured with the inductance of the sensitive element 1 and the capacitive element can be used for the magnetic sensor 110 because, when the inductance of the sensitive element 1 changes in the external magnetic field, the LC resonance frequency changes and thereby the oscillatory frequency changes. In addition, the delay time caused in an RLC series circuit, which combines the resistance and inductance of the sensitive element 1 and the capacitive element, changes because the resistance and inductance of the sensitive element 1 in the external magnetic field change. A relaxation oscillation circuit utilizing the change in the delay time can also be used for the magnetic sensor 110 because the oscillation frequency changes in the external magnetic field. As shown in
The sensitive element part 11 is connected in series with the capacitive element part 12. The sensitive element 1 can be represented by an equivalent circuit in which the resistance and the inductance are connected in series, and each of the resistance and the inductance changes in response to the change in the external magnetic field. The RLC series circuit is formed by connecting the sensitive element part 11 and the capacitive element part 12 in series. Note that, in the exemplary embodiment, the sensitive element part 11 and the capacitive element part 12 function as an example of a delay generation part.
In the frequency setting part 10, the capacitive elements 2 in the capacitive element part 12 repeat the charge and discharge via the sensitive element part 11, and thereby the alternating current flows through the sensitive elements 1 in the sensitive element part 11. On this occasion, in the sensitive element 1, the resistance and the inductance change due to the magnetic field or the change in the magnetic field. Consequently, in the frequency setting part 10, the cycle of the capacitive element 2 repeatedly charging and discharging changes. In other words, the frequency of the alternating current flowing through the sensitive element 1 changes.
In addition to the sensitive elements 1, the sensitive element part 11 may also include other electronic elements, such as resistive elements and inductance elements, that are connected in series or in parallel. In addition to the capacitive elements 2, the capacitive element part 12 may also include other electronic elements, such as resistive elements and inductance elements, that are connected in series or in parallel. Hereinafter, for simplifying the explanation, the description will be given assuming that the sensitive element part 11 is configured with the sensitive elements 1, and the capacitive element part 12 is configured with the capacitive elements 2. Then, the sensitive element part 11 is represented as the sensitive element 1, and the capacitive element part 12 is represented as the capacitive element 2. In other words, the sensitive element 1 is the sensitive element part 11, and the capacitive element 2 is the capacitive element part 12. In
The information processing device 120 processes the alternating current flowing through the magnetic sensor 110 to obtain the strength of the magnetic field.
The information processing device 120 includes a frequency measurement part 30 that measures the frequency of the alternating current flowing through the magnetic sensor 110, and a magnetic field calculation part 40 that calculates the magnetic field or the change in the magnetic field sensed by the sensitive element 1, which will be described later, based on the frequency measured by the frequency measurement part 30.
The frequency measurement part 30 measures the frequency of the alternating current emitted from the magnetic sensor 110, and outputs thereof to the magnetic field calculation part 40. As shown in the figure, the frequency measurement part 30 includes an obtaining part 31 and a frequency determination part 32. Details of the frequency measurement part 30 will be described later.
The magnetic field calculation part 40 calculates the magnetic field or the change in the magnetic field that is sensed by the sensitive element 1 based on the frequency determined by the frequency measurement part 30. The magnetic field calculation part 40 stores the relationship between the resistance and inductance of the sensitive element 1 and the strength of the magnetic field to be sensed. Therefore, the magnetic field calculation part 40 calculates the impedance of the sensitive element 1 from the frequency measured by the frequency measurement part 30, and calculates the magnetic field or the change in the magnetic field that is to be sensed by the sensitive element 1 based on the impedance.
As shown in
Note that a cross-sectional structure of the sensitive element 1 will be described in detail later.
Here, a soft magnetic material in the magnetic material has a small, so-called coercive force, the soft magnetic material being easily magnetized by an external magnetic field, but, upon removal of the external magnetic field, quickly returning to a state with no magnetization or a little magnetization. The hard magnetic material in the magnetic material has a large, so-called coercive force, the hard magnetic material being once magnetized by an external magnetic field, even upon removal of the external magnetic field, maintaining the magnetized state.
Note that, in the present specification, an element constituting the sensitive element 1 (the thin film magnet 60 or the like) is indicated by a two-digit number, and a layer processed into an element (the hard magnetic material layer 503 or the like) is indicated by a number in the five hundreds. Then, for an element, a layer processed into the element is written in parentheses. For example, the case of the thin film magnet 60 is written as the thin film magnet 60 (hard magnetic material layer 503). In the figure, the case is written as 60 (503). In addition, for a layer processed into an element, the element is written in parentheses. For example, the case of the hard magnetic material layer 503 is written as the hard magnetic material layer 503 (thin film magnet 60). The same is true in other cases.
With reference to
The sensitive circuit 70 includes: plural sensitive parts 71 configured with the soft magnetic materials (a soft magnetic material layers 505); connection parts 72 each serially connecting sensitive parts 71 windingly; and terminal parts 73 provided at one end portion and the other end portion of the sensitive parts 71 that are connected in series.
The sensitive part 71 has a reed-shaped planar shape with a longitudinal direction and a short direction. It is assumed that, in the sensitive part 71 shown in
Each sensitive part 71 has, for example, the length in the longitudinal direction of 1 mm to 10 mm, and the width in the short direction of 50 μm to 150 μm. The thickness of the sensitive part 71 (the thickness of the soft magnetic material layer 505) is 0.5 μm to 5 μm. The spacing between the adjacent sensitive parts 71 is 50 μm to 150 μm. Then, the number of sensitive parts 71 is four in
Note that the size of each sensitive part 71 (the length, the area, the thickness, etc.), the number of sensitive parts 71, the spacing between the sensitive parts 71, or the like may be set in accordance with the magnitude of the magnetic field to be sensed, in other words, to be measured. Note that the number of the sensitive parts 71 may be one.
The connection part 72 is provided between end portions of the adjacent sensitive parts 71 to connect the plural sensitive parts 71 in series. In other words, the connection parts 72 are provided to connect the adjacent sensitive parts 71 windingly (meanderingly). In the sensitive element 1 with four sensitive parts 71 shown in
The terminal parts 73 are provided to one end portion and the other end portion of the sensitive parts 71 that are connected in series. In
As described above, the sensitive circuit 70 is configured so that the sensitive parts 71 are windingly connected in series by the connection parts 72, and the electric currents flow from the terminal parts 73a and 73b provided at both end portions of the connected sensitive parts 71. Therefore, the circuit is referred to as the sensitive circuit 70.
Further, the sensitive element 1 includes yokes 80 each of which is provided to face the end portions of the sensitive parts 71 in the longitudinal direction thereof. Here, there are provided two yokes 80a and 80b, each of which is provided to face each of both end portions of the sensitive part 71 in the longitudinal direction thereof. Note that, in the case where the yokes 80a and 80b are not distinguished from each other, the yokes are referred to as the yokes 80. The yoke 80 guides the magnetic force lines to the end portions of the sensitive part 71 in the longitudinal direction thereof. For this reason, the yoke 80 is constituted by the soft magnetic material through which the magnetic force lines are likely to pass. In the example, the sensitive part 71 and the yoke 80 are constituted by the soft magnetic material layer 505. Note that, in the case where the magnetic force lines sufficiently pass in the longitudinal direction of the sensitive parts 71, it is unnecessary to provide the yokes 80.
Next, with reference to
The substrate 50 is composed of a non-magnetic material; for example, an electrically-insulated oxide substrate, such as glass or sapphire, a semiconductor substrate, such as silicon, or a metal substrate, such as aluminum, stainless steel, or a nickel-phosphorus-plated metal.
The adhesive layer 501 is a layer for improving adhesiveness of the control layer 502 to the substrate 50. As the adhesive layer 501, it is preferable to use an alloy containing Cr or Ni. Examples of the alloy containing Cr or Ni include CrTi, CrTa and NiTa. The thickness of the adhesive layer 501 is, for example, 5 nm to 50 nm. Note that, if there is no problem in adhesiveness of the control layer 502 to the substrate 50, it is unnecessary to provide the adhesive layer 501. Note that, in the present specification, composition ratios of alloys containing Cr or Ni are not shown. The same applies hereinafter.
The control layer 502 controls the magnetic anisotropy of the thin film magnet 60 constituted by the hard magnetic material layer 503 to be likely to express in the in-plane direction of the film. As the control layer 502, it is preferable to use Cr, Mo or W, or an alloy containing thereof (hereinafter, referred to as an alloy containing Cr or the like to constitute the control layer 502). Examples of the alloy containing Cr or the like to constitute the control layer 502 include CrTi, CrMo, CrV and CrW. The thickness of the control layer 502 is, for example, 10 nm to 300 nm.
It is preferable that the hard magnetic material layer 503 constituting the thin film magnet 60 uses an alloy that contains Co as a main component and also contains at least one of Cr and Pt (hereinafter, referred to as a Co alloy constituting the thin film magnet 60). Examples of the Co alloy constituting the thin film magnet 60 include CoCrPt, CoCrTa, CoNiCr and CoCrPtB. Note that Fe may be contained. The thickness of the hard magnetic material layer 503 is, for example, 1 μm to 3 μm.
The alloy containing Cr or the like to constitute the control layer 502 has a bcc (body-centered cubic) structure. Consequently, the hard magnetic material constituting the thin film magnet 60 (the hard magnetic material layer 503) preferably has an hcp (hexagonal close-packed) structure easily causing crystal growth on the control layer 502 composed of the alloy containing Cr or the like having the bcc structure. When crystal growth of the hard magnetic material layer 503 having the hcp structure is caused on the bcc structure, the c-axis of the hcp structure is likely to be oriented in a plane. Therefore, the thin film magnet 60 configured with the hard magnetic material layer 503 is likely to have the magnetic anisotropy in the in-plane direction. Note that the hard magnetic material layer 503 is polycrystalline composed of a set of different crystal orientations, and each crystal has the magnetic anisotropy in the in-plane direction. The magnetic anisotropy is derived from crystal magnetic anisotropy.
Note that, to promote the crystal growth of the alloy containing Cr or the like to constitute the control layer 502 and the Co alloy constituting the thin film magnet 60, the alloys should be heated to 100° C. to 600° C. By the heating, the crystal growth of the alloy containing Cr or the like constituting the control layer 502 is likely to be caused, and thereby crystalline orientation is likely to be provided so that the hard magnetic material layer 503 having the hcp structure includes an axis of easy magnetization in a plane. In other words, the magnetic anisotropy is likely to be imparted in a plane of the hard magnetic material layer 503.
The dielectric layer 504 is configured with a nonmagnetic dielectric material and electrically insulates the thin film magnet 60 and the sensitive circuit 70. Examples of the dielectric material constituting the dielectric layer 504 include oxide, such as SiO2, Al2O3, or TiO2, or nitride, such as Si3N4 or AlN. In addition, the thickness of the dielectric layer 504 is, for example, 0.1 μm to 30 μm.
The sensitive part 71 in the sensitive circuit 70 is provided with uniaxial magnetic anisotropy in a direction crossing the longitudinal direction, for example, an intersecting short direction (the width direction). Note that the direction intersecting the longitudinal direction may have an angle exceeding 45° and not more than 90° with respect to the longitudinal direction.
As the soft magnetic material layer 505 constituting the sensitive part 71, it is preferable to use an amorphous alloy, which is an alloy containing Co as a main component doped with a high melting point metal, such as Nb, Ta or W (hereinafter, referred to as a Co alloy constituting the sensitive part 71). Examples of the Co alloy constituting the sensitive part 71 include CoNbZr, CoFeTa and CoWZr. The thickness of the soft magnetic material constituting the sensitive parts 71 is, for example, 0.2 μm to 2 μm.
The connection parts 72 and the terminal parts 73 in the sensitive circuit 70 are configured with a conductor layer 506 having excellent conductivity. For example, Ag, Cu, Au, or Al can be used, but the material is not particularly limited thereto. Note that the connection part 72 and the terminal part 73 may be configured with different conductor layers. In addition, the connection parts 72 and the terminal parts 73 may be formed integrally with the sensitive parts 71. Thus, it is unnecessary to form the sensitive parts 71, the connection parts 72 and the terminal parts 73 separately.
The adhesive layer 501, the control layer 502, the hard magnetic material layer 503 and the dielectric layer 504 are processed to have a quadrangular planar shape (refer to
In the sensitive element 1, the magnetic force lines outputted from the north pole of the thin film magnet 60 once go to the outside of the sensitive element 1. Then, part of the magnetic force lines passes through the sensitive parts 71 via the yoke 80a and goes to the outside again via the yoke 80b. The magnetic force lines that have passed through the sensitive parts 71 return to the south pole of the thin film magnet 60 together with the magnetic force lines that have not passed through the sensitive parts 71. In other words, the thin film magnet 60 applies the magnetic field to the longitudinal direction of the sensitive parts 71.
Note that the north pole and the south pole of the thin film magnet 60 are collectively referred to as both magnetic poles, and when the north pole and the south pole are not distinguished from each other, they are referred to as the magnetic poles.
Note that, as shown in
Here, the spacing between the yoke 80 (the yokes 80a and 80b) and the sensitive circuit 70 should be, for example, 1 μm to 100 μm.
In the above, the sensitive part 71 was assumed to be configured with the single soft magnetic material layer 505; however, it may be possible to configure the soft magnetic material layer 505 with two layers, namely, an upper soft magnetic material layer and a lower soft magnetic material layer, and provide an antiferromagnetically coupled layer between the upper soft magnetic material layer and the lower soft magnetic material layer to antiferromagnetically couples (AFC) the upper soft magnetic material layer and the lower soft magnetic material layer. Examples of materials of such an antiferromagnetically coupled layer include Ru. Provision of the antiferromagnetically coupled layer suppresses demagnetizing fields and improves the sensitivity of the sensitive element 1.
In addition, a conductor layer that reduces the electrical resistance of the sensitive part 71 may be provided between the upper soft magnetic material layer and the lower soft magnetic material layer constituting the sensitive part 71. As the conductor layer, it is preferable to use metal or an alloy having high conductivity, and more preferable to use metal or an alloy that is highly conductive and non-magnetic. Examples of materials of such a conductor layer include metal, such as aluminum, copper, and silver. The thickness of the conductor layer is, for example, 10 nm to 500 nm. Provision of the conductor layer can increase the frequency of the alternating current to be applied to the sensitive circuit 70.
Further, a magnetic domain suppression layer that suppresses occurrence of a closure magnetic domain in the upper soft magnetic material layer and the lower soft magnetic material layer may be provided between the upper soft magnetic material layer and the lower soft magnetic material layer constituting the sensitive part 71. Examples of materials of such a magnetic domain suppression layer include non-magnetic materials, such as Ru and SiO2, and non-magnetic amorphous metals, such as CrTi, AlTi, CrB, CrTa, and CoW. By suppressing occurrence of the closure magnetic domain in the sensitive parts 71, occurrence of noise due to so-called the Barkhausen effect based on magnetic domain wall displacement is also suppressed.
Note that it may be possible to form the soft magnetic material layer 505 constituting the sensitive part 71 to have multiple layers more than two layers, and provide the antiferromagnetically coupled layer, the conductor layer, or the magnetic domain suppression layer between the multiple layers. In addition, it may also be possible to use two or all of the above-described antiferromagnetically coupled layer, conductor layer and magnetic domain suppression layer in combination.
In the above, the sensitive element 1 was assumed to be provided with the thin film magnet 60 and the yokes 80 in addition to the sensitive circuit 70. The thin film magnet 60 is provided to apply the bias magnetic field Hb, which will be described later, to the sensitive parts 71 in the sensitive circuit 70. In the case where the bias magnetic field Hb is applied from the outside of the sensitive element 1, it is unnecessary for the sensitive element 1 to include the thin film magnet 60. In this case, it is also unnecessary to include the adhesive layer 501, the control layer 502, the hard magnetic material layer 503, and the dielectric layer 504, which are provided for the thin film magnet 60. In other words, the sensitive element 1 should be configured with the sensitive circuit 70 provided on the substrate 50. In this case, the yokes 80 may or may not be provided.
Examples of methods for applying the bias magnetic field Hb from the outside of the sensitive element 1 include methods using the permanent magnet or bias coil. It may be possible only to place them close to the sensitive element 1; however, it is preferable to use a magnetic path configured with a soft magnetic material, such as a ferrite core, to suppress the leakage of the bias magnetic field due to the permanent magnet or the bias coil to the outside.
In the case where the sensitive circuit 70 is provided on the substrate 50, if the substrate 50 is configured with a semiconductor substrate, such as silicon, or a metal substrate, such as aluminum, stainless steel, or a nickel-phosphorus-plated metal, the substrate 50 has high conductivity. In such a case, an insulating material layer to electrically insulate the substrate 50 from the sensitive circuit 70 should be provided on the surface of the substrate 50 on which the sensitive circuit 70 is provided. Examples of the insulating material constituting the insulating material layer include, similar to the dielectric material constituting the dielectric layer 504, oxide such as SiO2, Al2O3, or TiO2, or nitride such as Si3N4 or AlN.
The sensitive element 1 in the present specification may be, in addition to the one that includes the thin film magnet 60 as shown in
Subsequently, the action of the sensitive element 1 will be described.
As shown in
The sensitive element 1 is brought into the state in which the bias magnetic field Hb is applied in advance by the thin film magnet 60 shown in
By looking at the sensitive element 1 as an impedance element, the above description can be given; however, the sensitive element 1 can be represented by an equivalent circuit connecting the resistance and the inductance in series, and is combined with the capacitive element to form an RLC series circuit; therefore, in considering the characteristics thereof, it is necessary to consider the characteristics of the resistance R, the reactance X, and the inductance L.
Here, the impedance Z, the resistance R, the reactance X, and the inductance L are in the following relation, where ω is the angular frequency (rad/sec), and f is the frequency (Hz):
Z=R+jX=R+jωL ω=2πf
In the sensitive element shown in
As shown in
The inverter INV1 includes an input terminal IN1 and an output terminal OUT1. The inverter INV2 includes an input terminal IN2 and an output terminal OUT2. In the following description, the input terminal is referred to as input, and the output terminal is referred to as output.
The inverters INV1 and INV2 are elements that invert the logic level of the input signal to generate the output signal. In other words, when the logic level “H” is inputted to the input IN1, the inverter INV1 outputs the inverted logic level “L” from the output OUT1, and when the logic level “L” is inputted to the input IN1, the inverter INV1 outputs the inverted logic level “H” from the output OUT1. Note that, in the case where the inverters INV1 and INV2 are not distinguished from each other, the inverters are referred to as inverters INV. Moreover, inversion of the output from the inverter INV is sometimes referred to as switching.
Here, it is assumed that the connection point where one of the terminal parts 73 (for example, the terminal part 73a (refer to
The inverters INV1 and INV2 are connected in series. In other words, the output OUT1 of the inverter INV1 is connected to the input IN2 of the inverter INV2. Then, the point β of the frequency setting part 10 is connected to the connection point between the output OUT1 of the inverter INV1 and the input IN2 of the inverter INV2. That is, the point β, the output OUT1, and the input IN2 have the same potential. Consequently, these are sometimes referred to as the point β (output OUT1), the point β (input IN2), or the like to indicate to have the same potential. Then, the point α of the frequency setting part 10 is connected to the input IN1 of the inverter INV1, and the point γ of the frequency setting part 10 is connected to the output OUT2 of the inverter INV2. In other words, the point α and the input IN1 of the inverter INV1 have the same potential, and the point γ and the output OUT2 of the inverter INV2 have the same potential. Consequently, these are sometimes referred to as the point α (input IN1), the point γ (output OUT2), to indicate to have the same potential.
In
The operation of the inverters INV1 and INV2 will be described with reference to the inverter INV1.
When the input IN1 of the inverter INV1 is at the ground potential GND (the logic level “L”), the transistor pTr1 is turned on, the transistor nTr1 is turned off, and the output OUT1 is at the power supply potential VCC (the logic level “H”). Conversely, when the input IN1 of the inverter INV1 is at the power supply potential VCC (the logic level “H”), the transistor pTr1 is turned off, the transistor nTr1 is turned on, and the output OUT1 is at the ground potential GND (the logic level “L”). Then, when the input IN1 transitions from the ground potential GND side to the power supply potential VCC side beyond the threshold voltage Vth, when the input IN1 reaches the threshold voltage Vth, the output OUT1 inverts from the power supply potential VCC (the logic level “H”) to the ground potential GND (the logic level “L”). Conversely, when the input IN1 transitions from the power potential VCC side to the ground potential GND side below the threshold voltage Vth, when the input IN1 reaches the threshold voltage Vth, the output OUT1 inverts from the ground potential GND (the logic level “L”) to the power supply potential VCC (the logic level “H”).
First, description will be given of the case where the frequency setting part is an RC series circuit.
In
In
At the time t1, when the point α (the input IN1) reaches the threshold voltage Vth (2.5V) from the 0V side, the inverter INV1 inverts and the point β (the output OUT1) is shifted from 5V to 0V. This causes the point β of the inverter INV2 (the input IN2) to shift from 5V to 0V, and thereby the inverter INV2 inverts and the point γ (the output OUT2) is shifted from 0V to 5V. Just before the inverter INV1 is inverted, the point α is the threshold voltage Vth (2.5V), and the point γ is 0V. Since the voltage applied to the capacitive element 2(C) is the potential difference between the points α and γ, the voltage applied to the capacitive element 2(C) is 2.5 V. When the inverter output is inverted instantaneously, the capacitive element 2(C) is not charged or discharged instantaneously, and to maintain the potential difference of 2.5 V, when the point γ (the output OUT2) is shifted from 0V to 5V due to the inversion of the inverter INV2, the point α is shifted to 7.5V, which is the result of addition of 5V to the threshold voltage Vth (2.5V).
Thereafter, from the point α of 7.5V to the point β of 0V, the electric charges accumulated in the capacitive element 2(C) flow through the sensitive element 1(R). Consequently, the potential at the point α gradually decreases. The transient response of the decrease is determined by the time constant τ, which is determined by the resistance R of the sensitive element 1 and the capacitance C of the capacitive element 2. Here, the time constant τ is given by
τ=RC.
Since the voltage at time t1 is VCC+threshold voltage Vth, the potential V(t) at the point α after t seconds have elapsed from the time t1 is
V(t)=(VCC+Vth)e−t/τ, and the potential gradually decreases from 7.5V.
Then, at the time t2, when the point α (the input IN1) decreases from 7.5V to reach the threshold voltage Vth, the inverter INV1 inverts and the point β (the output OUT1) is shifted from 0V to 5V. This causes the point β of the inverter INV2 (the input IN2) to shift from 0V to 5V, and thereby the inverter INV2 inverts and the point γ (the output OUT2) is shifted from 5V to 0V. Just before the inverter INV1 is inverted, the point α was the threshold voltage Vth (2.5V). Accordingly, when the inverter INV2 is inverted and the point γ (the output OUT2) is shifted from 5V to 0V, the point α is shifted to −2.5V, which is the result of subtraction of 5V from the threshold voltage Vth (2.5V). The transient response of the increase is also determined by the time constant τ.
Thereafter, by the current flowing from the point β of 5V to the point α of −2.5V through the sensitive element 1(R) to accumulate electric charges in the capacitive element 2(C), the potential of the point α gradually increases.
Then, at the time t3, when the point α (the input IN1) increases from −2.5V to reach the threshold voltage Vth (2.5V), the inverter INV1 inverts and the point β (the output OUT1) is shifted from 5V to 0V. This causes the point β of the inverter INV2 (the input IN2) to shift from 5V to 0V, and thereby the inverter INV2 inverts and the point γ (the output OUT2) is shifted from 0V to 5V. Just before the inverter INV1 is inverted, the point α was the threshold voltage Vth (2.5V). Accordingly, when the inverter INV2 is inverted and the point γ (the output OUT2) is shifted to 5V, the point α is shifted to 7.5V, which is the result of addition of 5V to the threshold voltage Vth (2.5V). In other words, the time t3 is the same as the time t1.
Thereafter, the potential change from the time t1 to the time t3 is repeated.
In
The voltage at the point α based on of the point γ means the voltage applied to the capacitive element 2(C). In addition, the voltage at the point β based on the point γ means the voltage applied to the frequency setting part 10.
The potential at the point β is shifted from 5V to −5V at the time t1, and shifted from −5V to 5V at the time t2. Then, at the time t3, the shift from 5V to −5V occurs similar to the time t1. In other words, the potential difference between the points β and γ point is always +/−5V. Since the inverters INV1 and INV2 constitute a full bridge, the voltage change of 10 V occurs at the frequency setting part 10 when the output of the inverter is inverted.
On the other hand, the point α is the threshold voltage Vth (2.5V) at the time t1, shifted to the voltage deviated by the threshold voltage Vth toward the negative side (−2.5V) at the time t2, and returns to the threshold voltage Vth (2.5V) at the time t3.
Consequently, the capacitive element 2(C) is charged and discharged at the voltage of +/−Vth.
Then, at both points α and β, the voltage change from the time t1 to the time t3 is repeated.
In
Since the voltage at the point γ is the potential difference between the points α and γ, the voltage at the point α shown in
At the time t1, the voltage at the point β is shifted from the threshold voltage Vth (2.5V) to −7.5V, which is the result of addition of the threshold voltage Vth (2.5V) to the power supply potential VCC (5V) on the negative side. Then, the voltage gradually increases from the time t1 to the time t2, and at the time t2, the voltage reaches the threshold voltage Vth on the negative side (−2.5V). At the time t2, the voltage is shifted from −2.5V to 7.5V, which is the result of addition of the threshold voltage Vth (2.5V) to the power supply potential VCC (5V). Then, the voltage gradually decreases from the time t2 to the time t3, and at the time t3, the voltage reaches the threshold voltage Vth (2.5V). At the time t3, the voltage is shifted from 2.5V to −7.5V, which is the result of addition of the threshold voltage Vth (2.5V) to the power supply potential VCC (5V) on the negative side.
Then, at the point β, the voltage change from the time t1 to the time t3 is repeated.
As described above, an AC voltage with a period from the time t1 to the time t3 is applied to the sensitive element 1(R), and an alternating current flows therethrough. The frequency of the alternating current is determined by the resistance R of the sensitive element 1 and the capacitance C of the capacitive element 2. Then, the sensitive element part 11 including the sensitive element 1 and the capacitive element part 12 including the capacitive element 2 are collectively referred to as the frequency setting part 10. The resistance R of the sensitive element 1 changes depending on the magnetic field H as shown in
Next, description will be given of the case where the frequency setting part is an RLC series circuit.
Here, the sensitive element 1(RL) is a series circuit with the resistance R and the inductance L as the equivalent circuit of the sensitive element, and is further connected in series with the capacitive element 2(C) to form an RLC series circuit. Description will be given of the operation of the magnetic sensor 110, to which the first exemplary embodiment is applied, in this case.
The basic operation is similar to the case shown in
Thus, the frequency of the alternating current is determined by the resistance R, the inductance L of the sensitive element 1, and the capacitance C of the capacitive element 2. Then, the resistance R and the inductance L of the sensitive element 1 change depending on the magnetic field H as shown in
Note that the description was given of the magnetic sensor 110 in the first exemplary embodiment as the relaxation oscillation circuit by the RLC series circuit; however, various types of oscillation circuits can be used for the magnetic sensor 110 as long as the oscillation circuit has frequency change caused by the change in the characteristics of the sensitive element 1 due to the external magnetic field.
For example, an LC resonance circuit is configured with the inductance of the sensitive element and the capacitive element, to thereby make it possible to use an LC oscillation circuit utilizing the change in the LC resonance frequency due to the change in the inductance of the sensitive element caused by the external magnetic field. Examples of the LC oscillation circuit include a Colpitts oscillation circuit. If the sensitive element 1(RL) is used instead of the inductance L of the LC oscillation circuit, the frequency of the oscillation circuit changes because the inductance L of the sensitive element 1(RL) changes due to the change in the external magnetic field.
As described above, the frequency measurement part 30 includes the obtaining part 31 and the frequency determination part 32.
The obtaining part 31 obtains the output signal from the magnetic sensor 110. The output signal can be said to be an alternating current oscillating at a frequency determined in accordance with the strength of the magnetic field by use of the magnetic sensor 110.
Note that the obtaining part 31 not only utilizes the output signal from the magnetic sensor 110 as it is, but also performs predetermined processing thereon. For example, the obtaining part 31 shapes the output signal from the magnetic sensor 110 to form a square wave. Then, the square wave is used as an output signal. In addition, it is preferable that the obtaining part 31 integrates and averages the output signal as the predetermined processing. This reduces phase noise and improves the accuracy of the frequency measurement of the output signal.
Note that, if the output signal of the magnetic sensor 110 has less noise and the frequency is stable, the output of the magnetic sensor 110 can be outputted directly to the frequency determination part 32 without the obtaining part 31.
The frequency determination part 32 determines the frequency of the output signal. In the exemplary embodiment, the frequency determination part 32 utilizes interference between the output signal and a reference signal having a reference frequency, which is the frequency used as a reference, to determine the frequency of the output signal.
The obtaining part 31 and the frequency determination part 32 are achieved by cooperation of the hardware resources and software. Examples of the hardware resources include computer devices, such as logic ICs to be described later, mixers, D-flip-flop circuits, and personal computers (PCs). In addition, regarding the software, a not-shown CPU in a computer device loads a program as the software into a main memory (not shown) and executes thereof to perform processing of various types of signals.
Next, a configuration of the frequency determination part 32 will be described in detail.
Conventionally, the amplitude of the output signal was measured, and the strength of the magnetic field was calculated based on the measured amplitude. This could be said that the voltage of the output signal was measured, and the strength of the magnetic field was calculated based on the measured voltage. In this case, it is difficult to detect the minute changes in voltage, and the accuracy of the strength of the magnetic field calculated from the detected minute changes in voltage is likely to decrease. For example, the difference of 1 ppm corresponds to 1 μV for a voltage of 1V, which is the difference buried in noise. There is also a method to amplify the voltage and increase the difference, but the noise created prior to amplification is also amplified, and therefore, it is difficult to detect a change in voltage.
Therefore, in the exemplary embodiment, the frequency of the output signal is measured to solve the problem. This can also be said to be a time measurement of the output signal. As compared to measurement of the voltage, it is easier to measure minute changes in frequency, and the accuracy of the strength of the magnetic field to be calculated from the measured minute changes in frequency is likely to be improved. For example, the difference of 1 ppm corresponds to the accuracy of 2.6 seconds per month; accordingly, the difference can be detected by the time measurement.
However, if the frequency of the output signal becomes high frequency, it is sometimes difficult to simply detect the changes in frequency. In such a case, it is necessary to increase the frequency resolution to detect the minute changes in frequency. In other words, it is necessary to increase the temporal resolution, and high-speed sampling is required. On the other hand, if the frequency of an output signal becomes 100 MHz or more, it is difficult to perform high-speed sampling that measures one period at high resolution. For example, in the case where the frequency of the output signal is 200 MHz, one period is 5 n seconds. At this time, 1 ppm is 5 f seconds, which is too short and difficult to measure.
On the other hand, if the frequency of the output signal is too low, there is a problem that the temporal resolution of the magnetic field strength, which varies in time, is reduced. For example, with a signal frequency of 1 Hz, the waveform of the magnetic field strength that varies at a period of 1 Hz cannot be represented. In the case where the magnetic field strength that varies at a period of 1 Hz is expressed by 100 points/period, the temporal resolution of 10 m seconds is required. To achieve this, it is necessary to accurately read the signal frequency within 10 m seconds; however, if the signal frequency is 1 Hz, it is required to read the signal frequency from a waveform of 1/100 of one period, and therefore, it is difficult to measure accurately. Consequently, it is necessary to convert the signal frequency into an adequate frequency, with which the minute changes in frequency can be detected and the temporal resolution of the magnetic field strength variation can be sufficiently high.
To solve this, for example, if the frequency of the output signal is divided, the frequency can be converted into a lower frequency, and the frequency resolution at a predetermined sampling rate is improved. However, this method also reduces the amount of change in frequency; therefore, the rate of change in frequency is not improved.
For example, in the case where the difference of 1 ppm in the frequency of 200 MHz is detected, if the frequency is divided to 1/1,000,000, the time variation of 1 ppm becomes 5 n seconds, which can be measured. However, since the frequency of the output signal is also converted to 1/1,000,000, the output signal of 200 MHz output signal is converted into 2 Hz, and thereby the temporal resolution in the magnetic field signal detection is reduced. For this reason, a method that increases the rate of change in the frequency by reducing the frequency without changing the amount of change in the frequency as much as possible is required.
Accordingly, in the exemplary embodiment, in measuring the frequency of the output signal, interference between the output signal and a reference signal having a reference frequency, which is the frequency used as a reference, is utilized to perform the time measurement, to thereby solve the problem, as described below.
To begin with, a first exemplary embodiment will be described. In the first exemplary embodiment, the frequency determination part 32 determines the frequency of the output signal on the basis of the waveform obtained based on the phase difference between the output signal and the reference signal as a result of interference between the output signal and the reference signal.
Here, description will be given of the case in which the logical operation causes the output signal and the reference signal to interfere with each other.
The circuit configuration shown in the figure includes: a reference signal output part 311 that outputs the reference signal; an output signal division part 312 that divides the output signal emitted by the magnetic sensor 110; a reference signal division part 313 that divides the reference signal outputted from the reference signal output part 311; a logical operation part 314 that performs logical operations; a triangular wave generation part 315 that generates triangular waves; an A/D conversion part 316 that performs A/D conversion; a signal processing part 317 that obtains the frequency of the triangular wave; and an output part 318 that outputs the frequency.
The reference signal output part 311 outputs a reference signal at a predetermined frequency. Here, the output signal from the magnetic sensor 110 has been converted into a square wave in advance by the obtaining part 31. Then, the reference signal output part 311 outputs a square wave at a predetermined frequency as the reference signal. It is preferable that the reference signal output part 311 has a frequency that is relatively close to the frequency of the output signal. Therefore, the frequency of the reference signal is not necessarily the one, but a reference signal with plural frequencies is prepared. The reference signal with plural frequencies can be generated by dividing the reference signal in the reference signal division part 313.
The output signal division part 312 divides the output signal as needed. Although the phase noise is reduced by being integrated and averaged by dividing, the temporal resolution of magnetic field detection is also reduced; accordingly, it is desirable to keep the division ratio to the minimum capable of reducing the phase noise that can be accurately processed by the logical operation part 314. In addition, the reference signal division part 313 divides the reference signal as needed. The output signal division part 312 and the reference signal division part 313 can be configured with a binary counter or a prescaler. In addition, a phase locked loop (PLL), such as a clock generator, can provide a configuration that integrates the reference signal output part 311 and the reference signal division part 313.
The logical operation part 314 performs a logical operation of the output signal divided by the output signal division part 312 and the reference signal divided by the reference signal division part 313. These signals are square waves, and the logical operation is performed assuming 1 when the pulse is ON and assuming 0 when the pulse is OFF. The logical operation is not particularly limited, and any of XOR (exclusive OR), NOR (negative OR), NAND (not AND), OR (logical OR), and AND (logical AND) can be used. The logical operation part 314 can be configured, for example, by the logic IC. In addition, the phase frequency comparator of the PLL can also be used.
The triangular wave generation part 315 generates a triangle wave based on the waveform outputted by the logical operation part 314.
Of these,
Then,
As shown in the upper side of
In addition, as shown in the upper side of
In other words, the waveform generated by the XOR (exclusive OR) in the logical operation part 314 is a square wave with a shorter pulse width as the phases of the output signal and the reference signal match better, and is a square wave with a longer pulse width as the phases of the output signal and the reference signal are shifted more from each other. That is, pulse width modulation (PWM) output is obtained.
Then, the triangular wave generation part 315 averages the waveform of the square waves generated as shown in
Then, in the case where this is applied to the longer time as shown in
The frequency of the triangular wave Ws1 is determined by the amount of shift between the frequency of the output signal and the reference frequency. In other words, the more the frequency of the output signal and the frequency of the reference signal match, the longer the period is. That is, the frequency of the triangular wave Ws1 is further reduced. In contrast thereto, as the frequency of the output signal and the frequency of the reference signal are shifted further and differ from each other, the period becomes shorter. That is, the frequency of the triangular wave Ws1 is more increased.
The frequency of the triangular wave Ws1 is represented by an absolute value of the difference between the frequency of the output signal and the frequency of the reference signal.
Consequently, based on the frequency of the triangular wave Ws1, the difference between the frequency of the output from the output signal division part 312 that was inputted to the logical operation part 314 and the frequency of the output from the reference signal division part 313 can be obtained, and further, the difference between the frequency of the output signal and the frequency of the reference signal can be obtained from the division ratio of the output signal division part 312 and the reference signal division part 313. Since the frequency of the reference signal is determined in advance, the frequency of the output signal can be obtained from the difference. On the other hand, since the frequency of the triangular wave Ws1 is represented by an absolute value of the difference between the frequency of the output signal and the frequency of the reference signal, the amount of change in the frequency of the output signal that has been minutely changed by the external magnetic field is maintained as it is. In the triangular wave Ws1, the frequency change rate increases because the frequency decreases while the amount of change in the frequency of the output signal is maintained. Accordingly, it becomes possible to measure the minute frequency difference caused by the external magnetic field.
Note that
The frequencies of the triangular waves Ws1 and Ws2 are lower than the frequency of the output signal. Accordingly, the frequencies of the triangular waves Ws1 and Ws2 can be detected at a lower sampling rate than to directly detect the frequency of the output signal. In other words, by detecting the frequencies of the triangular waves Ws1 and Ws2, the frequency of the output signal can be obtained without conducting the high-speed sampling.
In this case, it can also be said that, as a result of interference between the output signal and the reference signal, the frequency determination part 32 reduces the frequency of the output signal to a low frequency and increases the rate of change, and uses the output signal of the low frequency and the large rate of change, to thereby determine the frequency of the output signal.
Returning to
The signal processing part 317 obtains the frequency of the triangular wave, and then obtains the frequency of the output signal based on the frequency of the triangular wave. This can be carried out by, for example, the fast Fourier transform (FFT). The FFT can be carried out, for example, by taking the digital signal converted by the A/D conversion part 316 into the PC and executing a predetermined program to perform the processing of the FFT.
In addition, not only the FFT, but also various methods can be used, such as measuring the time taken for one period of a triangular wave, or counting the number of periods of a triangular wave within a predetermined time.
Note that, in the case where the output signal is divided by the output signal division part 312, the frequency is corrected in accordance with the division ratio N.
The output part 318 outputs the obtained frequency of the output signal to the magnetic field calculation part 40.
In the above-described example, the logical operation part 314 can achieve the similar function by using a phase frequency comparator (PFC) of, not only the logic IC, but also the phase locked loop (PLL) IC. In other words, the output signal divided by the output signal division part 312 and the reference signal divided by the reference signal division part 313 are inputted to the phase frequency comparator. The phase frequency comparator compares the phases of the divided signals to obtain the phase difference, to thereby detect the digital lock. In the case where the phase difference between the divided signals is within 15 n seconds, for example, and if such a phase difference is detected three consecutive times, a digital lock is detected, and alternatively, if the phase difference is 30 n seconds or more, for example, the digital lock is released. Then, the error signal pulse of the phase frequency comparator when the digital lock is detected is used. The error signal pulse is shown in
Next, a second exemplary embodiment will be described. In the second exemplary embodiment, the frequency determination part 32 mixes the output signal and the reference signal to cause interference, and converts the frequency of the output signal by the superheterodyne system.
The circuit configuration shown in the figure includes: a reference signal output part 321 that outputs the reference signal; a mixing part 322 that mixes the output signal and the reference signal; an A/D conversion part 323 that performs A/D conversion; a signal processing part 324 that obtains the frequency of the triangular wave; and an output part 325 that outputs the frequency.
The reference signal output part 321 outputs a reference signal at a predetermined frequency. The reference signal is generated, for example, by the clock generator. Then, as the reference signal, a clock signal with a square wave is outputted.
The mixing part 322 mixes the output signal and the reference signal. This causes the mixing part 322 to generate a mixing signal with an intermediate frequency, which is the difference between the frequency of the output signal and the frequency of the reference signal. The intermediate frequency is lower than the frequency of the output signal. In other words, the mixing by mixing part 322 causes interference between the output signal and the reference signal, which results in a low frequency of the output signal. That is, the output signal is changed to have a low frequency by the superheterodyne system. For example, in the case where the frequency of the output signal is 100 MHz and the frequency of the reference signal is 99.9 MHz, the intermediate frequency is 100 kHz.
The A/D conversion part 323 that performs A/D conversion, the signal processing part 324 that obtains the frequency of the A/D converted waveform, and the output part 325 that outputs the frequency are the same as the A/D conversion part 316, the signal processing part 317, and the output part 318 in
Since the magnetic field signal thus obtained can reduce the frequency while maintaining the rate of change due to the interference by the superheterodyne system, the frequency change rate can increase while converting the frequency to the adequate value to be detected; therefore, it becomes possible to perform high sensitivity measurement of the minute magnetic field signal.
Next, a third exemplary embodiment will be described. In the third exemplary embodiment, the frequency determination part 32 utilizes the aliasing that occurs between the output signal and the reference signal to determine the frequency of the output signal.
The circuit configuration shown in the figure includes: a reference signal output part 331 that outputs the reference signal; an interference part 332 that causes the output signal and the reference signal to interfere with each other to generate aliasing; a signal processing part 333 that obtains the frequency of the waveform observed by aliasing; and an output part 334 that outputs the frequency.
The reference signal output part 331 outputs a reference signal at a predetermined frequency. The reference signal is generated, for example, by the clock generator. Then, as the reference signal, a clock signal with a square wave is outputted.
Note that, similar to the first exemplary embodiment, it may be possible to provide a division part to divide at least one of the output signal and the reference signal in advance.
The interference part 332 causes the output signal and the reference signal to interfere with each other. The interference part 332 is, for example, a D-flip-flop.
Here, the frequency observed when sampling a signal, which is a sine wave, is shown. In
Of these,
Further,
The aliasing occurs when the frequency of the signal is more than ½ of the sampling frequency, as shown in
Note that the frequency of the signal is changed here; however, in practice, the interference part 332 changes the frequency of the sampling to cause aliasing. That is, the frequency of the clock signal, which is the reference signal, is changed. In this case, it can be said that the interference part 332 changes the sampling frequency for the output signal as a reference signal, to thereby cause interference.
To understand the concept of aliasing, the description was given by the analogue signal of the sine wave; however, since the digital signal causes similar aliasing, it is possible to use aliasing to measure minute frequency changes.
Here, the frequency observed when sampling a digital signal is shown. In
First, description will be given of the example in which the frequency of the output signal from the magnetic sensor 110 does not cause aliasing with respect to the reference signal from the reference signal output part 331.
The interference part 332 is a D-flip-flop. The D-flip-flop inputs and outputs the digital signal. That is, signals at the logic level “H” and the logic level “L” are inputted and outputted.
Here, a D-flip-flop including three terminals, a CLK terminal, a D terminal, and a Q terminal, is used. The CLK terminal is an input terminal for the reference signal. The D-flip-flop stores the input to the D terminal at the moment when the reference signal rises. The D terminal inputs the output signal from the magnetic sensor 110 to the D-flip-flop. The Q terminal outputs the state of the logic level stored by the D-flip-flop.
The reference signal shown in
The arrows (1) to (4) in
In the arrow (1), since the logic level in
Next, description will be given of the example in which the frequency of the output signal from the magnetic sensor 110 causes aliasing with respect to the reference signal from the reference signal output part 331.
The reference signal shown in
The arrows (1) to (5) in
In the arrow (1), since the logic level in
Thereafter, at the timing of the arrow (2), the logic level in
The logic level in
Then, when the signals in
As described above, aliasing occurs when the frequency in
Due to the shift between the change in the logic level in
However, if the frequencies are too close, the interference frequency by aliasing becomes too low, and thereby the temporal resolution of the magnetic field signal detection is reduced. In addition, if the frequencies are too close, there is a problem that stable interference signals cannot be obtained due to the jitter of the frequency of the output signal from the magnetic sensor 110.
Consequently, it is preferable that the difference between the frequency of the output signal from the magnetic sensor 110 and the frequency of the reference signal of the reference signal output part 331 satisfies the temporal resolution of the magnetic field signal detection and is kept as small as possible within the range capable of obtaining the stable interference signals.
The signal processing part 333 obtains the frequency of the waveform observed by aliasing and then obtains the frequency of the output signal based on the frequency that has been obtained. This can be carried out by FFT, for example, as in the first exemplary embodiment. In addition, not only the FFT, but also various methods can be used, such as measuring the time taken for one period of the output from the interference part 332, or counting the number of periods of the output from the interference part 332 within a predetermined time, similar to the first exemplary embodiment.
Next, a fourth exemplary embodiment will be described. The basic operations are the same as those in the third exemplary embodiment, and the frequency of the output signal is determined by using aliasing that occurs between the output signal and the reference signal.
The circuit configuration is also the same as
Usually, the digital input port samples at a predetermined sampling frequency or a sampling frequency that can be arbitrarily set. The mechanism is similar to that described with reference to
Accordingly, even though the interference part 332 is not provided, if the frequency difference between the output signal of the magnetic sensor 110 and the sampling frequency causes aliasing, the interference between the frequencies causes aliasing, and thereby the frequency of the input signal is reduced, and the rate of change is increased.
To obtain a frequency difference that causes aliasing, the output signal from the magnetic sensor 110 should be divided and adjusted to be close to the sampling frequency of the digital input port.
In addition, the similar effect can also be obtained by setting the sampling frequency of the digital input port to be close to the output frequency of the magnetic sensor 110.
Moreover, it may also be possible to divide the output frequency of the magnetic sensor 110, and to set the sampling frequency of the digital input port to have a predetermined frequency.
The signal sampled at the digital input port is converted into a frequency in the signal processing part 333, and then converted into a signal magnetic field and output from the output part 334.
The fourth exemplary embodiment does not require the interference part 332; accordingly, since the device can be made compact and at a lower cost, it is suitable to configure a system that measures plural channels simultaneously.
To verify the performance of the exemplary embodiment, operations were verified by using a clock generator instead of the magnetic sensor 110 in
The output signal of the clock generator is subjected to sampling, data processing, and output at an FPGA. The FPGA includes the functions of the reference signal output part 331, the interference part 332, the signal processing part 333, and the output part 334. The function of the reference signal output part 331 is achieved by generating a sampling frequency by the PLL in the FPGA. The sampling frequency was set to 300 MHz. The function of the interference part 332 is achieved by sampling the input signal by the digital input port. Since the digital input port samples the input signal at the sampling frequency generated by the PLL, aliasing occurs when the frequency of the input signal is close to the sampling frequency, and the digital signal is measured at the frequency difference between both frequencies.
The frequency was measured by counting the number of sampling clocks between the rising edges of the digital signal measured at the digital input port (corresponding to one period of the measured frequency). For example, in the case where the measured frequency is 100 kHz, the time between the rising edges of the measured frequency is 10 μ seconds, during which the sampling clock of 300 MHz is counted 3000 times. The counted number is converted into the frequency by a numerical operation programmed into the FPGA, and further converted into the frequency of the input signal in consideration of aliasing (the function of the signal processing part 333). This value is shown on the display as a graph or value and is written to the external storage device (the function of the output part 334). Note that, in this case, it can be said that the interference part 332 uses the sampling clock signal at the digital signal input port to cause interference.
The “sampling frequency” is the sampling frequency of the digital input port set by the PLL in the FPGA, the “input frequency” is the frequency of the signal inputted from the clock generator to the digital input port of the FPGA, and “Δ” is the difference between the “sampling frequency” and the “input frequency.”
The “measured frequency” is the frequency reduced by aliasing at the digital input port of the FPGA and measured, the “after conversion” is the input frequency calculated from the difference between the “sampling frequency” and the “measured frequency,” the “error” is the measurement error (Hz) calculated from the difference between the “input frequency” and the “after conversion,” and the ratio (ppm) obtained by dividing the measurement error by the “input frequency,” and the “frequency variation” is a value obtained by dividing the standard deviation of the “measured frequency” by the “input frequency.”
The frequency of the “after conversion” is approximately the same as the “input frequency,” which indicates that the frequency can be measured accurately and theoretically in the fourth exemplary embodiment.
In the above-described first to fourth exemplary embodiments, the frequency determination part 32 reduces the frequency of the output signal to the low frequency as a result of interference between the output signal and the reference signal. This is the frequency based on the phase difference between the output signal and the reference signal. Then, the output signal of the low frequency is used to determine the frequency of the output signal. Consequently, even in the case where the output signal from the magnetic sensor 110 is a high-frequency signal of 100 MHz or higher, due to reduction to the low frequency, sufficient frequency resolution can be obtained even at the normal sampling rate. The accuracy of the frequency measurement is less likely to decrease, and the magnetic field strength can be measured more accurately. In addition, this method does not decrease the amount of change in the frequency and increases the rate of change in the frequency; accordingly, it becomes possible to measure the magnetic field with high sensitivity. Moreover, even in the case where the division is used, since the division ratio can be smaller than before, the problem of reduction in the rate of change in the frequency is less likely to occur.
The foregoing description of the exemplary embodiments of the present invention has been provided for the purposes of illustration and description. It is not intended to be exhaustive or to limit the invention to the precise forms disclosed. Obviously, many modifications and variations will be apparent to practitioners skilled in the art. The exemplary embodiments were chosen and described in order to best explain the principles of the invention and its practical applications, thereby enabling others skilled in the art to understand the invention for various embodiments and with the various modifications as are suited to the particular use contemplated. It is intended that the scope of the invention be defined by the following claims and their equivalents.
| Number | Date | Country | Kind |
|---|---|---|---|
| 2021-103331 | Jun 2021 | JP | national |
