This nonprovisional application claims priority under 35 U.S.C. § 119(a) on German Patent Application No. DE 102006017188, which was filed in Germany on Apr. 12, 2006, and which is herein incorporated by reference.
1. Field of the Invention
The present invention relates to an integrated differential oscillator circuit which has an amplifier circuit with an input and an output, and has a frequency-selective feedback network with a first inductor and a DC power supply.
2. Description of the Background Art
An oscillator circuit is known from WO 99/43079, which corresponds to U.S. Pat. No. 6,002,303. This document shows a differential oscillator design with two resonant circuits that are deattenuated through an amplifier circuit of two transistors in common-base configuration. In the terminology of WO 99/43079, the first resonant circuit has a first resonant element, a first feedback path, and a differential coupling element. Various embodiments are specified, which result from different combinations of resistive, capacitive, magnetic and inductive embodiments of the components.
In one embodiment, which appears to illustrate an inductive feedback, the first resonant element has a resistive character, the feedback path has an inductive character, and the differential coupling element has a capacitive character. In the drawings, the feedback path is parallel to the collector-emitter path of one of the two transistors and is closed through an inductive component, which would mean a DC short circuit of the collector-emitter path in an embodiment of the inductive component as a coil.
In three other embodiments, capacitive feedback paths are specified. The differential coupling element lies between nodes to which are connected the emitters of the transistors, the feedback paths, and, in each case, one bias element that connects one of the nodes to a ground. This ground obviously represents a DC ground, since WO 99/43079 expressly distinguishes this ground from a “virtual ground point,” which is to say from AC ground. Current sources or current sinks are disclosed as bias elements. The terminals of the current sources/current sinks connected to the transistors are separated from one another only by the differential coupling element. A separate bias element in the form of a current source or current sink is thus required in each case in order to prevent an AC short circuit of the differential coupling element.
Such oscillators are also called feedback oscillators because of the feedback path. Also known are so-called reflection oscillators, for example from the publication “Optimizing MMIC Reflection-Type Oscillators,” 2004 IEEE MTT-S Digest, pp. 1341 ff. According to this document, such an oscillator has an active component that is connected to an AC ground through three impedances. In this context, two terminals are connected to ground in such a manner that a negative impedance is produced at the third terminal. A third impedance is connected to the AC ground there in order to set the resonant frequency.
As already described in WO 99/43079, when designing an oscillator it is always necessary to make compromises between requirements, one of which often can only be satisfied at the expense of another. A list of such requirements—which is not exhaustive—includes, for example, manufacturability in large quantities at the lowest possible costs, small space requirements for the oscillator circuit, low power consumption, a high signal-to-noise ratio, and low sensitivity to production-related variations in the circuit characteristics.
It is therefore an object of the present invention to provide an integrated differential oscillator circuit with an improved signal-to-noise ratio, a relatively wide tuning range and/or a relatively high quality, a relatively high efficiency and relatively small effects from production-related variations on the circuit characteristics.
This object is attained by an oscillator circuit of the aforementioned type in that the output is transformer-coupled to the input through a first inductor and a second inductor of the feedback network, wherein the output is connected to a first DC voltage through the first inductor and a first DC path, and the input is connected to a second DC voltage of the DC power supply through the second inductor and a second DC path.
As a result of the connection of the second inductor to the second DC reference voltage, the DC path required for deattenuating the resonant circuit and establishing the operating point of the amplifier circuit is routed through the second inductance to the amplifier circuit. As a general rule, inductors are implemented by metallic means, and have a negligibly small ohmic resistance as compared to bias elements of semiconductor material.
At such small ohmic resistance values, small differences in the resistance values, such as can arise from process variations in the production of integrated oscillator circuits, play only a secondary role. By contrast, in the customary DC connection of the amplifier circuit with the aid of resistors of semiconductor material or with the aid of active current sources or current sinks that contain transistors, process variations result in relatively large dispersions in the resistance values.
Moreover, the noise voltages u_r arising in the connecting lines are directly proportional to the value R of their resistances (u_r2=4kBTR, where kB=Boltzmann's constant and T=absolute temperature).
Because of the small resistance values of the inductors, the invention provides a low-noise DC connection of the amplifier circuit with a reduced range of effects due to process variations. This advantage is of great importance precisely because of the differential signal processing: Differential signal processing requires the best possible symmetry in the DC supply of the amplifier circuit. Deviations in the symmetry can lead to differences in the DC voltage at terminals of the differential input of the amplifier circuit. In the aforementioned prior art, such voltage differences can arise as a result of manufacturing-related variation of the properties of the two current sources, and can lead to different operating points of the transistors serving as amplifiers there. These transistors then are no longer driven in a precisely differential manner, producing adverse effects on the quality of the signal-to-noise ratio of the output signal of the oscillator circuit.
In contrast, as a result of the inventive connection of the input of the amplifier circuit to the second DC voltage of the DC power supply through the second inductor and the second DC path, a very low resistance of the DC power supply is achieved overall. Because of the differential design, separate DC path sections to the terminals of the differential input are still necessary. However, these sections are implemented through the extremely low resistance inductors. The total resistance of the DC power supply is thus dominated at the input side of the amplifier arrangement by components such as resistors or transistors of a current source of the DC power supply, which components are arranged in a circuit section that is common to both terminals of the differential input. As a result of these influences, asymmetries in the DC power supply of the amplifier circuit are avoided almost completely.
The transformer coupling permits a feedback of AC signals while it blocks DC currents. For configurations of the amplifier circuit with transistors, it thus permits the collector and emitter DC voltages or drain and source DC voltages required for transistor operation, in particular.
Since the tuning range, which is to say the bandwidth over which the resonant frequency can be tuned, is limited with increasing frequency by parasitic capacitances of the resonant circuit and/or the amplifier circuit, a reduction in the parasitic capacitances and thus an increase in the width of the frequency tuning range is produced as an additional great advantage of the transformer coupling. The reduction in parasitic capacitances achieved with the transformer coupling can be used either to achieve a maximum increase in the tuning range for constant quality, or to achieve a maximum increase in the quality for constant tuning range, or to achieve a simultaneous improvement of quality and tuning range to submaximal levels.
Further scope of applicability of the present invention will become apparent from the detailed description given hereinafter. However, it should be understood that the detailed description and specific examples, while indicating preferred embodiments of the invention, are given by way of illustration only, since various changes and modifications within the spirit and scope of the invention will become apparent to those skilled in the art from this detailed description.
The present invention will become more fully understood from the detailed description given hereinbelow and the accompanying drawings which are given by way of illustration only, and thus, are not limitive of the present invention, and wherein:
In this connection, like elements are labeled with like reference symbols in all figures. Specifically,
The oscillator circuit 16 shown in
The differential output 22.l, 22.r is transformer-coupled (magnetically) to the input 20.l, 20.r through the first inductor 26.l, 26.r and the second inductor 28.l, 28.r of the feedback network 24. In this regard, the output 22.l, 22.r is connected to a first DC voltage V1 of the DC power supply 32 through the first inductor 26.l, 26.r and a first DC path 36. The input 20.l, 20.r is connected to a second DC voltage V2 of the DC power supply 32 through the second inductor 28.l, 28.r and a second DC path 38.
The DC path 36 for the connection to the first DC voltage V1 is preferably connected to a center tap of the first inductor 26.l, 26.r. Similarly, the DC path 38 for the connection to the second DC voltage V2 is preferably connected to a center tap of the second inductor 28.l, 28.r. Because of the symmetry of the arrangement, each center tap then constitutes an AC ground 30 at which no AC component arises.
In this way, all voltages required for the operation of the oscillator circuit 16 can be supplied externally by existing components such as the inductors 26.l, 26.r, 28.l, 28.r, which themselves are connected to AC voltages that are in a sense static, which is to say to AC grounds 30 having different DC voltages.
When the embodiment 18.1 is used as an amplifier circuit 18 in
A signal at the collector of one of the two transistors 40, 42 is fed back to the emitter of the same transistor 40, 42 through the associated transformer coupling, by which means the transistor 40, 42 is modulated at its emitter. With such modulation, the signal at the collector as the output of the amplifier circuit 18 follows the input signal at the emitter with like phase. The phase condition for oscillation is met to this extent.
As an alternative to the embodiment 18.1 in
In this embodiment, the input 20.l (20.r) of the amplifier circuit 18.2 is connected to the base of the transistor 46 (44), while the output 22.l (22.r) is connected to the collector of the transistor 44 (46). In an application of the embodiment 18.2 as an amplifier circuit 18 from
With modulation of a transistor by an input signal at its base, the output signal at the collector of the same transistor always follows the input signal with a phase shift of π. Since the parallel resonant circuit having the first inductor 26.l 26.r and the capacitor lies between the collectors of the two transistors 44 and 46, and since an AC voltage arises across the parallel resonant circuit in the operation of the oscillator circuit 16, the parallel resonant circuit creates an additional phase shift of π between the two connected collectors. Thus, a phase shift of π arises at the collector of the transistor 44 relative to the collector of the transistor 46. Depending on the sign of the phase shift, the total phase shift between the base of the transistor 46 and the collector of the transistor 44 is thus either equal to 0 or equal to 2π. As a result of the cross-coupling 48, wherein the base of the left (right) transistor 44 (46) is connected to the right input 20.r (left input 20.l), the signal propagating from the collector of the transistor 44 to the base of the transistor 46 arrives there with an overall phase shift of zero or 2π relative to the input signal. The converse also applies, so that the phase prerequisite for oscillation is also met to this extent with the common-emitter configuration of the embodiment 18.2.
Although the above-described embodiments 18.1, 18.2 of amplifier circuits 18 have been discussed using bipolar NPN transistors 40, 42, 44, 46, it is understood that corresponding embodiments can also be built with bipolar PNP transistors or with unipolar transistors of the n-channel or p-channel type. In the embodiments with unipolar transistors, such transistors are used in (unipolar) common-gate, common-source or common-drain configurations analogous to the (bipolar) common-base, common-emitter or common-collector configuration.
In another embodiment, the values of the capacitor 34 in
The adjustable capacitor 34 is shown schematically in
With the adjustable capacitor 34, the oscillator circuit 16 constitutes, for example, a voltage-controlled oscillator VCO 16. For technical reasons, almost exclusively capacitive components 34 are used as drivable control components for frequency tuning in a VCO 16. In this context, the tuning range, which is to say the bandwidth over which the resonant frequency can be tuned, is limited with increasing frequency by parasitic capacitances of the resonant circuit and/or the amplifier circuit 18. This yields another great advantage of transformer coupling over the capacitive couplings otherwise used. With regard to the width of the frequency tuning range, the capacitive couplings count among the problematic parasitic capacitances.
The tuning range is proportional to the square root of the quotient of the difference of the maximum and minimum resonant circuit capacitances in the numerator, and the sum of the maximum and minimum resonant circuit capacitances in the denominator. In this regard, the value of the resonant circuit capacitance is comprised of the tunable and parasitic components or capacitances. In contrast to a capacitive coupling, the transformer coupling results in smaller values of the parasitic capacitances, since the coupling capacitances can be eliminated. As a rule, the values of the coupling capacitances are greater than the value of the tunable component 34 of the resonant circuit capacitance. Since the parasitic capacitances always drop out of the difference in the numerator, the width of the tuning range increases with decreasing parasitic capacitance values. Since the parasitic capacitance value is small with transformer coupling, the denominator is correspondingly small for transformer coupling, resulting in a correspondingly larger tuning range.
In addition, the quality factor Q of the resonant circuit depends on the quotient of the maximum capacitance in the numerator and the minimum capacitance in the denominator. The value of the quality factor drops with increasing quotient, first gradually and then more steeply. The steeply dropping quality factor thus limits the maximum tuning range.
As the size of the parasitic capacitances decreases, the quotient itself increases monotonically from a limit value of 1 to a value of the quotient that is determined only by the minimum and maximum values of the tunable capacitance component. The smaller the parasitic capacitances become, the larger the quotient becomes.
If one plots the quality factor Q as a function of the tuning range A, the qualitative result is the family of curves shown in
The conductor loops can be nearly circular, elliptical, or rectangular. In place of a pure rectangular, circular, or elliptical shape, other embodiments can also have conductor loops with piecewise straight segments in regular or irregular and convex or concave polygonal shapes and/or conductor loops with piecewise curved convex or concave segments or composite shapes composed of curved and straight segments.
In another embodiment, the frequency-selective network 24 composed of the resonant circuit inductors and capacitors has an additional capacitive coupling between the first inductor 26 and the second inductor 28, as is shown schematically in
In the additional embodiment in
With the exception of the abstract embodiment in
The invention being thus described, it will be obvious that the same may be varied in many ways. Such variations are not to be regarded as a departure from the spirit and scope of the invention, and all such modifications as would be obvious to one skilled in the art are to be included within the scope of the following claims.
Number | Date | Country | Kind |
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DE102006017188 | Apr 2006 | DE | national |