I. Field
The present disclosure relates generally to electronics, and more specifically to techniques for improving performance of a transmitter and a receiver.
II. Background
A wireless communication device such as a cellular phone typically includes a transmitter and a receiver to support bi-directional communication. The transmitter may condition and upconvert inphase (I) and quadrature (Q) output baseband signals with transmit local oscillator (LO) signals to obtain an output radio frequency (RF) signal that is more suitable for transmission via a wireless channel. The receiver may receive an input RF signal via the wireless channel and may downconvert the input RF signal with receive LO signals to obtain I and Q input baseband signals.
The transmitter typically includes an I branch and a Q branch to condition and quadrature upconvert the I and Q output baseband signals. The receiver also typically includes an I branch and a Q branch to quadrature downconvert the input RF signal and condition the I and Q input baseband signals. The I and Q branches may also be referred to as I and Q paths, etc. Ideally, for both the transmitter and the receiver, the I branch should be in quadrature (or 90° out of phase) with respect to the Q branch, and the two branches should have equal gain. However, I/Q imbalances typically exist between the I and Q branches and may include gain imbalance and/or phase error. Gain imbalance refers to error between the gains of the I and Q branches. Phase error refers to deviation/error from quadrature between the I and Q branches. The I/Q imbalances may degrade the performance of the transmitter and the receiver.
The word “exemplary” is used herein to mean “serving as an example, instance, or illustration.” Any design described herein as “exemplary” is not necessarily to be construed as preferred or advantageous over other designs.
I/Q imbalance estimation and compensation techniques are described herein. These techniques may be used for various electronics devices such as wireless communication devices, cellular phones, personal digital assistants (PDAs), handheld devices, wireless modems, laptop computers, cordless phones, Bluetooth devices, broadcast receivers, etc. For clarity, the use of the techniques for a wireless communication device is described below.
A receiver or a transmitter may be implemented with a super-heterodyne architecture or a direct-conversion architecture. In the super-heterodyne architecture, a signal is frequency converted between RF and baseband in multiple stages, e.g., from RF to an intermediate frequency (IF) in one stage, and then from IF to baseband in another stage for a receiver. In the direct-conversion architecture, a signal is frequency converted between RF and baseband in one stage. The super-heterodyne and direct-conversion architectures may use different circuit blocks and/or have different requirements. In the exemplary design shown in
In the receive path, an antenna 110 receives signals transmitted by base stations and/or other transmitter stations and provides an input RF signal, which is routed through a duplexer or switch 122 and provided to receiver 130. Within receiver 130, the input RF signal is amplified by a low noise amplifier (LNA) 132 and filtered by a filter 134 to obtain a filtered RF signal. A downconverter 136 receives the filtered RF signal and I and Q receive (RX) LO signals (IRX
In the transmit path, digital processor 170 processes data to be transmitted and provides I and Q output baseband signals (IBBout and QBBout) to transmitter 150. Within transmitter 150, lowpass filters 152a and 152b filter the I and Q output baseband signals, respectively, to remove undesired images caused by the prior digital-to-analog conversion. Amplifiers 154a and 154b amplify the signals from lowpass filters 152a and 152b, respectively, and provide I and Q amplified signals. An upconverter 156 receives the I and Q amplified signals and I and Q transmit (TX) LO signals (ITX
LO signal generator 126 generates the I and Q RX LO signals used by receiver 130 for frequency downconversion as well as the I and Q TX LO signals used by transmitter 150 for frequency upconversion. A phase locked loop (PLL) 124 receives timing information from digital processor 170 and generates control signals used to adjust the frequency and/or phase of the RX LO signals and the TX LO signals provided by LO signal generator 126.
Digital processor 170 may include various processing units for data transmission and reception and other functions. For example, digital processor 170 may include a digital signal processor (DSP), a reduced instruction set computer (RISC) processor, a central processing unit (CPU), etc. Memory 172 may store program codes and data for wireless device 100. Digital processor 170 and/or memory 172 may be implemented on one or more application specific integrated circuits (ASICs) and/or other ICs.
Receiver 130 may have I/Q imbalances, which may result from circuit blocks in the I branch (e.g., mixer 138a, lowpass filter 140a, and amplifier 142a) not matching circuit blocks in the Q branch (e.g., mixer 138b, lowpass filter 140b, and amplifier 142b). I/Q imbalances may result from gain imbalance between the I and Q branches and phase error between the I and Q branches. I/Q imbalances may also result from different frequency responses for the I and Q branches. In any case, I/Q imbalances may degrade the performance of receiver 130.
In an aspect, I/Q imbalances in receiver 130 may be estimated and compensated in order to improve the performance of the receiver. I/Q imbalances may be estimated for different frequencies to obtain frequency-dependent gain imbalances and phase errors between the I and Q branches of the receiver. I/Q imbalances may then be compensated across frequency, which may provide better performance than compensation at a single frequency.
Within digital processor 170, analog-to-digital converters (ADC) 212a and 212b receive and digitize the I and Q input baseband signals and provide I and Q input samples (IIN and QIN), respectively. An RX I/Q imbalance estimation unit 220 receives the I and Q input samples and estimates I/Q imbalances between the I and Q branches of receiver 130, as described below. An RX I/Q imbalance compensation unit 230 also receives the I and Q input samples and the measured I/Q imbalances from estimation unit 220. Compensation unit 230 processes the I and Q input samples to compensate for I/Q imbalances between the I and Q branches, as described below, and provides I and Q compensated input samples (ICIN and QCIN). The I and Q compensated input samples may be further processed (e.g., filtered, demodulated, decoded, etc.) to recover data sent in the input RF signal. Although not shown in
In an exemplary design, I/Q imbalances in receiver 130 (and also I/Q imbalances in transmitter 150) may be decomposed into (i) frequency-independent I/Q imbalance that is applicable across frequency and (ii) frequency-dependent I/Q imbalances that may vary across frequency. The frequency-independent and frequency-dependent I/Q imbalances may be estimated and compensated as described below.
In a first exemplary design of estimating I/Q imbalances in receiver 130, a continuous wave (CW) signal may be applied as the input RF signal. A CW signal is a periodic signal at a single frequency (ideally) but may include other spectral components (e.g., harmonics). The CW signal may be swept across frequency. In one design, the CW signal may be generated with a PLL and injected into receiver 130 via a switch. In another design, the CW signal may be provided by transmitter 150 via a loopback switch. In any case, I/Q imbalances in receiver 130 may be determined by examining the I and Q input samples from ADCs 212.
The CW signal may be expressed in discrete time as:
where n is a sample index,
k is a frequency index, which may also be referred to as a tone index,
fin,k is the frequency of the CW signal for tone k,
fsamp is the sampling rate of ADCs 212, and
Ideal I and Q RX LO signals may be expressed in discrete time as:
where fLO is the frequency of the I and Q RX LO signals, and
c(n) denotes a complex RX LO signal comprising the I and Q RX LO signals.
In an actual system, the CW signal and the LO signals are typically analog signals and are sampled after the signals are mixed and I/Q imbalances have affected the signals.
The I and Q RX LO signals may be set at a fixed frequency of fLO, and the CW signal may be varied in frequency. The frequency of the I and Q input baseband signals may then be given as:
f
k
=f
in,k
−f
LO, Eq (3)
where fk is the frequency of the I and Q input baseband signals corresponding to tone k.
The CW signal and the I and Q RX LO signals are typically not pure sinusoidal signals. Consequently, the I and Q input samples from ADCs 212 typically include odd harmonics such as 3rd and 5th harmonics. The I and Q input samples may be digitally filtered by estimation unit 220 to attenuate harmonics. The digitally filtered I and Q input samples may be used for I/Q imbalance estimation and may be referred to as simply I and Q samples.
For tone k, the I and Q samples used for I/Q imbalance estimation may be expressed as:
where β1 and βQ denote the phase errors of the I and Q RX LO signals, respectively,
In the exemplary design shown in equation set (4), I/Q imbalances in receiver 130 are approximated or modeled by a frequency-independent component and a frequency-dependent component. The frequency-independent component is modeled by phase errors βI and βQ and gain errors γI and γQ between the I and Q branches, which are not a function of tone k. The frequency-dependent component is modeled by gain vI,k and phase φI,k for the I branch and gain vQ,k and phase φQ,k for the Q branch, which are functions of tone k. The frequency-dependent component may include the effects of mixers 138, lowpass filters 140, amplifiers 142, and ADCs 212. The frequency responses of lowpass filters 140 may be a major contributor to the frequency-dependent component.
For each tone k, the root-mean-square (RMS) of the I and Q samples may be computed as follows:
where Nk is the number of I and Q samples used to compute the RMS,
xI,RMS denotes the RMS of the I samples for tone k, and
xQ,RMS denotes the RMS of the Q samples for tone k.
The RMS should be computed for an integer number of cycles and for a sufficient number of cycles of a baseband CW signal at tone k in order to obtain a more accurate RMS measurement. Nk may be selected to include a sufficient number of samples for a sufficient integer number of cycles at tone k.
A gain imbalance gerr,k at tone k may be determined by taking the ratio of the RMS of the I and Q samples, as follows:
A phase error θtotal,k at tone k may be obtained by taking the arcsin of the average of the products of normalized I and Q samples, as follows:
The gain imbalance gerr,k and the phase error θtotal,k may be determined for each of K different tones, where K may be any suitable value. The bandwidth of lowpass filters 140 may be dependent on the system bandwidth and may be denoted as fBW. The K tones may be spaced uniformly across±the filter bandwidth or at specific frequencies by stepping the CW signal from fin,1=fLO−fBW to fin,K=fLO+fBW. The
K tones may be mirrored so that for each tone at a frequency of +fx there is a corresponding tone at a frequency of −fx.
The phase error θtotal,k for each tone k may be decomposed into (i) a frequency-independent phase error θfreq
The frequency-dependent phase error θk for each tone k may be expressed as:
θk=θtotal,k−θfreq
The I/Q imbalance at each tone k may then be expressed as:
H
k
=g
err,k
·e
jθ
, Eq (10)
where Hk denotes the complex I/Q imbalance at tone k. Frequency independent gain imbalance is included in gerr,k and does not need to be decomposed and compensated separately.
In a second exemplary design of estimating I/Q imbalances in receiver 130, the input RF signal may be used directly to estimate I/Q imbalances. This design may avoid the need to generate and inject a CW signal as the input RF signal. This design may also enable I/Q imbalance estimation without the need for a preamble in the input RF signal.
As shown in
The I and Q input samples from ADCs 212, with the I and Q RX LO signals being offset from the input RF signal, may be digitally filtered by estimation unit 220 with a Hamming window, a Hanning window, a Kaiser-Bessel window, a Gaussian window, or some other window. The window may attenuate side lobes resulting from the signal being “chopped” in the time domain into blocks of samples for subsequent processing. The digitally filtered I and Q input samples may be referred to as simply I and Q samples and may be expressed as:
where r(n) denotes the input RF signal,
hI(m) denotes the impulse response of the I branch of receiver 130,
hQ(m) denotes the impulse response of the Q branch of receiver 130,
yI(n) denotes an I sample at sample index n,
yQ(n) denotes a Q sample at sample index n, and
denotes a convolution operation.
Equation set (11) assumes that there is no side lobe, i.e., the signal has a sharp cutoff.
The I samples may be transformed to the frequency domain with an N-point fast Fourier transform (FFT) to obtain I symbols for N frequency bins (or simply, bins). Similarly, the Q samples may be transformed with an N-point FFT to obtain Q symbols for N frequency bins. The I samples have real values whereas the I symbols have complex values. Similarly, the Q samples have real values whereas the Q symbols have complex values. For clarity, in much of the description herein, a sample refers to a time-domain value whereas a symbol refers to a frequency-domain value. N may be selected to obtain the desired frequency resolution and may be equal to 128, 256, 512, 1024, 2048, etc. The I and Q symbols may be expressed as:
where γI, γQ, βI and βQ are described above,
vI(k) denotes the gain and φI(k) denotes the phase of the I branch at bin k,
vQ(k) denotes the gain and φQ(k) denotes the phase of the Q branch at bin k,
R(k±kos) denotes the RF signal at bin k±kos,
kos denotes the bin corresponding to the RX LO frequency offset fos,
YI(k) denotes an I symbol for bin k, and
YQ(k) denotes a Q symbol for bin k.
A gain imbalance gerr,os(k) for bin k with frequency offset fos may be determined by taking the ratio of the magnitudes of the I and Q symbols for bin k, as follows:
A phase error θtotal,os(k) for bin k with LO frequency offset fos may be obtained by taking the difference of the angles of the I and Q symbols for bin k, as follows:
where Re{ } denotes the real part and Im{ } denotes the imaginary part.
For the second exemplary design, the frequency of the I and Q RX LO signals may be offset below the center frequency of the input RF signal and may be given as fLO=fin−fos. I and Q samples obtained with low-side frequency offset may be processed as described above to obtain gain imbalance gerr,low(k) and phase error θtotal,low(k) for each bin k within a frequency range of interest. This frequency range may correspond to±the filter bandwidth fBW.
The frequency of the I and Q RX LO signals may also be offset above the center frequency of the input RF signal and may be given as fLO=fin+fos. I and Q samples obtained with high-side frequency offset may be processed as described above to obtain gain imbalance gerr,high(k) gerr,high(k) and phase error θtotal,high(k) for each bin k within the frequency range of interest.
A gain imbalance gerr(k) for bin k may be determined by averaging the gain imbalances for the low-side and high-side frequency offsets, as follows:
A phase error θtotal(k) for bin k may be obtained as follows:
The phase error θtotal(k) may be offset by ±π/2.
The gain imbalance gerr(k) and the phase error θtotal(k) may be determined for each of K different bins within the frequency range of interest, where K may be any suitable value. K may be dependent on the sampling rate fsamp, the filter bandwidth fBW, the FFT size N, etc. The K bins may also cover mirrored±frequencies, as described above.
The phase error θtotal(k) for each bin k may be decomposed into (i) a frequency-independent phase error θfreq
The frequency-dependent phase error θ(k) for each bin k may be expressed as:
θ(k)=θtotal(k)−θfreq
The I/Q imbalance at each bin k may then be expressed as:
H(k)=gerr(k)·ejθ(k), Eq (19)
where H(k) denotes the complex I/Q imbalance for bin k.
To improve accuracy of I/Q imbalance estimation, the I/Q imbalance may be estimated as described above based on a set of I and Q samples. The process may be repeated for one or more additional sets of I and Q samples. The I/Q imbalance estimates obtained with different sets of I and Q samples may be averaged to obtain a final I/Q imbalance estimate having better accuracy.
In one exemplary design of compensating for I/Q imbalances in receiver 130, a digital filter may be used to compensate for frequency-dependent I/Q imbalances between the I and Q branches of receiver 130. The digital filter may be a finite impulse response (FIR) filter, an infinite impulse response (IIR) filter, some other types of filter, or a combination thereof. Multipliers with scalars may be used to compensate for frequency-independent I/Q imbalances between the I and Q branches of receiver 130.
In one exemplary design, an L-tap FIR filter may be used to compensate for frequency-dependent I/Q imbalances, where L may be equal to 3, 4, 5, 6, 7, 8, 9 or some other suitable value. The number of taps (L) may be selected based on frequency variation of I/Q imbalance and the desired accuracy in I/Q imbalance compensation. The L coefficients of the FIR filter may be determined by finding the least squares best-fit for the measured I/Q imbalance. For clarity, the description below is for determining the coefficients of the FIR filter for the measured I/Q imbalance Hk, for k=0, . . . , K−1, for the first exemplary design.
A K×L matrix F may be defined as follows:
where τl=l·τ and τ is one sample period for the FIR filter, for l=0, . . . , L−1, and
ωk is the frequency corresponding to tone k or bin k, for k=0, . . . , K−1.
The L coefficients for the FIR filter may then be determined as follows:
where αl is the coefficient for the l-th tap of the FIR filter.
The FIR filter may be applied to either the I branch or the Q branch of receiver 130, depending on how the gain imbalances are computed. In particular, the FIR filter may be applied to the Q branch if the gain imbalances are computed based on xI,RMS/xQ,RMS or |YI(k)|/|YQ(k)|, as described above. The FIR filter may be applied to the I branch if the gain imbalances are computed based on xQ,RMS/xI,RMS or |YQ(k)|/|YI(k)|. The phase may be flipped depending on which branch is used as the reference.
In an exemplary design, a scalar c to compensate for frequency-independent I/Q imbalance may be determined as follows:
L coefficients of a FIR filter for the measured I/Q imbalance H(k), for k=0, . . . , K−1, for the second exemplary design may be determined in similar manner. In this case, the vector in the right hand side of equation (21) may be replaced with H(0) through H(K−1). The computation shown in equation (21) may then be performed to obtain the coefficients of the FIR filter for the second exemplary design.
In the exemplary design shown in
Within compensation unit 530, a multiplier 532a multiplies the I filtered samples with scalar c and provides scaled I filtered samples. A multiplier 532b multiplies the Q corrected samples with scalar c and provides scaled Q corrected samples. A summer 534a sums the I filtered samples with the scaled Q corrected samples and provides I compensated input samples (ICIN). A summer 534b sums the Q corrected samples with the scaled I filtered samples and provides Q compensated input samples (QCIN).
I/Q imbalance may be estimated once or periodically in order to track variation over time. I/Q imbalance may be compensated continuously.
Gain imbalance between the I and Q branches for each tone may be determined based on the I and Q samples for the tone (block 614). In one exemplary design, the gain imbalance between the I and Q branches for each tone may be determined based on a ratio of an RMS of the I samples for the tone and an RMS of the Q samples for the tone, e.g., as shown in equation (6). The phase error between the I and Q branches for each tone may be determined based on the I and Q samples for the tone (block 616). In one exemplary design, the phase error for each tone may be determined based on an average of the products of normalized I samples and normalized Q samples for the tone, e.g., as shown in equation (7).
A frequency-independent phase error may be determined based on an average of the phase errors for the plurality of tones, e.g., as shown in equation (8) (block 618). A frequency-dependent phase error for each tone may be determined based on the phase error for the tone and the frequency-independent phase error, e.g., as shown in equation (9) (block 620).
The first set of I and Q samples may be transformed to the frequency domain to obtain a first set of I and Q symbols for a plurality of frequency bins (block 716). The second set of I and Q samples may also be transformed to the frequency domain to obtain a second set of I and Q symbols for the plurality of frequency bins (block 718).
Gain imbalance between the I and Q branches may be determined based on the first and second sets of I and Q symbols (block 720). In an exemplary design, gain imbalance between the I and Q branches for each of multiple frequency bins may be determined based on a ratio of the magnitude of an I symbol for the frequency bin to the magnitude of a Q symbol for the frequency bin, e.g., as shown in equation (13).
Phase error between the I and Q branches may be determined based on the first and second sets of I and Q symbols (block 722). In an exemplary design, the phase error between the I and Q branches for each of multiple frequency bins may be determined based on an angle of an I symbol for the frequency bin and an angle of a Q symbol for the frequency bin, e.g., as shown in equation (14). The phase error for each frequency bin corresponding to positive frequency may be determined based on I and Q symbols for that frequency bin in the first set of I and Q symbols, e.g., as shown in equation (16). The phase error for each frequency bin corresponding to negative frequency may be determined based on I and Q symbols for that frequency bin in the second set of I and Q symbols, e.g., as also shown in equation (16).
A frequency-independent phase error may be determined based on an average of phase errors for the multiple frequency bins (block 724). A frequency-dependent phase error for each of the multiple frequency bins may be determined based on the phase error for the frequency bin and the frequency-independent phase error (block 726).
The first and third digital filters may be of the same type (e.g., IIR filters) and may have the same frequency response. The second digital filter may be an FIR filter having a frequency response determined based on I/Q imbalances between the first and second branches of the receiver. The frequency response of the second digital filter may be determined based on gain imbalances and phase errors between the first and second branches for multiple frequencies, e.g., multiple tones or multiple frequency bins.
First compensated samples (e.g., QCIN in
The first corrected samples for the first branch and the second filtered samples for the second branch (e.g., QCOR and IFIL in
Referring back to
In another aspect, I/Q imbalances in transmitter 150 may be estimated and compensated in order to improve the performance of the transmitter. I/Q imbalances may be estimated for different frequencies to obtain frequency-dependent gain imbalances and phase errors between the I and Q branches of transmitter 150. I/Q imbalances may then be compensated across frequency, which may provide better performance than I/Q compensation at a single frequency.
In the exemplary design shown in
ADCs 212a and 212b digitize the I and Q input baseband signals and provide I and Q input samples, respectively. A TX I/Q imbalance estimation unit 240 receives the I and Q input samples and the I and Q output samples, estimates I/Q imbalances between the I and Q branches of transmitter 150 as described below, and provides frequency responses of the I and Q branches of transmitter 150. TX I/Q imbalance compensation unit 250 receives the frequency responses of the I and Q branches from estimation unit 220 and determines coefficients and scalars to compensate for I/Q imbalances in transmitter 150.
In a first exemplary design of estimating I/Q imbalances in transmitter 150, a reference signal may be applied as the I and Q output samples. The reference signal is a known signal and may be a pseudo-random noise signal, a signal having a constant envelop and a flat frequency response, etc. Receiver 130 and transmitter 150 may operate in a loopback configuration, as shown in
The I and Q input samples from ADCs 212 in the loopback configuration may be expressed as:
{circumflex over (x)}I(n)=xI(n)hI(m)+b·xQ(n)hQ(m), and Eq (23a)
{circumflex over (x)}Q(n)=xQ(n)hQ(m)+a·xI(n)hI(m), Eq (23b)
where xI(n) and xQ(n) denote the I and Q output samples, respectively,
hI(m) denotes the impulse response of the I branch of transmitter 150,
hQ(m) denotes the impulse response of the Q branch of transmitter 150,
a and b are scalars modeling frequency-independent I/Q phase imbalances, and
{circumflex over (x)}I(n) and {circumflex over (x)}Q(n) denote the I and Q input samples, respectively.
A block of N I input samples may be transformed to the frequency domain with an N-point FFT to obtain a block of N I input symbols for N bins. Similarly, a block of N Q input samples may be transformed with an N-point FFT to obtain a block of N Q input symbols for N frequency bins. The I and Q input symbols may be expressed as:
where {circumflex over (X)}I(k) and {circumflex over (X)}Q(k) denote the FFTs of {circumflex over (x)}I(n) and {circumflex over (x)}Q(n), respectively,
HI(k) and HQ(k) denote the FFTs of hI(m) and hQ(m), respectively, and
XI(k) and XQ(k) denote the FFTs of xI(n) and xQ(n), respectively.
Multiple (R) blocks of I and Q input samples may be transformed to obtain R blocks of I and Q input symbols. For each bin k, the I input symbols in the R blocks may be stacked. For each bin k, the Q input symbols in the R blocks may also be stacked. The stacked I and Q symbols for each bin k may be expressed as:
where XI,r(k) and XQ,r(k) denote I and Q output symbols in the r-th block, and
{circumflex over (X)}I,r(k) and {circumflex over (X)}Q,r(k) denote I and Q input symbols in the r-th block.
In equation set (25), XI,r(k) and XQ,r(k) output symbols may be obtained from the known I and Q output samples xI(n) and xQ(n) provided to DACs 252 on the transmitter side. In particular, R blocks of I output samples may be transformed to obtain R blocks of I output symbols, and R blocks of Q output samples may be transformed to obtain R blocks of Q output symbols. The output samples may be reference output samples, arbitrary samples, transmit samples, etc. {circumflex over (X)}I,r(k) and {circumflex over (X)}Q,r(k) input symbols may be obtained from the I and Q input samples provided by ADCs 212 on the receiver side. Acquisition may be performed to time-align the I and Q input samples with the I and Q output samples.
The frequency responses of the I and Q branches of transmitter 150 for each bin k may be determined as follows:
Both HI(k) and a·HI(k) as well as HQ(k) and b·HQ(k) may be obtained from equation set (26). The scalars a and b may be determined as follows:
In one exemplary design of compensating for I/Q imbalances in transmitter 150, separate FIR filters may be used to compensate for the frequency responses of the I and Q branches of transmitter 150. Multipliers with scalars may be used to compensate for frequency-independent I/Q imbalance between the I and Q branches of transmitter 150.
The coefficients for the FIR filters for the I and Q branches of transmitter 150 may be determined as follows:
where pl is the coefficient for the l-th tap of the FIR filter for the I branch, and
ql is the coefficient for the l-th tap of the FIR filter for the Q branch.
The FIR filters for the I and Q branches may have the same length or different lengths. The FIR filters for TX I/Q imbalance compensation may have the same or different lengths as the FIR filters for RX I/Q imbalance compensation.
In the exemplary design shown in
Within compensation unit 1030, an IIR filter 1032a receives and filters the first I samples. FIR filter 1034a further filters the output samples from IIR filter 1032a and provides I compensated output samples (ICOUT). An IIR filter 1032b receives and filters the first Q samples. FIR filter 1034b further filters the output samples from IIR filter 1032b and provides Q compensated output samples (QCOUT). IIR filters 1032a and 1032b may be noise reduction filters having a wider bandwidth (e.g., 1.5 fBW).
Frequency responses of the I and Q branches of the transmitter may be determined based on the I and Q input symbols and the I and Q output symbols (block 1118). In one exemplary design, a first complex gain (e.g., HI(k) and a·HI(k)) for the I branch and a second complex gain (e.g., HQ(k) and b·HQ(k)) for the Q branch for each of multiple frequency bins may be determined based on I and Q input symbols for the frequency bin in the multiple blocks of I and Q input symbols, e.g., as shown in equation set (26). The frequency response of the I branch may comprise first complex gains for the I branch for the multiple frequency bins. The frequency response of the Q branch may comprise second complex gains for the Q branch for the multiple frequency bins. The frequency response may also comprise phase information for the I and Q branches. The frequency responses of the I and Q branches may be used to compensate for frequency-dependent I/Q imbalances between the I and Q branches of the transmitter.
A first scalar for the I branch and a second scalar for the Q branch may also be determined based on the I and Q input symbols and the I and Q output symbols (block 1120). The first and second scalars may be used to compensate for frequency-independent I/Q imbalance between the I and Q branches of the transmitter
The first I samples for the I branch may be filtered with a first digital filter (e.g., FIR filter 1034a in
In one exemplary design, the frequency response of the I branch of the transmitter may be determined and used to determine the coefficients of the first digital filter. The frequency response of the Q branch of the transmitter may also be determined and used to determine the coefficients of the second digital filter. The frequency responses of the first and second digital filters may thus compensate for frequency-dependent I/Q imbalance between the I and Q branches of the transmitter. The first and second scalars may be selected to compensate for frequency-independent I/Q imbalance between the I and Q branches of the transmitter.
The I/Q imbalance estimation and compensation techniques described herein may be used to estimate and compensate for frequency-dependent I/Q imbalances, which may arise due to mismatched analog lowpass filters and other circuit blocks. The techniques may enable a receiver and a transmitter to achieve higher image rejection across an operating band, which may improve performance.
Those of skill would further appreciate that the various illustrative logical blocks, modules, circuits, and algorithm steps described in connection with the disclosure herein may be implemented as electronic hardware, computer software, or combinations of both. To clearly illustrate this interchangeability of hardware and software, various illustrative components, blocks, modules, circuits, and steps have been described above generally in terms of their functionality. Whether such functionality is implemented as hardware or software depends upon the particular application and design constraints imposed on the overall system. Skilled artisans may implement the described functionality in varying ways for each particular application, but such implementation decisions should not be interpreted as causing a departure from the scope of the present disclosure.
The various illustrative logical blocks, modules, and circuits described in connection with the disclosure herein may be implemented or performed with a general-purpose processor, a DSP, an ASIC, a field programmable gate array (FPGA) or other programmable logic device, discrete gate or transistor logic, discrete hardware components, or any combination thereof designed to perform the functions described herein. A general-purpose processor may be a microprocessor, but in the alternative, the processor may be any conventional processor, controller, microcontroller, or state machine. A processor may also be implemented as a combination of computing devices, e.g., a combination of a DSP and a microprocessor, a plurality of microprocessors, one or more microprocessors in conjunction with a DSP core, or any other such configuration.
The steps of a method or algorithm described in connection with the disclosure herein may be embodied directly in hardware, in a software module executed by a processor, or in a combination of the two. A software module may reside in RAM memory, flash memory, ROM memory, EPROM memory, EEPROM memory, registers, hard disk, a removable disk, a CD-ROM, or any other form of storage medium known in the art. An exemplary storage medium is coupled to the processor such that the processor can read information from, and write information to, the storage medium. In the alternative, the storage medium may be integral to the processor. The processor and the storage medium may reside in an ASIC. The ASIC may reside in a user terminal. In the alternative, the processor and the storage medium may reside as discrete components in a user terminal.
In one or more exemplary designs, the functions described may be implemented in hardware, software, firmware, or any combination thereof. If implemented in software, the functions may be stored on or transmitted over as one or more instructions or code on a computer-readable medium. Computer-readable media includes both computer storage media and communication media including any medium that facilitates transfer of a computer program from one place to another. A storage media may be any available media that can be accessed by a general purpose or special purpose computer. By way of example, and not limitation, such computer-readable media can comprise RAM, ROM, EEPROM, CD-ROM or other optical disk storage, magnetic disk storage or other magnetic storage devices, or any other medium that can be used to carry or store desired program code means in the form of instructions or data structures and that can be accessed by a general-purpose or special-purpose computer, or a general-purpose or special-purpose processor. Also, any connection is properly termed a computer-readable medium. For example, if the software is transmitted from a website, server, or other remote source using a coaxial cable, fiber optic cable, twisted pair, digital subscriber line (DSL), or wireless technologies such as infrared, radio, and microwave, then the coaxial cable, fiber optic cable, twisted pair, DSL, or wireless technologies such as infrared, radio, and microwave are included in the definition of medium. Disk and disc, as used herein, includes compact disc (CD), laser disc, optical disc, digital versatile disc (DVD), floppy disk and blu-ray disc where disks usually reproduce data magnetically, while discs reproduce data optically with lasers. Combinations of the above should also be included within the scope of computer-readable media.
The previous description of the disclosure is provided to enable any person skilled in the art to make or use the disclosure. Various modifications to the disclosure will be readily apparent to those skilled in the art, and the generic principles defined herein may be applied to other variations without departing from the scope of the disclosure. Thus, the disclosure is not intended to be limited to the examples and designs described herein but is to be accorded the widest scope consistent with the principles and novel features disclosed herein.