1. Field of the Invention
The present invention relates generally to phase-locked loops (PLLs).
2. Description of Related Art
Phase-locked loops are widely used in electronic devices, such as computers, telecommunications equipment and radio. However, known PLLs have been insufficiently noise insensitive.
The PFD 101 may compare the feedback frequency ωfb with the reference frequency ωref. When ωfb is lower than the reference frequency ωref, the PFD 101 may output switching signals PU and PD to the charge pump 102, closing a switch 1021 and keeping a switch 1022 open, so as to charge a charge storage device C1 in the loop filter 103. Consequently, the control voltage Vctrl at the output of the loop filter 103 is up, the control current IC supplied to the ICO 1043 is up, and the output frequency ωout is up until it equals N ωref.
When ω is higher than the reference frequency ωref, the PFD 101 may generate switching signals PU and PD to open the switch 1021 and close the switch 1022, so that the charge storage device C1 in the loop filter 103 may discharge via the switch 1022. Consequently, the control voltage Vctrl at the output of the loop filter 103 is down, the control current IC is down, and the output frequency ωout is down, until ωout=N ωref.
When ωfb equals ωref, the PFD 101 may keep both switches 1021 and 1022 closed to maintain the relationship.
The pole position and zero position of the PLL 100 are:
So the ratio between the pole and zero position is:
The gain bandwidth of the PLL 100 is:
For the PLL 100, the output phase noise at the output frequency ωout contributed by R1 is:
wherein K is the Boltzmann constant, T is a temperature value, and s=jω, which is a variable in frequency domain.
One limitation of the known PLL 100 is its high frequency tuning gain (KVCO), which is the frequency/voltage gain of the VCO. The high frequency tuning gain, together with a wide loop bandwidth used to suppress phase noise, may make the PLL more sensitive to the noise from the PFD, the charge pump and the loop filter. Therefore, it may be desirable to provide a PLL which may have a decreased frequency tuning gain KVCO and the wide loop bandwidth.
Another limitation of the known PLL 100 is that the loop filter 103 may occupy a large chip area when being integrated on a chip. One known solution uses an active loop filter, but the active device may bring additional noise. Another known solution uses a passive feed forward loop filter with noiseless resistor multiplication, but it may increase sensitivity to switching glitches of the charge pump. Therefore, it may be desirable to provide a PLL with a loop filter which has high chip area efficiency but low noise and switching glitch sensitivity.
A voltage-controlled oscillator (VCO), comprising: a voltage supply; a voltage to current converter for converting a voltage signal from the voltage supply into a control current, the voltage to current converter comprising a first voltage to current converter and a second voltage to current converter coupled in parallel to produce the control current; and a current-controlled oscillator (ICO), generating an output frequency in response to the control current.
A phase-locked loop (PLL), comprising: a phase and frequency detector (PFD) for comparing a feedback frequency and a reference frequency and generating a first switching signal and a second switching signal; a first charge pump, comprising a first current source coupled in series with a first switch, a second switch and a second current source, wherein the first switch is controlled by the first switching signal, the second switch is controlled by the second switching signal and the first current source provides a current ICP; a loop filter, comprising a first branch and a second branch coupled in parallel, wherein the first branch comprises a resistance device coupled in series with a first charge storage device, and the second branch comprises a second charge storage device; and a voltage controlled oscillator (VCO) providing an output frequency, wherein the output frequency is sent to the PFD as the feedback frequency. The VCO comprise: a voltage supply; a voltage to current converter for converting a voltage signal from the voltage supply into a control current, the voltage to current converter comprising a first voltage to current converter and a second voltage to current converter coupled in parallel to produce the control current; and a current-controlled oscillator (ICO), generating the output frequency in response to the control current.
A capacitor multiplication structure may be used in the PLL to increase the capacitance of the first charge storage device without increasing its size. The PLL may further comprise a second charge pump coupled in parallel with the first charge pump, wherein the second charge pump comprises a third current source, a third switch, a fourth switch and a fourth current source coupled in series, to charge or discharge the junction of the resistance device and the first charge storage device. The third current source provides a current α*ICP to decrease voltage variations over the first charge storage device, and wherein α<1. The third switch is turned on and off approximately simultaneously with the first switch, but in the opposite direction. The fourth switch is turned on and off approximately simultaneously with the second switch, but in the opposite direction.
A method for controlling a voltage-controlled oscillator (VCO), comprising: converting a voltage signal into a first control current with a first voltage to current converter; converting a voltage signal from the voltage supply into a second control current with a second voltage to current converter; combining the first control current and the second control current into a control current; and generating an output frequency in response to the control current with a current-controlled oscillator (ICO).
Embodiments are described herein with reference to the accompanying drawings, similar reference numbers being used to indicate functionally similar elements.
The following discussion describes a PLL with an oscillator structure having a low frequency tuning gain KVCO and a passive loop filter structure with noiseless capacitor multiplication.
A VCO 204 may include a voltage supply such as a high PSR regulator 2041, a voltage to current converter 2042 and an ICO 2043. The voltage to current converter 2042 may include two voltage to current converters, 2042A and 2042B, coupled in parallel between the voltage supply 2041 and the ICO 2043. The converter 2042A may be controlled by a control voltage Vctrl
In one embodiment, gm
gm=gm
gm
gm
According to equation (7), when β is considerably greater than 1, the trans-conductance gm
When Vctrl
VC=Vctrl
Assuming that KVCO′ is the tuning gain from Vctrl
When the trans-conductance of the VCO 204 is similar to the trans-conductance of the VCO 104 in the known PLL 100, by using two converters 2042A and 2042B and controlling the converter 2042B, whose trans-conductance is β times the trans-conductance of 2042A, with the control voltage Vctrl
The charge pump 302 may have a current source 3021, switches 3022 and 3023, and a current source 3024 coupled in series between a fixed voltage AVDD and a fixed voltage P. Switches 3022 and 3023 may be turned on and off by switching signals PU and PD from the PFD 301 respectively, and may be, e.g., transistors. The current sources 3021 and 3024 may provide a current ICP.
In the loop filter 303, a charge storage device C1 and a resistance device R1 may be coupled in series between the output of the charge pump 302 and the fixed voltage P, and a charge storage device C2 may be coupled between the output of the charge pump 302 and the fixed voltage P as well, in parallel with the circuit branch including the charge storage device C1 and the resistance device R1. The charge storage devices C1 and C2 may be, e.g., capacitors, and the resistance device R1 may be, e.g., a resistor. A voltage to current converter 3042A may take its control voltage Vctrl
The current source 3021 may charge the charge storage device C1 via the switch 3022 and the resistance device R1 to raise the control voltages Vctrl
As discussed above with reference to
In the circuit shown in
gm
In particular, the gate of MNA may receive the control voltage Vctrl
In the circuit shown in
gm
In particular, the gate of MPA may receive the control voltage Vctrl
In the circuit of
In particular, the gate of MNA may receive the control voltage Vctrl
The current mirror 7044 may have P-type FETs MPP1 and MPP2. The junction of the gates of MPP1 and MPP2 and the drain of MPP1 may be coupled to the junction of the drains of MNA and MNB. The source of MPP1 may be coupled to the voltage supply 7041 via a resistor Rc, and the source of MPP2 may be coupled to the voltage supply 7041 via a resistor Rd. In one embodiment, Rd=Rc/m. The drain of MPP2 may be coupled to an ICO 7043.
The control voltage Vctrl
Some properties of the PLL 300 are as follows:
wherein “∥” is a functor which means:
When IA=Vctrl
So the open loop transfer function may be:
From equation (15), the pole and zero position may be:
The gain bandwidth may be:
As the impedance of C1 is far smaller than that of C2, the thermal noise contributed by R1 at the junction between R1 and C1 is negligible compared to that at the junction R1 and C2. As a result, the output phase noise contributed by R1 at the output frequency ωout of the PLL 300 may be:
where φn2 is the output phase noise contributed by R1 at the output frequency ωout of the PLL 100.
Compared to equation (3), equation (18) indicates that the pole over zero ratio of the PLL 300 is 1/(1+β) times of that of the known PLL 100. This will make the PLL loop unstable. In addition, equation 20 indicates that, due to the frequency tuning gain KVCO is reduced to 1/(1+β) times, the phase noise contribute by R1 is reduced to 1/(1+β)2 times.
One way to improve the loop stability of the PLL 300 may be to increase the value of C1 to its (1+β) times. However, this may make the die size consumption unacceptable, since C1 is a big capacitor.
Another way to improve the loop stability of the PLL 300 may be to decrease the value of C2 to its 1/(1+β) and increase the value of R1 to its (1+β) times. Since the resistor may become R1*(1+β), which will contribute 4*K*T*R1*(1+β) to the noise, the noise reduction benefit depicted by equation 20 may be substituted with equation 23. Equation 23 also indicates that, in the PLL 300, the phase noise contributed by the filter resistor R1 may be improved to 1/(1+β) times of that in the PLL 100. Properties of such a PLL may be:
One way to both improve the loop stability and further reduce the output phase noise of the PLL 300 is to use capacitor multiplication.
When ωfb is lower than the reference frequency ωref, the PFD 301 may output switching signals PU and PD to the charge pump 302 and the replica charge pump 1002, closing switches 3022 and 10023 and keeping switches 3023 and 10022 open. When f<1/(C1R1), since 1/(sC2)>>(R1+1/(sC1), most of the current ICP coming from the current source 3021 in the charge pump 302 may go through C1. At the same time, since 1/(sC1)<<(R1+1/sC2)), most of the current from the current source 10024 in the replica charge pump 1002, which is in the opposite direction of the current ICP coming from the current source 3021 and is αICP, may go through C1 as well. Thus, the actual current going through C1 is ICP*(1−α), and the actual voltage drop over C1 is about ICP*(1−α)/(s*C1), which means that the equivalent value of C1 may be amplified by 1/(1−α).
When α=⅞, for example, for each charge unit charged to C1 from the current source 3021 via the switch 3022 and R1, ⅞ of the charge unit may be drawn from C1 by the current source 10024 via R1 and the switch 10023. Thus, the value of C1 may be amplified by 8 times, without changing the size of C1 and reducing chip area efficiency, and the variation of Vctrl
When ωfb is higher than the reference frequency ωref, the PFD 301 may send switching signals PU and PD to the charge pump 302 and 1002, keeping switches 3022 and 10023 open while closing switches 3023 and 10022. Consequently, C1 may discharge via R1 and the switch 3023 and the discharge current is ICP. At the same time, the current source 10022 may charge C1 via the switch 10022 and R1, and the charging current may be α*ICP.
When ωfb equals ωref, the PFD 301 may keep the switches 3022 and 3023 in the charge pump 302 and switches 10022 and 10023 in the replica charge pump closed to maintain the relationship.
The frequency tuning gain of the PLL 1000 may be similar to that of PLL 300. However, since the variation of Vctrl
Properties of the PLL 1000 may be:
When IA=Vctrl
then:
The open loop transfer function may be:
From equation (27), especially when α=β/(β+1), the pole and zero position may be:
The output phase noise contributed by R1 may be:
Equation (30) may indicate that the ratio of pole and zero position of the PLL 1000 in
Equation (31) may indicate that the output phase noise contributed by R1 in the PLL 1000 may be 1/(1+β) times of that in the PLL 300 (expressed by equation (19)), and 1/(1+β)2 times of that in the PLL 100.
In sum, PLL 1000 and PLL 300 may reduce the frequency tuning gain to about 1/(1+β) of that of the known PLL 100. PLL 1000 may reduce the output noise contributed by R1 to 1/(1+β) times of that in the PLL 300, and 1/(1+β)2 times of that in the PLL 100. In addition, with the capacitor multiplication provided by the replica charge pump, the PLL 1000 may achieve similar loop stability of that of PLL 100 without increasing the size of C1, C2 or R1. Table 1 is an exemplary performance comparison between a known PLL 100 and a PLL 1000 according to one embodiment.
Several features and aspects have been illustrated and described in detail with reference to particular embodiments by way of example only, and not by way of limitation. Alternative implementations and various modifications to the disclosed embodiments are within the scope and contemplation of the present disclosure. For example, bipolar junction transistors (BJTs) or junction gate field-effect transistors (JFETs) may be used to replace the FETs in the embodiments. Therefore, it is intended that the invention be considered as limited only by the scope of the appended claims.
This application claims the benefit of priority to previously filed U.S. provisional patent application Ser. No. 61/078,962, filed Jul. 8, 2008, entitled LOW-JITTER, RING BASED PHASE LOCKED LOOP WITH A VOLTAGE CONTROLLED OSCILLATOR STRUCTURE OF LOW FREQUENCY TUNING GAIN AND A LOOP FILTER STRUCTURE OF HIGH AREA EFFICIENCY. That provisional application is hereby incorporated by reference in its entirety.
Number | Name | Date | Kind |
---|---|---|---|
5691669 | Tsai et al. | Nov 1997 | A |
7812650 | Song et al. | Oct 2010 | B2 |
20040095188 | Puma et al. | May 2004 | A1 |
20080042759 | Watanabe | Feb 2008 | A1 |
Number | Date | Country | |
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61078962 | Jul 2008 | US |