Low-Noise Amplifier Circuit

Abstract

The low-noise amplifier circuit exhibits reduced noise.

Description

FIELD OF THE INVENTION

The invention relates to a low-noise amplifier circuit. A circuit and a mobile comprising such low-noise amplifier circuit are also proposed. A use for amplifying signals and a method for reducing the noise of a low-noise amplifier circuit are further considered.

BACKGROUND OF THE INVENTION

A low-noise amplifier circuit is an electronic amplifier used to amplify very weak signals. Weak signals may, for instance, be captured by a detection device such as an antenna. The low-noise amplifier circuit is often named after its acronym LNA circuit. For instance, LNA may be used for a WLAN band which extends from 2.41 GHZ to 2.48 GHz.

The low-noise amplifier circuit is usually located very close to the detection device to reduce losses in the feed line. Using a LNA circuit, the effect of noise from subsequent stages of the receive chain is reduced by the gain of the LNA circuit, while the noise of the LNA circuit itself is injected directly into the received signal. Thus, it is preferred when using a LNA circuit that the desired signal power be boosted while adding as little noise and distortion as possible, so that the retrieval of this signal is possible in the later stages in the system.

A good LNA circuit has a low noise figure. Noise figure is often named after its acronym NF. NF is a measure of degradation of the signal-to-noise ratio SNR, caused by components in a radio frequency RF signal chain. The noise figure is defined as the ratio of the output noise power of a device to the portion thereof attributable to thermal noise in the input termination at standard noise temperature T₀, usually 290 K. The noise figure is thus the ratio of actual output noise to that which would remain if the device itself did not introduce noise. It is a number by which the performance of a radio receiver can be specified.

A good LNA circuit further exhibits a large enough gain (20 dB may be considered as a large gain) and should have large enough intermodulation and compression point. In telecommunications, a third-order intercept point IIP3 or TOI is a measure for weakly nonlinear systems and devices, for example receivers, linear amplifiers and mixers. It is based on the idea that the device nonlinearity can be modeled using a low-order polynomial, derived by means of Taylor series expansion. The third-order intercept point relates nonlinear products caused by the third-order nonlinear term to the linearly amplified signal, in contrast to the second-order intercept point that uses second order terms.

Further criteria are operating bandwidth, gain flatness, stability and input and output voltage standing wave ratio (VSWR).

In addition, package PIN reduction imposes to share the LNA circuit input with other blocks like power amplifier also named after its acronym PA. Thus, low input impedance for the LNA circuit is preferred. In accordance, low consumption and low area are also some key important points for the LNA circuit. The common gate LNA circuit is a right way to provide low input impedance. However, such arrangement exhibits a high NF compared to the common source LNA circuit. In other words, common gate LNA circuits are usually too noisy.

It is known from the article “Using Capacitive Cross Coupling Technique in RF Low noise Amplifiers and Down Conversion Mixer Design” by Wei Zhuo et al. an implementation of the cross coupling technique on a differential NMOS LNA circuit. The LNA circuit exhibits a NF value of 3 dB, a gain of 12 dB and a linearity IIP3 of 6.7 dBm under 2.7 V voltage supply.

SUMMARY OF THE INVENTION

The object of the present invention is to alleviate at least partly the above mentioned drawbacks.

More particularly, the invention aims to provide a low-noise amplifier circuit with reduced noise. Indeed, it will be shown that reduced NF of the common gate LNA circuit below 2 dB, notably for transistor belonging to the class of CMOS 40 nm, can be obtained notably by using cross coupling technique on the PMOS input transistor and without using any inductor.

This object is achieved with a low-noise amplifier circuit comprising a first transistor NMOS and a second transistor NMOS, each transistor having a source terminal, a gate terminal and a drain terminal; and a third transistor NMOS and a fourth transistor NMOS, each transistor having a source terminal, a gate terminal and a drain terminal; wherein: the source of the second transistor is coupled to the drain terminal of the first transistor, and the source of the third transistor is coupled to the drain terminal of the fourth transistor; and the gate terminal of the first transistor has a first capacitive coupling to the source terminal of the fourth transistor and the gate terminal of the fourth transistor has a second capacitive coupling to the source terminal of the first transistor; characterised in that the low-noise amplifier circuit further comprises a fifth transistor PMOS and a sixth transistor PMOS, each transistor having a source terminal, a gate terminal and a drain terminal wherein: the drain of the fifth transistor is coupled to the drain terminal of the second transistor, and the drain of the sixth transistor is coupled to the drain terminal of the third transistor; the source terminal of the fifth transistor has a third capacitive coupling to the source terminal of the first transistor; the gate terminal of the fifth transistor has a fourth capacitive coupling to at least the source terminal of the sixth transistor or the source terminal of the fourth transistor; the gate terminal of the sixth transistor has a fifth capacitive coupling to at least the source terminal of the fifth transistor or the source terminal of the first transistor; and the source terminal of the sixth transistor has a sixth capacitive coupling to the source terminal of the fourth transistor.

Preferred embodiments comprise one or more of the following features:

• • the low-noise amplifier circuit has inputs and outputs, the source terminals of the first transistor and the fourth transistor being the inputs of the low-noise amplifier circuit and the drain terminals of the second transistor and the third transistor being the outputs of the low-noise amplifier circuit.
  • the first capacitive coupling increases an effective transconductance of the first transistor and decreases an effective equivalent input resistance of the first transistor both by a factor superior or equal to two, and/or wherein the second capacitive coupling increases an effective transconductance of the fourth transistor and decreases an effective equivalent input resistance of the fourth transistor both by a factor superior or equal to two, and/or wherein the fourth capacitive coupling increases an effective transconductance of the fifth transistor and decreases an input resistance of the fifth transistor both by a factor superior or equal to two, and/or wherein the fifth capacitive coupling increases an effective transconductance of the sixth transistor and decreases an input resistance of the sixth transistor both by a factor superior or equal to two.
  • input low-noise amplifier circuit has an inductive coupling to the ground, preferably with at least a wired inductance.
  • gate potential of the fifth and sixth transistors is fixed by a common loop circuit via two resistances, and wherein the reference voltage of the common loop is taken between outputs of the low noise amplifier circuit.
  • at least one of the source terminal of the fifth or sixth transistors has a resistive coupling to a DC-Voltage, preferably with at least a wired resistor, and wherein gate potential of the first and fourth transistors is fixed by a current mirror via two resistances.
  • at least one of the gate terminal of the first or fourth transistors either has a resistive coupling to a DC-Voltage, preferably with at least a wired resistor, or has a coupling to a DC-Voltage through a MOS transistor.
  • the low-noise amplifier circuit is adapted to amplify at least one signal in the WLAN band or at least one signal in the Bluetooth band.
  • the noise figure values over the WLAN band and/or Bluetooth band of the low-noise amplifier circuit are inferior to 2 dB, preferably inferior to 1.8 dB.
  • a circuit comprising a low-noise amplifier circuit according to some embodiments of the invention, a power amplifier, wherein the low-noise amplifier circuit input is shared with the power amplifier across a Balun.
  • A mobile device comprising a low-noise amplifier circuit according to some embodiments of the invention.

Further features and advantages of the invention will appear from the following description of embodiments of the invention, given as non-limiting examples, with reference to the accompanying drawings listed hereunder.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic view of an example of LNA circuit,

FIG. 1 bis is a schematic view of another example of LNA circuit,

FIG. 2 shows a schematic view of a mathematically equivalent of a single side circuit of an example of LNA circuit according to FIG. 1,

FIG. 3 shows a schematic representation of a transistor M_X with an element equivalent to a cross-coupling configuration,

FIG. 4 is a graphic showing the evolution of the NF with frequency for an example of LNA circuit according to FIG. 1.

FIG. 5 shows an example of application with a Balun of LNA circuit according to an embodiment of the invention.

DETAILED DESCRIPTION OF THE INVENTION

The present invention is related to a low-noise amplifier circuit. FIG. 1 illustrates a schematic view of an example of a LNA circuit 56. Such low-noise amplifier circuit 56 comprises a first transistor M₁ NMOS, a second transistor M₂ NMOS, a third transistor M₃ NMOS and a fourth transistor M₄ NMOS arranged in “a common gate configuration”. The LNA circuit 56 further comprises a fifth transistor M₅ PMOS and a sixth transistor M₆ PMOS arranged with the four previous transistors in a “double common gate configuration”.

The metal-oxide-semiconductor field-effect transistor (MOSFET, MOS-FET, or MOS FET) is a transistor used for amplifying or switching electronic signals. The basic principle of this kind of transistor was first proposed by Julius Edgar Lilienfeld in 1925. In MOSFETs, a voltage on the oxide-insulated gate electrode can induce a conducting channel between the two other contacts called source and drain. The channel can be of n-type or p-type, and is accordingly called an nMOSFET or a pMOSFET (also commonly NMOS, PMOS). It is by far the most common transistor in both digital and analog circuits, though the bipolar junction transistor was at one time much more common.

In the case of LNA circuits, JFETs and HEMTs are often used because a high amplification in the first stage of the amplifier is required. The junction gate field-effect transistor JFET or JUGFET is the simplest type of field effect transistor. High electron mobility transistor HEMT, also known as heterostructure FET HFET or modulation-doped FET MODFET, is a field effect transistor incorporating a junction between two materials with different band gaps (i.e., a heterojunction) as the channel instead of a doped region, as is generally the case for MOSFET.

The two types of transistors NMOS and PMOS are slightly different, however both comprise three terminals labelled gate, source, and drain. In the case of FET, a voltage at the gate can control a current between source and drain.

In the LNA circuit 56 of the FIG. 1 transistors are connected together in order to obtain the low-noise amplification desired. Thus, the source of the second transistor M₂ is coupled to the drain terminal of the first transistor M₁, and the source of the third transistor M₃ is coupled to the drain terminal of the fourth transistor M₄. Besides the gate terminal of the first transistor M₁ has a first capacitive coupling C₁ to the source terminal of the fourth transistor M₄ and the gate terminal of the fourth transistor M₄ has a second capacitive coupling C₂ to the source terminal of the first transistor M₁. Both capacitive couplings C₁ and C₂ provide means for reducing noise in the LAN circuit 56.

Moreover, the two additional transistors PMOS M₅ and M₆ further improve this reduction effect. Thus, the drain of the fifth transistor M₅ is coupled to the drain terminal of the second transistor M₂, and the drain of the sixth transistor M₆ is coupled to the drain terminal of the third transistor M₃.

For coupling each side circuit 10 and 54 of the LNA circuit 56 together, transistors are also cross-coupled between them. Thus, the source terminal of the fifth transistor M₅ has a third capacitive coupling C₃ to the source terminal of the first transistor M₁, the gate terminal of the fifth transistor M₅ has a fourth capacitive coupling C₄ to at least the source terminal of the sixth transistor M₆ (as in FIG. 1bis) or the source terminal of the fourth transistor M₄ (as in FIG. 1), the gate terminal of the sixth transistor M₆ has a fifth capacitive coupling C₅ to at least the source terminal of the fifth transistor M₅ (as in FIG. 1 bis, where the potential of source terminal of fifth transistor M₅ is labelled V_A) or the source terminal of the first transistor M₁ (as in FIG. 1) and the source terminal of the sixth transistor M₆ has a sixth capacitive coupling C₆ to the source terminal of the fourth transistor M₄.

The fourth capacitive coupling C₄ can in fact either be connected the source terminal of the sixth transistor M₆ (as in FIG. 1bis, where the potential of source terminal of sixth transistor M₆ is labelled V_B) or the source terminal of the fourth transistor M₄ (as in FIG. 1), because these both source terminals are also connected by the sixth capacitive coupling C₆. Furthermore, with a RF signal a capacitive coupling, which is preferably at least one wired capacitor, can be equivalent to a short cut. Thus in this regime, the source terminal of the sixth transistor M₆ and the source terminal of the fourth transistor M₄ are similar.

In a preferred embodiment, illustrated on FIG. 1, the fourth capacitive coupling C₄ is made between the gate terminal of the fifth transistor M₅ and the source terminal of the fourth transistor M₄ in terms of phase shift, as well as the fifth capacitive coupling C₅ is made between the gate terminal of the sixth transistor M₆ and the source terminal of the first transistor M₁ for the same reasons.

As the following description will further details these points, the capacitive couplings C₃ and C₆ provide an equivalent input resistance for each side circuit 10 and 54 of the LNA circuit 56. Besides, the capacitive couplings C₄ and C₅ allow in a similar way than the capacitive couplings C₁ and C₂ to provide means for reducing noise in the LAN circuit 56.

In order to detail the process of noise reduction for in the LNA circuit 56, explanations will be focused based on one side circuit 10 of the LNA circuit 56. For this purpose, FIG. 2 describes one side circuit 10 of the LNA circuit of FIG. 1. FIG. 2 is an equivalent circuit in order to better explain the invention, but the actual electronic circuit remains the circuit of FIG. 1 or FIG. 1bis.

The capacitive coupling C₁ and C₄ coming from the side circuit 10 to the side circuit 54 are replaced by mathematically equivalent circuits 32 and 38.

In FIG. 2, the three transistors M₁, M₂ and M₅ are arranged in the so-called “double common gate configuration”. This should be understood as the fact that the RF signal V_in of the circuit 56 is connected to the source terminal of the first transistor M₁ and the source terminal of the fifth transistor M₅ via a capacitor 24. The gate terminal of transistors M₁, M₂ and M₅ are at a different DC voltage level.

The transistors are arranged in serial such that a middle transistor may be defined. In the side circuit 10, the first transistor M₁ and the fifth transistor M₅ are arranged on each side of the middle transistor which is the second transistor M₂.

The side circuit 10 further comprises a first circuit 12 at the source terminal of the first transistor M₁, which with the third capacitive coupling C₃ to the source terminal of the fifth transistor M₅, provides an equivalent input resistance R_in=R_L//Zin_n//Zin_p.

According to the example of FIG. 2, such first circuit 12 comprises a resistor 13 and a voltage source 14. The value of the resistance of resistor 13 is R_s1 which is equal to R_S over 2. The RF voltage source 14 provides a tension equal to V_in=V_RF/2.

Furthermore, the source terminal of the first transistor M₁ is connected to an end of the resistor 13. The other end of the resistor 13 is connected to the RF voltage source 14. The potential of the source terminal of the first transistor M₁ is labelled V_X.

The gate terminal of the first transistor M₁ is connected to an end of a resistor 16. The value of the resistance of resistor 16 is R₁. The other end of the resistor 16 is connected to a potential labelled V_DC1. Such potential may be provided by a voltage source which is not represented on FIG. 2. Instead of resistor 16, a MOS transistor could be used to bias first transistor M₁. But using a resistor is better, because it minimizes parasitic capacitor and it improves matching.

The drain terminal of the first transistor M₁ is connected to the source terminal of the second transistor M₂ as a cascode. The gate of the second transistor M₂ is connected to a potential equal to V₂. Such potential may be provided by a voltage source which is not represented on FIG. 2.

The gate terminal of the fifth transistor M₅ is connected to an end of a resistor 18. The value of the resistance of resistor 18 is R₂. The other end of the resistor 18 is connected to a potential labelled V_DC2. Such potential may be provided by a voltage source which is not represented on FIG. 2.

The source terminal of the fifth transistor M₅ is connected to an end of a resistor 20 which is at a potential V_A. The value of the resistance of resistor 20 is R_L. The other end of the resistor 20 is connected to a potential labelled V₃. Such potential may be provided by a voltage source which is not represented on FIG. 2. The source terminal of the fifth transistor M₅ is also connected to the source terminal of the first transistor M₁, the potential V_X, via a branch 22 including a capacitor 24.

The drain terminal of the fifth transistor M₅ is both connected to the drain terminal of the second transistor M₂ and an output branch 26 of the LNA circuit 10. Such branch 26 comprises a capacitor 28 and a resistor 30. The value of the resistance of resistor 30 is R_C. The current issued from the drain terminal of the fifth transistor M₅ is labelled i_p, the current issued from the drain terminal of the second transistor M₂ is labelled i_n and the current circulating in the branch 26 is labelled i_out. According to Kirchhoff s first law applied in this case, it can be written that the sum of the currents i_n and i_p is equal to i_out.

The input matching is mainly done by sizing and adjusting the bias current of the input transistors M₁ and M₅ such that Zin_n*Zin_p/(Zin_n+Zin_p) be equal to the input impedance which is the resistance Rs₁. For the sake of illustration, the value of Rs₁ will be set to 10 Ω, be it understood that any other value may be considered.

The side circuit 10 also comprises a second circuit 32 for increasing the effective transconductance of the fifth transistor M₅ and decreasing the equivalent input resistance R_p. This second circuit 32 is mathematically similar to the fourth capacitive coupling C₄ in the LNA circuit 56.

In field effect transistors, and MOSFETs in particular, transconductance is the change in the drain/source current divided by the change in the gate/source voltage with a constant drain/source voltage. The transconductance is labelled gm_x with X the number associated to the transistor. Typical values of gm for a small-signal field effect transistor are 1 to 30 millisiemens. In such case, increasing the effective transconductance of the fifth transistor M₅ means that the transconductance gm₅ of the fifth transistor M₅ is higher with the second circuit 32 in the side circuit 10 than without the second circuit 32 in the side circuit 10.

The side circuit 10 further comprises a third circuit 38 for increasing the effective transconductance of the first transistor M₁ and for decreasing the effective equivalent input resistance R_n. In such case, decreasing the effective equivalent input resistance R_p and R_n means that the equivalent input resistance is lower with the circuits 32 and 38 than without.

Such side circuit 10 allows matching the input under low impedance.

Furthermore, the addition of second and third circuits 32 and 38 which respectively increase the effective transconductance of the first transistor M₁ and the fifth transistor M₅ and decrease the effective equivalent input resistance R_P and R_n gives significant improvement on NF. To show that increasing the effective transconductance of the fifth transistor M₅ and decreasing the effective equivalent input resistance R_P results in a reduced NF, some calculations will be presented in the following. Only the relevant equations are presented here, other equations are given in appendix II.

Noise in a LNA circuit is mainly due to the current thermal noise of the transistors. This section is also only based on side circuit 10 of the LNA circuit 56. Such noise may indeed be expressed as in the following equation relatively to side circuit 10:

$\begin{array}{cc}\begin{array}{c}\mathrm{NF}=10\log \left(1+\frac{\stackrel{\_}{i_{\mathrm{out}}^2}}{g_m^2\stackrel{\_}{{\mathrm{vsn}}^2}}\right)\\ =10\log \left(1+\frac{\stackrel{\_}{i_n^2+i_p^2}}{g_m^2\stackrel{\_}{{\mathrm{vsn}}^2}}\right)\end{array}& \left(\mathrm{Equation}1\right)\end{array}$

Wherein:

• • g_m=i_out/V_in is the complete common gate transconductance, vsn is the resistor input thermal noise {square root over (4KTR_s1)}.

The expressions of both currents i_p and i_n are also known. For the expression of the current i_p, Equation 2 is obtained:

$\begin{array}{cc}\stackrel{\_}{i_p^2}=\frac{4\mathrm{KT}{\gamma }_pg_{m_5}}{{\left(1+\frac{R_{s_1}}{R_n}+{\mathrm{Gm}}_pR_{s_1}\right)}^2}& \left(\mathrm{Equation}2\right)\end{array}$

Wherein:

• • K is the Boltzmann constant,
  • T is the temperature of the LNA circuit 10,
  • γ_p is the input channel thermal noise,
  • gm₅ is the transconductance of the fifth transistor M₅,
  • R_n is the input equivalence resistance of the transistor M₁,
  • Gm_p is the effective transconductance of the fifth transistor M₅,
  • R_s1 is the value of the resistance of resistor 13 in circuit 12.

It can be noticed from this Equation 2 that when the transconductance Gm_p is increased, the current i_p is reduced. This results in a reduction of the thermal noise due to this current.

For the expression of the current i_n, Equation 3 is obtained:

$\begin{array}{cc}\stackrel{\_}{i_n^2}=\frac{4\mathrm{KT}{\gamma }_n{\mathrm{gm}}_1}{{\left(1+\frac{R_{s1}}{R_p}+{\mathrm{Gm}}_nR_{s1}\right)}^2}& \left(\mathrm{Equation}3\right)\end{array}$

Wherein:

• • γ_n is the output channel thermal noise,
  • g_m1 is the transconductance of the first transistor M₁,
  • R_p is the input equivalent resistance of the fifth transistor M₅.
  • Gm_n is the effective transconductance of the first transistor M₁.

It can be noticed from this Equation 3 that when the effective input equivalent resistance R_p is decreased, the current i_n is reduced. This results in a reduction of the thermal noise due to this current.

Therefore, it has been shown that increasing the transconductance Gm_p of the fifth transistor M₅ and decreasing the input equivalent resistance R_p results in a reduced NF for the side circuit 10.

This results in a side circuit 10 with reduced noise, the reduction being stable independently from the use and the temperature. Compared to other noise cancelling technique, the current cost of such technique is reduced. In addition, the proposed technique does not exhibit huge NF dispersion with process.

The second circuit 32 may increase the effective transconductance Gm_p of the fifth transistor M₅ by a factor superior or equal to two. Indeed, in such case, the effect on the noise reduction is more sensitive.

The second circuit 32 may decrease the equivalent input resistance R_p of the fifth transistor M₅ by a factor superior or equal to two. Indeed, in such case, the effect on the noise reduction is more sensitive.

Similar configurations, effects and advantages are present on side circuit 54 thanks to capacitive couplings C₂, C₅ and C₆. Besides effects on both side circuits 10 and 54 can be simultaneous and additives.

This enables to avoid the use of common source LNA circuit. Therefore, the advantage of the common gate structure LNA circuit is kept. Notably, the linearity which can be expressed in terms of IIP3 is better.

The low-noise amplifier will thus exhibits noise figure NF values which are inferior to 2 dB, and even preferably 1.8 dB in either the WLAN band or the Bluetooth band. Bluetooth is often named after its acronym which is BT.

This enables to provide a LNA circuit 56 with improved properties, notably in term of linearity, and reduced noise compared to other LNA circuit in the common gate configuration.

Each side circuit 10 and 54 may be without any coil. It should be understood that coil means a component with a significant inductance and not unwanted inductance due to the imperfectness of wires for instance. Indeed, a side circuit 10 without any coil enables to better follow the input signal that is desired to be amplified.

The second and/or third circuits 32 and 38 of the side circuit 10 may only comprise passive elements. As explained above, the passive elements are intended to increase the effective transconductance of the fifth transistor M₅ and to decrease the input equivalence resistance R_p. This enables to avoid consuming additional power in the second and/or third circuits 32 and 38.

According to the example of FIG. 2, the second circuit 32 of the side circuit 10 comprises a fourth capacitor C₄ cross-coupled between the source and the gate terminals of the fifth transistor M₅. This enables to increase the effective transconductance of the fifth transistor M₅. This configuration aims to create a mathematically equivalent configuration to the fourth capacitive coupling C₄ in the LNA circuit 56 of FIG. 1bis.

Preferably, in the second circuit 32, the fourth capacitor C₄ is cross-coupled between the source and the gate terminals of the fifth transistor M₅ by introducing an amplifier 40 with a gain of −1 in serial with the fourth capacitor C₄. This enables to increase in an even better way the effective transconductance of the fifth transistor M₅.

Adding a second circuit 32 in cross coupling for the fifth transistor M₅ brings a noise cancelling path which is entirely new.

The way calculation may be led in this case is illustrated by FIG. 3. FIG. 3 is a schematic representation of a transistor with a circuit in equivalent cross coupling configuration.

FIG. 3 is a schematic view of a configuration wherein a transistor 48 labelled M_X. FIG. 3 is an equivalent circuit in order to better explain the invention, but the actual electronic circuit remains the circuit of FIG. 1 or FIG. 1bis. The transconductance of the transistor M_X is gm_x. It corresponds to the ratio between the current entering the drain terminal and the voltage of the source terminal. The current entering the drain terminal of the transistor M_X is labelled i_X and the source terminal is put at the tension V_X of a voltage source which delivers such tension V_X. When this transistor M_X is used in a circuit, as the circuit 44 of FIG. 3, the ratio between the current i_X and the voltage V_X changes. This ratio is then called the effective transconductance of the transistor M_X and labelled Gm_x.

In the case of FIG. 3, there is a feedback from the source terminal to the gate terminal of the transistor M_X with an amplifier 50 of gain −A. Thus, the gate terminal is controlled by a −AV_X potential.

Therefore, by using, for instance Kirchhoff's laws, it can be obtained:

$\begin{array}{cc}{\mathrm{Gm}}_x=\frac{i_X}{V_X}=g_{\mathrm{mx}}\left(1+A\right)& \left(\mathrm{Equation}4\right)\end{array}$

Applied to the case of the second circuit 32, this gives the following equation:

Gm ₅ =gm ₅ (1+A)   (5)

It thus appears that the effective transconductance is increased by a factor 1+A. If A is equal to 1, this results in an increase of the effective transconductance by a factor 2.

Such effect is illustrated by the simulation of the graphic of FIG. 4. This graphic shows the evolution of the NF with the frequency, in the range from 2.4 GHz to 2.5 GHz. The curve 50 corresponds to the evolution simulated for the side circuit 10 without the fourth capacitor C₄ whereas the curve 52 illustrates the evolution simulated for the side circuit 10 with the fourth capacitor C₄. In the case without the fourth capacitor C₄, it can be noticed that NF is superior to 2.25 dB at each frequency. For comparison, it can be noticed that NF is inferior to 1.85 dB at each frequency for the case of curve 52. The presence of the fourth capacitor C₄ therefore enables to reduce the noise of the side circuit 10.

According to the example of FIG. 2, the third circuit 38 of the low-noise side circuit 10 comprises a first capacitor C₁ cross-coupled between the source and the gate terminals of the first transistor M₁. This enables to obtain a reduction of effective equivalent input resistance R_n. This configuration aims to create a mathematically equivalent configuration to the first capacitive coupling C₁ in the LNA circuit 56 of FIG. 1.

Preferably, in the third circuit 38, the first capacitor C₁ is cross-coupled between the source and the gate terminals of the first transistor M₁ by introducing an amplifier 36 with a gain of −1 in serial with the first capacitor C₁. This enables to obtain an even better reduction of effective equivalent input resistance R_n.

Cross-coupling C₁ brings another noise cancelling path, increases the transconductance gm₁ by a factor 2 and allows to reduce power consumption of the side circuit 10. The noise factor of the NMOS input stage can be written has:

$\begin{array}{cc}F=1+\frac{1}{{{\mathrm{gm}}_1\left(1+A\right)}^2{\mathrm{Rs}}_1}·\left(\frac{\gamma }{\alpha }+\frac{{\left(1+{\mathrm{gm}}_1{\mathrm{Rs}}_1\right)}^2}{\frac{{\mathrm{gm}}_1}{{\mathrm{gm}}_2}}\right)& \left(\mathrm{Equation}6\right)\end{array}$

Wherein:

• • A represents the gain of the auxiliary cross-coupling amplifier,
  • γ is the channel thermal noise,
  • α is defined as the ratio between gm₁ to g_d0, g_d0 being the transconductance when the difference of voltage between the drain and the source terminal V_DS1 is equal to 0; thus it can be written α=(gm₁/g_d0).
  • g_m1 is the transconductance of the first transistor M₁.
  • g_m2 is the transconductance of the second transistor M₂.

In case, A=1 and 1/(gm₁(1+A))=Rs₁, then the improvement of the factor noise is given by:

$\begin{array}{cc}F=1+\frac{1}{2}·\left(\frac{\gamma }{\alpha }+\frac{4{\mathrm{gm}}_2}{{\mathrm{gm}}_1}\right)& \left(\mathrm{Equation}7\right)\end{array}$

Thus, this results in a reduction of the noise of the side circuit 10.

To sum up, the combination of the first and the fourth capacitors C₁ and C₄ enables to obtain a side circuit 10 which exhibits a low NF and low consumption compared to existing common gate LNA circuit solutions. Indeed, it only consumes 5 mA differential under 1.2 V and presents a linearity IIP3 of 10 dBm with a NF of 1.8 dB.

Many other elements may be considered for obtaining the same effect. The choice of the elements may be done thanks to a method for reducing the noise of a low-noise amplifier side circuit 10.

Such method, for a given low noise amplifier side circuit 10 comprises the step of choosing the elements of the second circuit 32 for increasing the effective transconductance of the fifth transistor M₅. This step of choosing encompasses both the notion of choosing which kind of elements should be considered and choosing their specific properties.. The properties to be chosen for a capacitor are for instance its capacitance, its internal resistance and its breakdown voltage. Among the different properties of an element, it should be distinguished between the properties which have an impact on the effective transconductance of the fifth transistor M₅ and the properties which have nearly no impact on these parameters. In the case of a capacitor, its capacitance, its internal resistance may modify the effective transconductance of the fifth transistor M₅ whereas the breakdown voltage does not have impact on the effective transconductance of the fifth transistor M₅. Thus, it should be understood that the choosing step may comprise an optimizing step of the properties of the elements. This optimizing step may be carried out by using a merit function which is minimized for a maximum effective transconductance of the fifth transistor M₅.

The method also comprises a step of choosing the elements of the third circuit 38 for decreasing the effective equivalent input resistance R_n.

Similar remarks made concerning the step of choosing the elements of the second circuit 32 apply there for this step.

Such method enables to obtain the side circuit 10 as previously described. This method is thus a way to obtain a side circuit 10 with reduced noise, the reduction being stable independently from the use and the temperature.

Further, such method may be performed based on a computer program comprising instructions for performing the method. The program is executable on a programmable device. The application program may be implemented on a high-level procedural or object-oriented programming language, or in assembly or machine language if desired. In any case, the language may be compiled or interpreted language. The program may be a full installation program, or an update program. In the latter case, the program is an update program that updates a programmable device, previously programmed performing parts of the method, to a state wherein the device is suitable for performing the whole method.

The program may be recorded on a data storage medium. The data storage medium may be any memory adapted for recording computer instructions. The data storage medium may thus be any form of nonvolatile memory, including by way of example semiconductor memory devices, such as EPROM, EEPROM, and flash memory devices; magnetic disks such as internal hard disks and removable disks; magneto-optical disks; and CD-ROM disks.

This method can be used in a similar way on side circuit 54 of the LNA circuit 56. The use of the above mentioned method on both sides provide a method for reducing the noise on the whole LNA circuit 56.

Some other elements can be present on the LNA circuit 56 as an inductive coupling between inputs 58 and 60 of low-noise amplifier circuit 56 and the ground, which is preferably at least one wired inductance.

The use of the previously described LNA circuit 56 in the circuit enables to benefit from its low noise properties.

Such LNA circuit 56 may be used for several different applications.

For instance, it may be proposed a circuit comprising a low-noise amplifier circuit as previously described and a power amplifier. An RF power amplifier is a type of electronic amplifier used to convert a low-power radio-frequency signal into a larger signal of significant power, typically for driving the antenna of a transmitter. It is usually optimized to have high efficiency, high output power compression, good return loss on the input and output, good gain, and optimum heat dissipation.

According to this example, the low-noise amplifier circuit input is shared with the power amplifier across a Balun. In the specific case of FIG. 4, the Balun is a Balun 5020.

A Balun is a type of electrical transformer that can convert electrical signals that are balanced about ground (differential) to signals that are unbalanced (single-ended), and the reverse. They are also often used to connect lines of differing impedance. The origin of the word balun is bal(ance)+un(balance). Baluns can take many forms and their presence is not always obvious. They always use electromagnetic coupling for their operation.

FIG. 5 shows an example of application with a Balun of LNA circuit according to an embodiment of the invention. The Balun is a 5020 ohms balun. There is an Antenna A on one side of the balun B, whereas there are a common gate low noise amplifier LNA and a power amplifier PA on the other side of the balun B. Low noise amplifier LNA presents an input LNA IN and an output LNA OUT. Power amplifier PA presents an input PA IN and an output PA OUT.

Low noise amplifier input LA IN is shared with a power amplifier output PA OUT through a Balun B. The power amplifier output PA OUT is connected, through two capacitors C7 and C8, to the low noise amplifier input LA IN with two GO2 switches M₁ and M₄. Switches M₁ and M₄ prevent low noise amplifier LNA over voltage when power amplifier PA is active.

When power amplifier PA is active, the switches M₁ and M₄ are off and the low noise amplifier LNA is off. The power amplifier PA only sees the 20 ohms of the balun B secondary access and the off impedance of the low noise amplifier input LNA IN.

When Antenna A receiver is active, the switches M₁ and M₄ are on and the power amplifier PA is off The low noise amplifier LNA only sees the 20 ohms and the off impedance of the power amplifier PA.

This application is full integrated and presents no external components for the receiver transmitter switch.

Alternatively, it may be considered a mobile device comprising a low-noise amplifier circuit 56 as previously described.

The low-noise amplifier circuit 56 may also be used to amplify at least one signal in the WLAN band.

Other applications may be considered since this LNA circuit enables the reception of WIFI or BT (BT stands for Bluetooth) either independently or simultaneously. Indeed, in terms of wireless networking communications, two of the currently dominant, standardized approaches are specified in Wireless LAN (WLAN, “Wi-Fi”, 802.11 abgn) standard and the Bluetooth standard.

WLAN devices are frequently used, for example, to provide wireless Internet connectivity and operate two frequency bands, i.e., a low band disposed in the 2.4 GHz Industrial, Scientific and Medical Band (ISM band) and a high band disposed in the 5 GHz range. Bluetooth devices also operate in the 2.4 GHz and are frequently used, for example, for short range communications, e.g., between a mobile phone and an associated earplug device.

The invention has been described with reference to preferred embodiments. However, many variations are possible within the scope of the invention.

APPENDICE I
TABLE OF ACRONYMS
ACRONYMS       MEANING
BT             Bluetooth
EPROM          Erasable Programmable Read-Only
               Memory
EEPROM         Electrically Erasable Programmable
               Read-Only Memory
FET            Field-Effect Transistor
HFET           Heterostructure FET
IIP3           Third Order Input Intercept Point
JFET or JUGFET Junction gate FET
LNA            Low-Noise Amplifier
MODFET         MOdulation-Doped FET
MOSFET         Metal-Oxide-Semiconductor FET
NF             Noise Figure
NMOS           NMOSFET = n-Channel MOSFET
PA             Power Amplifier
PMOS           PMOSFET = p-Channel MOSFET
RF             Radio Frequency
SNR            Signal-to-Noise Ratio
TOI            Third-Order Intercept Point
VSWR           Voltage Standing Wave Ratio
WLAN           Wireless Local Area Network

APPENDICE II: ADDITIONAL FORMULAS CONCERNING LNA CIRCUIT

The following expressions are related to the example of FIG. 2 and can be derived by applying known physical laws to the circuit of FIG. 2. Same notation as before is used.

1) Expression of the effective transconductance of the first transistor M₁

$\begin{array}{cc}{\mathrm{Gm}}_n=\frac{i_n}{V_X}=\frac{1+2\left({\mathrm{gm}}_1+{\mathrm{gmbs}}_1\right){\mathrm{rds}}_1}{{\mathrm{rds}}_1+\frac{1}{{\mathrm{gm}}_2+{\mathrm{gmbs}}_2}}& \left(\mathrm{Equation}8\right)\end{array}$

Wherein:

• • gmbs₁; gmbs₂ is the backgate transconductance of transistors M₁ and M₂
  • gm₁; gm₂ is the transconductance of transistors M₁ and M₂
  • rds₁ is the resistance between the drain and the source terminals of the first transistor M₁ in small-signal approximation,

2) Expression of the potential V_in

$\begin{array}{cc}{V}_{\mathrm{in}}=\mathrm{Vx}\left(1+{\mathrm{Gm}}_n{\mathrm{Rs}}_1+\frac{{\mathrm{Rs}}_1}{\mathrm{Rp}}\right)& \left(\mathrm{Equation}9\right)\end{array}$

3) Expression of the transconductance of the first transistor M₁

$\begin{array}{cc}{\mathrm{gm}}_n=\frac{{\mathrm{Gm}}_n}{1+\frac{{\mathrm{Rs}}_1}{\mathrm{Rp}}+{\mathrm{Gm}}_n{\mathrm{Rs}}_1}& \left(\mathrm{Equation}10\right)\end{array}$

4) Expression of the equivalent input resistance R_n of the first transistor M₁

$\begin{array}{cc}{R}_n=R_L//{\mathrm{Zin}}_n=R_L//\left(\frac{\frac{1}{{\mathrm{gm}}_2+{\mathrm{gmbs}}_2}+{\mathrm{rds}}_1}{1+2\left({\mathrm{gm}}_1+{\mathrm{gmbs}}_1\right){\mathrm{rds}}_1}\right)& \left(\mathrm{Equation}11\right)\end{array}$

Wherein:

• • rds₁ is the resistance between the drain and the source terminals of the first transistor M₁ in small-signal approximation,
  • gmbs₁; gmbs₂ is the backgate transconductance of transistors M₁ and M₂,
  • // means that the resistance considered is the equivalent resistance of two resistances in parallel. In other words, R=R_X//R_Y means that the resistance R is equal to

$R=\frac{R_X·R_Y}{R_X+R_Y}.$

5) Expression of the effective transconductance of the fifth transistor M₅

$\begin{array}{cc}{\mathrm{Gm}}_p=\frac{i_p}{\mathrm{Vx}}=\frac{1+2\left({\mathrm{gm}}_3+{\mathrm{gmbs}}_3\right){\mathrm{rds}}_3}{{\mathrm{rds}}_3+\left({\mathrm{rout}}_n//\mathrm{Rc}\right)}& \left(\mathrm{Equation}12\right)\end{array}$

Wherein:

• • Gm_p is the effective transconductance of the fifth transistor M₅,

6) Other expression of the potential V_in

$\begin{array}{cc}{V}_{\mathrm{in}}=V_x\left(1+{\mathrm{Gm}}_p{\mathrm{Rs}}_1+\frac{{\mathrm{Rs}}_1}{\mathrm{Rn}}\right)& \left(\mathrm{Equation}13\right)\end{array}$

7) Expression of the transconductance of the fifth transistor M₅

$\begin{array}{cc}{\mathrm{gm}}_p=\frac{{\mathrm{Gm}}_p}{1+\frac{{\mathrm{Rs}}_1}{\mathrm{Rn}}+{\mathrm{Gm}}_p{\mathrm{Rs}}_1}& \left(\mathrm{Equation}14\right)\end{array}$

8) Expression of the equivalent output resistance Rp of the fifth transistor M₅

$\begin{array}{cc}\mathrm{Rp}=R_L//\frac{\left({\mathrm{Rout}}_n//\mathrm{Rc}\right)+{\mathrm{rds}}_5}{1+2\left({\mathrm{gm}}_5+{\mathrm{gmbs}}_5\right){\mathrm{rds}}_5}& \left(\mathrm{Equation}15\right)\end{array}$

• • Rout_n is the output resistance seen by the cascode second transistor M₂,
  • rds₅ is the resistance between the drain and the source terminals of the fifth transistor M₅ in small-signal approximation.
  • gmbs₅ is the backgate transconductance of the fifth transistor M₅,

Claims

1. A low-noise amplifier circuit comprising: a first transistor NMOS and a second transistor NMOS, each transistor having a source terminal, a gate terminal and a drain terminal; anda third transistor NMOS and a fourth transistor NMOS, each transistor having a source terminal, a gate terminal and a drain terminal;

2. The low-noise amplifier circuit according to claim 1, wherein the low-noise amplifier circuit has inputs and outputs, the source terminals of the first transistor and the fourth transistor being the inputs of the low noise amplifier circuit and the drain terminals of the second transistor and the third transistor being the outputs of the low-noise amplifier circuit.

3. The low-noise amplifier circuit according to claim 1, wherein the first capacitive coupling increases an effective transconductance of the first transistor and decreases an effective equivalent input resistance Rn of the first transistor both by a factor superior or equal to two, and/or wherein the second capacitive coupling increases an effective transconductance of the fourth transistor and decreases an effective equivalent input resistance of the fourth transistor both by a factor superior or equal to two, and/or wherein the fourth capacitive coupling increases an effective transconductance of the fifth transistor and decreases an input resistance of the fifth transistor both by a factor superior or equal to two, and/or wherein the fifth capacitive coupling increases an effective transconductance of the sixth transistor and decreases an input resistance of the sixth transistor both by a factor superior or equal to two.

4. The low-noise amplifier circuit according to claim 1, wherein input low-noise amplifier circuit has an inductive coupling to the ground, preferably with at least a wired inductance.

5. The low-noise amplifier circuit according to claim 1, wherein gate potential of the fifth and sixth transistors is fixed by a common loop circuit via two resistances, and wherein the reference voltage of the common loop is taken between outputs of the low noise amplifier circuit.

6. The low-noise amplifier circuit according to claim 1, wherein at least one of the source terminal of the fifth or sixth transistors has a resistive coupling to a DC-Voltage, preferably with at least a wired resistor, and wherein gate potential of the first and fourth transistors is fixed by a current mirror via two resistances.

7. The low-noise amplifier circuit according to claim 1, wherein at least one of the gate terminal of the first or fourth transistors either has a resistive coupling to a DC-Voltage, preferably with at least a wired resistor, or has a coupling to a DC-Voltage through a MOS transistor.

8. The low-noise amplifier circuit according to claim 1, adapted to amplify at least one signal in the WLAN band or at least one signal in the Bluetooth band.

9. The low-noise amplifier circuit according to claim 8, wherein the noise figure values over the WLAN band and/or Bluethooth band of the low-noise amplifier circuit are inferior to 2 dB, preferably inferior to 1.8 dB.

10. A circuit comprising: a low-noise amplifier circuit according to claim 1, anda power amplifier,wherein the low-noise amplifier circuit input is shared with the power amplifier across a Balun.

11. A mobile device comprising a low-noise amplifier circuit according to claim 1.