BACKGROUND
Field of the Invention
The present invention relates to integrated circuits and more particularly generating a reference signal in integrated circuits.
Description of the Related Art
In general, a bandgap reference circuit provides a voltage reference with improved temperature stability and is less dependent on power supply voltage than other known voltage reference circuits. Bandgap reference circuits typically generate a reference voltage approximately equal to the bandgap voltage of silicon extrapolated to zero degrees Kelvin, i.e., VG0=1.205V. To achieve a target reference voltage, these circuits typically use voltage multiplication, which increases output noise. Typical voltage reference circuits include a current mirror coupled to the power supply and the voltage reference node to provide a current proportional to absolute temperature (i.e., PTAT) to the voltage reference node. These circuits can be made with relatively low cost, but have the disadvantages of having high noise for a particular power consumption and being sensitive to power supply noise, which reduces the accuracy of the voltage reference. Accordingly, improved techniques for generating reference voltages are desired.
SUMMARY OF EMBODIMENTS OF THE INVENTION
A low-noise voltage reference generator that utilizes internal gain and feedback to generate an output signal having reduced sensitivity to power supply variations and loading conditions is described. In at least one embodiment of the invention, a method includes generating a current based on a voltage drop across a resistor. The voltage drop is based on a second voltage drop across a gate terminal of a transistor and a source terminal of the transistor. The method includes the current using a reference voltage to generate a mirrored current through a node coupled to the drain terminal of the transistor. The method includes generating a level-shifted voltage using a voltage on the node. The method includes buffering the level-shifted voltage using a power supply voltage to generate the reference voltage.
In at least one embodiment of the invention, an apparatus includes a buffer amplifier configured to transfer a signal from an input node to an output reference node using a power supply voltage on a first power supply node. The apparatus includes a current mirror coupled to the output reference node and configured to generate a mirrored current through a first node based on a first current through a second node and a voltage on the output reference node. The apparatus includes a resistor coupled between the second node and a second power supply node. The apparatus includes a first transistor of a first type having a gate terminal coupled to the first node and a source terminal coupled to the second node. The first transistor is configured to develop a voltage drop across terminals of the resistor to generate the first current. The apparatus includes a level-shifting circuit configured to level shift a voltage on the first node to drive the input node of the buffer amplifier.
BRIEF DESCRIPTION OF THE DRAWINGS
The present invention may be better understood, and its numerous objects, features, and advantages made apparent to those skilled in the art by referencing the accompanying drawings.
FIG. 1 illustrates a circuit diagram of an exemplary bandgap voltage reference generator circuit.
FIG. 2 illustrates a circuit diagram of an exemplary VGS/R voltage reference generator circuit.
FIG. 3 illustrates a circuit diagram of an exemplary low noise voltage reference generator circuit and load regulator consistent with at least one embodiment of the invention.
FIG. 4 illustrates a circuit diagram of an exemplary low noise voltage reference generator circuit and load regulator including an active level shifting circuit consistent with at least one embodiment of the invention.
FIG. 5 illustrates a circuit diagram of an exemplary low noise voltage reference generator circuit and load regulator including an active level shifting circuit consistent with at least one embodiment of the invention.
FIG. 6 illustrates a circuit diagram of an exemplary low noise voltage reference generator circuit and load regulator including a passive level shifting circuit consistent with at least one embodiment of the invention.
FIG. 7 illustrates a circuit diagram of an exemplary low noise voltage reference generator circuit and load regulator including a passive level shifting circuit with additional voltage headroom consistent with at least one embodiment of the invention.
FIG. 8 illustrates an exemplary complementary version of low noise voltage reference generator circuit and load regulator of FIG. 3 consistent with at least one embodiment of the invention.
FIG. 9 illustrates a circuit diagram of an exemplary low noise voltage reference generator circuit and load regulator including an active level shifting circuit having selectable device parameters consistent with at least one embodiment of the invention.
The use of the same reference symbols in different drawings indicates similar or identical items.
DETAILED DESCRIPTION
Referring to FIG. 1, a typical low-noise, on-chip regulated voltage reference is implemented in two stages. Voltage reference generator 101 establishes a voltage that is relatively independent of the external supply (i.e. has sufficient power supply rejection for a target application) and has relatively low noise. Voltage reference generator 101 provides the low-noise and supply insensitive reference to buffer circuit 103, which provides adequate current for a variety of loading conditions. Voltage reference generator 101 is a typical bandgap voltage reference that utilizes temperature behavior of diodes to generate a voltage having a negative temperature coefficient (i.e., a negative first-order temperature coefficient) and a voltage having a positive temperature coefficient (i.e., a positive first-order temperature coefficient) and combines those voltages to produce an approximately zero temperature coefficient reference voltage. Voltage reference generator 101 takes advantage of two electrical characteristics to achieve the desired reference voltage Vbg: the base-emitter voltage VBE, of a bipolar transistor is nearly complementary to absolute temperature T, e.g., VBE=(−1.5 mV/° K×T+1.22)V, and thermal voltage VT is proportional to absolute temperature T, i.e, VT=kT/q, where k is the Boltzmann constant and q is the magnitude of the electrical charge on the electron. Although pure diodes are preferable because they generate a higher diode drop for the same current, the typical bandgap voltage reference manufactured in a complementary metal-oxide-semiconductor (CMOS) process uses diode-coupled, bipolar junction transistors (i.e., BJTs or bipolar transistors), which are readily available in a CMOS process (e.g., pnp bipolar transistors formed from p-type diffusion, an n-type well, and a p-type well in the CMOS process). The voltage across the diodes (or diode-coupled bipolar junction transistors) has a negative temperature coefficient, but the voltage difference between two diode drops in which the current densities differ is proportional to absolute temperature (PTAT). The use of two banks of bipolar junction transistors of different sizes (or two identical banks with different currents) can generate voltage difference ΔVBE. The typical bandgap forces voltage difference ΔVBE across a relatively temperature insensitive resistor (e.g., a polysilicon resistor) using negative feedback, which generates a PTAT current through the resistor. Another resistor is placed in series, which amplifies voltage difference ΔVBE to cancel the negative temperature coefficient of the diode drop.
For example, reference voltage Vbg is stable with respect to temperature variations. A voltage proportional to absolute temperature (i.e., a PTAT voltage) may be obtained by taking the difference between two base-emitter voltages of transistors biased at different current densities:
where J1 and J2 are the current densities of corresponding bipolar transistors. Accordingly, voltage reference circuit 101 includes a pair of pnp bipolar transistors (i.e., transistors 110 and 112) that are coupled in a diode configuration (i.e., the collectors and bases of these transistors are coupled together) and coupled to ground. Transistor 110 has area A that is m times larger than the area of transistor 112. In addition, the current mirror including transistors 120 and 122 and that is used to bias transistors 110 and 112 has a current ratio of n. Thus, the current density ratio of transistor 110 and transistor 112 varies by a factor of n×m. The emitter of transistor 112 is coupled to an inverting input of operational amplifier 104. The emitter of transistor 110 is coupled, via resistor R3, to the non-inverting input of operational amplifier 104. Operational amplifier 104 maintains equivalent voltages at nodes 114 and 118, i.e., V118=V114=VBE112. Hence, the difference between VBE112 and VBE110 (i.e., ΔVBE112,110) forms across resistor R3. Operational amplifier 104 and transistors 120 and 122 convert this voltage difference into a current (i.e., current I) proportional to the voltage difference, which is proportional to the thermal voltage VT:
Since the thermal voltage VT is proportional to absolute temperature via the constant factor k/q, k=1.38×10−23 J/K and q=1.6*10−19 C, the current proportional to the voltage difference is also proportional to an absolute temperature, i.e., I is a PTAT current.
Transistor 110 provides a voltage nearly complementary to absolute temperature (i.e., a CTAT voltage) because the base-to-emitter voltage VBE of a bipolar transistor is nearly complementary to absolute temperature. By compensating the PTAT current with a CTAT voltage, transistors 120, 122, 110, and 112, and resistors R1, R2, and R3, may be appropriately sized to generate a particular reference voltage output having an approximately zero temperature coefficient:
Vbg=VBE110+(1+p)VT ln (nm).
If n, m, and p are selected to generate Vbg with zero temperature coefficient at 300° K, then
Vbg=0.74V+0.45V=1.19V≈1.2V.
Vbg is approximately equal to the bandgap voltage of silicon extrapolated to zero degrees Kelvin VG0=1.205V.
Adding a PTAT voltage to a diode drop produces an approximately zero temperature coefficient point at approximately 1.2 V, resulting in a circuit that is not substantially sensitive to the effects of process variation on the bipolar junction transistor. The ratiometric manner in which the resistors are used also reduces effects of process variation, aging, and strain sensitivity. In an exemplary embodiment of the voltage reference, the ratio of R2 to R3 (i.e., the value p) is approximately 5 to 10 (i.e., p=R2/R3≈5-10). Operational amplifier 104 compares voltage difference ΔVBE (e.g., a voltage less than 100 mV) along with input-referred noise of operational amplifier 104 and thus substantially degrades the signal-to-noise ratio of reference voltage Vbg. To effectively reduce the noise, a higher power operational amplifier may be used to gain the input signal to obtain a target reference voltage level over temperature. Operational amplifier 104 has a feedback factor that causes a reduction in loop gain and bandwidth from the open loop gain. Buffer circuit 103 is series-coupled in the signal path for load regulation and is coupled to the power supply, which introduces power supply noise into the output signal VOUT.
A technique for reducing effects of noise on the output of a voltage reference generator as compared to noise sensitivity of the output of voltage reference generator 100 includes using a VGS/R topology. Referring to FIG. 2, voltage reference generator 200 uses the zero temperature coefficient point of transistor M5, i.e., the point where a constant current causes no change in the gate-to-source voltage VGS of transistor M5 due to cancellation of the negative temperature coefficient of the threshold voltage of transistor M5 with the positive temperature coefficient of overdrive voltage VDSAT of transistor M5. If transistor M5 is in the saturation region of operation, the gate-to-source voltage VGS of transistor M5 develops across resistor R4, causing current to flow through resistor R4. The transistor load including transistor M6 and transistor M7 mirror that current and cause the mirrored current to flow back into transistor M5, resulting in positive feedback. To ensure a stable operating point, transistor M8 provides negative feedback from the drain of transistor M5 to the gate of transistor M5. The topology of FIG. 2 is exemplary only, and other VGS/R voltage reference topologies may provide circuits having lower headroom constraints, but higher noise.
Still referring to FIG. 2, voltage reference generator 200 forces the voltage across resistor R4, which has substantially no temperature coefficient to be equal or approximately equal to the zero temperature coefficient gate-to-source voltage VGS of transistor M5. Although this circuit has lower thermal noise as compared to the circuit of FIG. 1, voltage reference generator 200 is sensitive to process variations since the reference voltage may be affected by the threshold voltage, resistance, mobility, oxide capacitance, and dimensions of transistor M5. In general, the threshold voltage of a metal-oxide-semiconductor field effect transistor (MOSFET) is particularly sensitive to process variations. Process variations may be addressed by trimming resistors R2 and R3 to adjust the ratio of R2/R3 to gain up gate voltage Vg or by using a programmable width transistor M5, varying the M6/M7 transistor ratio, or varying the resistance of resistor R1 or combination thereof.
Similarly, thermal variations may be addressed by trimming transistor M5 or trimming transistor M6 and transistor M7 to adjust the transistor ratio M6/M7, or by trimming resistor R1. In addition, since gate-to-source voltage VGS≈0.5V in a typical semiconductor manufacturing technology and the reference voltage is sensitive to loading at node 205, buffer circuit 203 may be required to generate a greater reference voltage level or to reduce sensitivity to load 108. Thus, operational amplifier 206 is coupled in series with node 205. Once the overdrive voltage is set to cancel the temperature coefficient of the threshold voltage variation, resistors R2 and R3 may be adjusted to provide sufficient gain to achieve a constant, target reference voltage (e.g., Vout). Buffer circuit 203 uses less voltage gain (e.g., 2×gain) than that provided by operational amplifier 104 of voltage reference circuit 100 of FIG. 1 (e.g., 10×gain). Accordingly, buffer circuit 203 contributes less noise than operational amplifier 104.
The two-stage topology of voltage reference generators 100 and 200 of FIGS. 1 and 2 allows for core circuit 101 and core circuit 201, respectively, to be designed independently from the loading conditions. Although the two-stage topology reduces design complexity, it increases noise contributions to the buffered reference signal and may result in a substantial impact on performance. Thus, operational amplifier 206 may be designed to achieve a target noise level, but as a result, consumes more power and area than desirable.
Referring to FIG. 3, the topology of voltage reference generator and load regulator 300 has a low output impedance and generates a low noise voltage reference signal with increased power supply rejection (PSR), lower power consumption, and less area than the voltage reference generators 100 and 200 of FIGS. 1 and 2. Referring back to FIG. 3, voltage reference generator and load regulator 300 embeds load regulation within the voltage reference generator circuit. Although some embodiments of the voltage reference generator and load regulator include an operational transconductance amplifier in level-shifting circuit 302, noise from the core circuit 301 dominates the noise performance, which may substantially reduce overall power consumption for a given on-chip regulated supply (e.g., at least by a factor of four) with a negligible impact on noise performance as compared to the topologies of voltage reference generators 100 and 200 of FIGS. 1 and 2. In addition, internal self-regulation provided by level-shifting circuit 302 of FIG. 3 improves the power supply rejection of voltage reference generator 300 over the topologies of FIGS. 1 and 2.
Referring to FIGS. 1, 2, and 3, voltage reference generator and load regulator 300 includes core circuit 301 having a VGS/R topology that is indirectly coupled to a power supply node via buffer circuit 303. Voltage reference generator and load regulator 300 includes level-shifting circuit 302 that shifts a voltage level on an internal, high gain node, rather than amplifying a voltage on a high-impedance node as in voltage reference generator 200. Voltage reference generator and load regulator 300 regulates output reference node 305 to be a low impedance node that may drive a load without sourcing current from core circuit 301, and thus without substantially affecting operation of core circuit 301. Level shifting circuit 302 shifts voltage Vint on node 306, a high-gain node of core circuit 301, to generate output reference voltage Vref,buf, which supplies core circuit 301. Voltage reference generator and load regulator 300 sources current from buffer circuit 303 instead of from core circuit 301 and utilizes the gain of core circuit 301 at node 306 to feedback a level-shifted version of the signal to an input of buffer circuit 303.
Buffer circuit 303 regulates the power supply voltage and supplies current to a load while only causing small changes in the voltage on node 306. Voltage variation at node 306 is level-shifted and used to control buffer circuit 303 to increase its output current to drive the load while rejecting power supply variation. Unlike core circuit 101 and core circuit 201, core circuit 301 is not directly coupled to the external power supply node. Instead, buffer circuit 303 protects core circuit 301 from external supply surges. By eliminating buffer circuit 103 and buffer circuit 203 and the associated voltage multiplication in series with the core circuit 101 and core circuit 201, respectively, to generate output reference voltage Vref,buf, voltage reference generator and load regulator 300 has improved noise performance as compared to voltage reference generator of 100 and voltage reference generator of 200.
When output reference voltage Vref,buf increases, voltage Vint decreases, providing negative feedback and decreases voltage Vh. Transistor M11 decreases output reference voltage Vref,buf to stabilize the voltage on the output node. Variations in output reference voltage Vref,buf appear across R13 and are sensed by transistor M9, which amplifies those variations and feeds back to the input terminal of buffer circuit 303. The voltage change from Vint to Vh configures buffer circuit 303 as a voltage source. The feedback provided by buffer circuit 303 reduces the equivalent impedance at node 305. In at least one embodiment, level-shifting circuit 302 increases voltage Vint by approximately the gate-to-source voltage VGS of transistor M12 (e.g., 0.5V) and an additional amount, to provide headroom for transistor M10 (e.g., 100-200 mV) for an exemplary voltage level shift of at least 0.6-0.7 V to ensure that node 306 provides a high impedance point and adequate gain. The current mirror formed by transistors M10 and M11 provides some gain, but the gain of transistor M9 exceeds the positive feedback provided by gain of the current mirror, thus the negative feedback dominates and provides gain from the output node 305 to node 306.
Still referring to FIG. 3, if process variations cause the threshold voltage of M11 to decrease, those process variations typically cause the threshold voltage of M9 to increase. Use of the overdrive voltages of M9 and M12 provides positive temperature dependence and device ratios may be altered to obtain target temperature performance. In applications where an inverter circuit is coupled to node 305, the current consumed by the inverter circuit at different process corners would track current consumption of core circuit 301. Various circuit parameters may be varied to alter the temperature response or output reference voltage of circuit 300. For example, resistor R13 may have a variable resistance. In at least one embodiment, the gate of transistor M11 is coupled to a selectable tap of resistor R13 to modify the temperature coefficient to blend gate-to-source voltage VGS of transistor M12 and gate-to-source voltage VGS of transistor M9 that level shifts to increase the voltage or decrease the voltage. In addition, the current mirror ratio or the width or length of transistor M9 may be varied, or a combination thereof.
Voltage reference generator and load regulator 300 uses the internal gain of core circuit 301 having a VGS/R circuit topology to gain a signal on an internal node rather than buffering an output of a core circuit of the voltage reference generator in series with the output reference node. In addition, rather than coupling the core circuit directly to the external supply, voltage reference generator and load regulator 300 sources current from a buffer circuit coupled to the external supply. Accordingly, voltage reference generator and load regulator 300 has the ability to deliver load current without substantially affecting core circuit 301.
Referring to FIGS. 4 and 5, in some embodiments, the voltage reference generator and load regulator includes an active level-shifting circuit. For example, voltage reference generator and load regulators 400 and 500 each include operational transconductance amplifier 402 as the level shifting circuit. Operational transconductance amplifier 402 is configured to provide a level-shifted voltage Vh that controls buffer circuit 303 to force a voltage difference between voltage Vint and voltage Vg to zero. In an exemplary low supply voltage application, operational transconductance amplifier 402 receives a power supply voltage from an on-chip charge pump supply, which may be different from the source of the power supply voltage received by buffer circuit 303. By coupling operational transconductance amplifier 402 with the high gain node in a feedback path to the buffer circuit, any operational transconductance amplifier noise is within the feedback loop and is suppressed by the intrinsic gain of core circuit 301. The core reference may be designed for a particular temperature coefficient or target output voltage level or may be trimmed after manufacture for a particular temperature coefficient (e.g., using non-volatile memory or input from a user interface to select the value) for a particular temperature coefficient or output voltage. Referring to FIG. 5, in at least one embodiment voltage reference generator 500 includes variable resistors R13 and R14 that may be selectively configured to have particular resistances for such adjustments. Referring to FIG. 9, in at least one embodiment, voltage reference generator 900 includes resistors R16 and R17 having a selectable tap-off point controlled by select circuit 902 to achieve a particular resistance for such adjustments. In addition, one or more of transistors M10 and M11 may have selectable W/L.
Referring back to FIG. 3, in some embodiments, level-shifting circuit 302 includes a passive level-shifting circuit to generate the level-shifted voltage Vh. Referring to FIG. 6, voltage reference generator and load regulator 600 includes a switched capacitor circuit controlled by first and second clock phases ϕ1 and ϕ2, which signal alternating time intervals, that are used to store charge on capacitor CLS during a first time interval and to provide that stored charge to integrating capacitor Ch during a second time interval to generate level-shifted voltage Vh. Voltage reference generator 600 forces the gate-to-source voltage drop of transistor M9 to be the difference between level-shifted voltage Vh and voltage Vint. Such embodiments may have little voltage headroom, but may operate sufficiently if VGS of M9 is greater than the combined voltage of the gate-to-source voltage of transistor M12 and the overdrive voltage of transistor M10.
Referring to FIG. 7, in another embodiment of a voltage reference generator and load regulator including passive level shifting, first and second clock phases ϕ1 and ϕ2 control a switched capacitor circuit using alternating time intervals to generate level-shifted voltage Vh. Capacitor CLS is configured to store charge during a first time interval and is configured to provide that stored charge to integrating capacitor Ch during a second time interval to control buffer circuit 303, which forces output voltage level Vref,buf to be the difference between level-shifted voltage Vh and voltage Vint. If transistors M9 and M12 have similar current densities and the same overdrive voltage, core circuit 301 will have enough gain to supply current to a load without affecting the operation of core circuit 301 and the output voltage level Vref,buf.
Referring to FIGS. 3-7, in other embodiments of voltage reference generator and load regulators 300, 400, 500, 600, and 700, the current mirror including transistors M10 and M11 may be a cascode current mirror including additional transistors configured to be in a saturation region of operation and coupled to transistors M10 and M11. In other embodiments, complementary versions of voltage reference generator and load regulators 300, 400, 500, 600, and 700 may be used. For example, n-type transistors are replaced with p-type transistors and p-type transistors are replaced with n-type transistors, as illustrated in FIG. 8. Voltage reference generator and load regulator 800 includes buffer circuit 803, which may include a common source p-type device buffer, coupled between an n-type current mirror and a ground reference node. The n-type current mirror formed by transistors M17 and M18 is configured to develop gate-to-source voltage VGS across resistor R15, which is coupled to a power supply node (e.g., VDD). Level-shifting circuit 802 is configured to drive the buffer circuit 803 based on a voltage developed on a node coupled to the drains of transistors M16 and M17.
Thus, embodiments of a voltage reference generator and load regulator that utilizes internal gain and feedback to generate a low-noise output with reduced sensitivity to power supply variations and loading have been described. While circuits and physical structures have been generally presumed in describing embodiments of the invention, it is well recognized that in modern semiconductor design and fabrication, physical structures and circuits may be embodied in computer-readable descriptive form suitable for use in subsequent design, simulation, test or fabrication stages. Structures and functionality presented as discrete components in the exemplary configurations may be implemented as a combined structure or component. Various embodiments of the invention are contemplated to include circuits, systems of circuits, related methods, and tangible computer-readable medium having encodings thereon (e.g., VHSIC Hardware Description Language (VHDL), Verilog, GDSII data, Electronic Design Interchange Format (EDIF), and/or Gerber file) of such circuits, systems, and methods, all as described herein, and as defined in the appended claims. In addition, the computer-readable media may store instructions as well as data that can be used to implement the invention. The instructions/data may be related to hardware, software, firmware or combinations thereof.
The description of the invention set forth herein is illustrative, and is not intended to limit the scope of the invention as set forth in the following claims. Variations and modifications of the embodiments disclosed herein, may be made based on the description set forth herein, without departing from the scope and spirit of the invention as set forth in the following claims.
Patent Grant
10296026
