Low-power inverter for crystal oscillator buffer or the like

Abstract

A low-power inverter (53) reduces power consumption over known inverter designs and is especially well-adapted for serving as a buffer in a Pierce crystal oscillator with a large load capacitance. The inverter (53) includes P- and N-side source-follower stages (310, 320) driving CMOS output transistor pairs (350, 360). The source followers are current-limited through current sources (311, 313, 321, 323) which are biased by a stable reference voltage such as a bandgap reference voltage. Clamping devices (331, 332) are provided to limit the voltages on the gates of the output transistors (350, 360), thereby limiting maximum currents thereof. In addition, a helper device (332) is connected to the gate of a P-channel output transistor (350). The P-channel output transistor (350) typically has a large gate area and thus a large capacitance, and the helper device (332) quickly increases the voltage at the gate when an input signal changes to a high voltage.

Description

CROSS-REFERENCE TO RELATED APPLICATION

Related subject matter is contained in my copending application, Ser. No. 08/111,793, filed concurrently herewith, entitled "Method and Apparatus for Reducing Jitter and Improving Testability of an Oscillator" and assigned to the assignee hereof.

1. Field of the Invention

This invention relates generally to electrical circuits, and more particularly, to inverters for crystal oscillator buffers or the like.

2. Background of the Invention

Crystal oscillators find many applications in clocked analog and digital systems. In general, crystal oscillators use a crystal's resonant frequency as an accurate reference to generate a clock output signal with a desired frequency. One type of crystal oscillator, known as a pulled-crystal oscillator, is useful in systems in which the nominal clock output signal frequency is fixed but the actual frequency may vary by a relatively-small amount. An example in which a pulled-crystal oscillator is useful is a telecommunications system in which the far end provides data at a nominal rate which may vary by a certain amount, but does not provide a separate clock signal. The pulled-crystal oscillator uses the crystal's resonant frequency to establish a baseline frequency at the nominal frequency. The pulled-crystal oscillator then varies a load capacitance on the crystal's terminals to "pull" the frequency by relatively-small amounts.

In a typical crystal oscillator, each terminal of the crystal is capacitively-loaded, and a resistor and an inverter are connected between the crystal terminals to ensure oscillation. This so-called Pierce oscillator is common in integrated circuits and typically uses a complementary metal-oxide-semiconductor (CMOS) inverter. While generally reliable and robust, the CMOS Pierce oscillator can have high power consumption caused by overlap currents conducted in the inverter's P- and N-channel transistors during switching when using a conventional 5-volt power supply. When used in a pulled-crystal oscillator with its additional capacitance, the power consumption of the Pierce oscillator is especially troublesome because the P- and N-channel transistors must be larger to ensure oscillation under the increased load. Thus what is needed is an inverter which can drive large loads with low power consumption for applications such as Pierce oscillators and the like.

SUMMARY OF THE INVENTION

Accordingly, the present invention provides, in one form, a low-power inverter for a crystal oscillator buffer or the like, including an input node for receiving an input signal, an output node for providing an output signal, a first P-channel transistor, a first N-channel transistor, a first source follower stage, and a first clamp portion. The first P-channel transistor has a source coupled to a first power supply voltage terminal, a gate coupled to the input node, and a drain coupled to the output node for providing the output signal. The first N-channel transistor has a drain coupled to the output node, a gate, and a source coupled to a second power supply voltage terminal. The first source follower stage has an input terminal coupled to the input node, and an output terminal coupled to the gate of the first N-channel transistor. The first clamp portion is coupled to the gate of the first N-channel transistor and to the second power supply voltage terminal, and clamps a voltage at the gate of the first N-channel transistor, thereby limiting a drain-to-source current of the first N-channel transistor.

In another form, the present invention provides an oscillator having a first node for being coupled to a first terminal of a crystal, a second node for being coupled to a second terminal of a crystal, first and second power supply voltage terminals, a resistor, first and second capacitors, a first P-channel transistor, a first N-channel transistor, first and second source follower stages, and first and second clamp portions. The resistor has a first terminal coupled to the first node, and a second terminal coupled to the second node. The first capacitor has a first terminal coupled to the first node, and a second terminal coupled to the second power supply voltage terminal. The second capacitor has a first terminal coupled to the second node, and a second terminal coupled to the second power supply voltage terminal. The first P-channel transistor has a source coupled to the first power supply voltage terminal, a gate, and a drain coupled to the second node. The first N-channel transistor has a drain coupled to the second node, a gate, and a source coupled to a second power supply voltage terminal. The first source follower stage has an input terminal coupled to the first node, and an output terminal coupled to the gate of the first N-channel transistor. The first clamp portion is coupled to the gate of the first N-channel transistor and to the second power supply voltage terminal, and clamps a voltage at the gate of the first N-channel transistor, thereby limiting a drain-to-source current of the first N-channel transistor. The second source follower stage has an input terminal coupled to the first node, and an output terminal coupled to the gate of the first P-channel transistor. The second clamp portion is coupled to the gate of the first P-channel transistor and to the first power supply voltage terminal, and clamps a voltage at the gate of the first P-channel transistor, thereby limiting a drain-to-source current of the first P-channel transistor.

These and other features and advantages will be more clearly understood from the following detailed description taken in conjunction with the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates in partial logic diagram and partial schematic diagram form a pulled-crystal oscillator according to the prior art.

FIG. 2 illustrates in partial block diagram, partial logic diagram, and partial schematic diagram form a pulled-crystal oscillator using a low-power inverter according to the present invention.

FIG. 3 illustrates in partial block diagram and partial schematic diagram form one of the CDACs of FIG. 2.

FIG. 4 illustrates in schematic form one of the TEST CDACs of FIG. 2.

FIG. 5 illustrates partial block diagram, partial logic diagram, and partial schematic diagram form a portion of the pulled-crystal oscillator of FIG. 2 useful in understanding a test mode thereof.

FIG. 6 illustrates a timing diagram of the pulled-crystal oscillator of FIG. 2 during the test mode.

FIG. 7 illustrates in partial logic diagram and partial schematic diagram form a portion of the pulled-crystal oscillator of FIG. 2 useful understanding an advantage thereof in normal operation mode.

FIG. 8 illustrates in schematic form a crystal oscillator using an inverter according to the prior art.

FIG. 9 illustrates in schematic form the inverter of FIG. 2.

DETAILED DESCRIPTION OF A PREFERRED EMBODIMENT

FIG. 1 illustrates in partial logic diagram and partial schematic diagram form a pulled-crystal oscillator 20 according to the prior art. Oscillator 20 includes generally a crystal 21, a resistor 22, an inverter 23, a first capacitor array 24, a second capacitor array 25, and a coupling circuit 26. Crystal 21 has first and second terminals. Resistor 22 has a first terminal connected to the first terminal of crystal 21, and a second terminal connected to the second terminal of crystal 21. Inverter 23 has a first terminal connected to the first terminal of crystal 21, and a second terminal connected to the second terminal of crystal 21.

Capacitor array 24 is connected to the first terminal of crystal 21 and includes N capacitors switched by a corresponding one of input signals labelled "D.sub.0 -D.sub.N-1 ". Each leg of capacitor array 24 includes a capacitor and a switch, in series between the first terminal of crystal 21 and a power supply voltage terminal labelled "V.sub.SS ". V.sub.SS is a more-negative power supply voltage terminal having a nominal potential of approximately zero volts. An illustrative leg is shown including a capacitor 30, and a switch 31. Capacitor 30 has a first terminal connected to the first terminal of crystal 21, and a second terminal. Switch 31 has a first terminal connected to the second terminal of capacitor 30, and a second terminal connected to V.sub.SS, and is opened and closed by a digital control signal labelled "D.sub.i ", where i corresponds to the significance of the capacitor. Capacitor array 24 also includes a capacitor 32 having a first terminal connected to the first terminal of crystal 21, and a second terminal connected to V.sub.SS.

Capacitor array 25 is connected to the second terminal of crystal 21 and includes N capacitors switched by a corresponding one of input signals D.sub.0 -D.sub.N-1. Like array 24, each leg of capacitor array 25 includes a capacitor and a switch, in series between the second terminal of crystal 21 and V.sub.SS. An illustrative leg is shown including a capacitor 34, and a switch 35. Capacitor 34 has a first terminal connected to the second terminal of crystal 21, and a second terminal. Switch 35 has a first terminal connected to the second terminal of capacitor 32, and a second terminal connected to V.sub.SS, and is opened and closed by signal D.sub.i. Capacitor array 25 also includes a capacitor 36 having a first terminal connected to the second terminal of crystal 21, and a second terminal connected to V.sub.SS.

Coupling circuit 26 includes an inverter 40, a resistor 41, a capacitor 42, and inverters 43 and 44. Inverter 40 has an input terminal, and an output terminal connected thereto. Resistor 41 has a first terminal connected to the output terminal of inverter 40, and a second terminal. Capacitor 42 has a first terminal connected to the second terminal of crystal 21, and a second terminal connected to the second terminal of resistor 41. Inverter 43 has an input terminal connected to the second terminals of resistor 41 and capacitor 42, and an output terminal. Inverter 44 has an input terminal connected to the output terminal of inverter 43, and an output terminal for providing a signal labelled "CLOCK OUTPUT SIGNAL".

Oscillator 20 is a conventional pulled-crystal oscillator which has the problems of open capacitor testing, CLOCK OUTPUT SIGNAL jitter, and circuit design as mentioned previously. In particular, coupling circuit 26 converts a sinusoidal waveform into a clean, digital square wave. The circuit design choices used for coupling circuit 26 affect power consumption and duty cycle distortion. Coupling circuit 26 includes self-biased complementary metal-oxide-semiconductor (CMOS) inverter 40 in order to keep the input of inverter 43 near its switchpoint for fast switching. This self-biasing increases power consumption in inverter 40 because the P- and N-channel transistors are always somewhat conductive. Furthermore, in order to reduce power, it is desirable to make inverter 40 weak; however, if inverter 40 is made too weak, it no longer matches the threshold characteristics of inverter 43. The result is that the input of inverter 43 is biased above or below its switchpoint, resulting in clock duty cycle distortion. Coupling circuit 26 is also sensitive to power supply noise and noise that is alternating current (AC) coupled through capacitor 42.

FIG. 2 illustrates in partial schematic diagram, partial logic diagram, and partial block diagram form a pulled-crystal oscillator 50 using a low-power inverter 53 according to the present invention. Pulled-crystal oscillator 50 includes generally a crystal 51, a resistor 52, inverter 53, multiplexers 54 and 55 each labelled "MUX", a comparator 56, a first capacitor digital-to-analog converter (CDAC) labelled "CDAC A" 57, a second CDAC labelled "CDAC B" 58, a decoder 59, and a testing circuit 60. Crystal 51 has a first terminal labelled "XTAL.sub.IN ", and a second terminal labelled "XTAL.sub.OUT ". Resistor 52 has a first terminal connected to XTAL.sub.IN, and a second terminal connected to XTAL.sub.OUT. Inverter 53 has an input terminal connected to XTAL.sub.IN, and an output terminal connected to XTAL.sub.OUT. MUX 54 has a first input terminal connected to a node labelled "TEST A", a second input terminal connected to XTAL.sub.IN, a control input terminal for receiving a signal labelled "TEST MODE", and an output terminal. MUX 55 has a first input terminal connected to a node labelled "TEST B", a second input terminal connected to XTAL.sub.OUT, a control input terminal for receiving signal TEST MODE, and an output terminal. Comparator 56 has a negative input terminal connected to the output terminal of MUX 54, a positive input terminal connected to the output terminal of MUX 55, and an output terminal for providing CLOCK OUTPUT SIGNAL.

CDAC 57 is connected to XTAL.sub.IN, to the first terminal of MUX 54 at node TEST A, and to V.sub.SS, and receives an N-bit digital code labelled "DA.sub.0 -DA.sub.N-1 " at an input terminal thereof and a signal labelled "SELECT A" at a control input terminal thereof. CDAC 58 is connected to XTAL.sub.OUT, to the second terminal of MUX 55 at node TEST B, and to V.sub.SS, and receives an N-bit digital code labelled "DB.sub.0 -DB.sub.N-1 " at an input terminal thereof and a signal labelled "SELECT B" at a control input terminal thereof. The differences between CDACs 57 or 58 and capacitor arrays 24 or 25 of FIG. 1 are understood by referring now to FIG. 3, which illustrates in partial block diagram and partial schematic diagram form CDAC 57. CDAC 57 includes N capacitors each selectively switching a terminal of the corresponding capacitor between V.sub.SS and TEST A. For illustrative purposes, CDAC 57 is described with reference to the i-th leg, which includes a capacitor 80, N-channel transistors 81 and 82, and a latch 83. Capacitor 80 has a first terminal connected to XTAL.sub.IN, and a second terminal. Transistor 81 has a drain connected to the second terminal of capacitor 80, a gate for receiving a signal labelled "S.sub.i ", and a source connected to V.sub.SS. Transistor 82 has a drain connected to the second terminal of capacitor 80, a gate for receiving a signal labelled "T.sub.i ", and a source connected to node TEST A. Latch 83 has a data input terminal for receiving signal DA.sub.i, a control input terminal for receiving signal SELECT A, a true output terminal for providing signal S.sub.i, and a gated complementary output terminal for providing signal T.sub.i.

CDAC 57 also includes a capacitor 85 and a resistor 86. Capacitor 85 has a first terminal connected to XTAL.sub.IN, and a second terminal connected to V.sub.SS. Capacitor 85 is a default capacitance which is always present and sets the basic frequency from which the frequency of oscillation is "pulled". Resistor 86 has a first terminal connected to node TEST A, and a second terminal connected to V.sub.SS. Resistor 86 is a high-valued resistor implemented as either a diffused resistor, or preferably a long-channel MOS transistor which may be made nonconductive when not needed. The advantage of resistor 86 will be described subsequently with reference to FIG. 7.

Signal SELECT A is active either during normal operation or when CDAC 57 is being tested. When signal SELECT A is inactive at a logic low, latch 83 functions to provide signal Si in response to signal DA.sub.i, and keeps signal Ti in an inactive state at a logic low. Thus, capacitor 81 will be selectively connected to V.sub.SS through transistor 81, based on the value of corresponding input DA.sub.i. When signal SELECT A is active at a logic high, latch 83 provides signals S.sub.i and T.sub.i as complementary output signals. Thus, if DA.sub.i =0, then signal S.sub.i is inactive at a logic low keeping transistor 81 nonconductive, and signal T.sub.i is active at a logic high to make transistor 82 conductive and to connect the second terminal of capacitor 80 to node TEST A. If DA.sub.i =1, then signal S.sub.i is active at a logic high making transistor 81 conductive, coupling capacitor 80 between XTAL.sub.IN and V.sub.SS. Signal T.sub.i is inactive at a logic low to keep transistor 82 nonconductive. Latch 83 makes signals S.sub.i and T.sub.i nonoverlapping with respect to each other in order to avoid momentary leakage from the first input terminal of MUX 54 through transistors 82 and 81 to V.sub.SS. Note that CDAC 58 is structurally identical to switched capacitor array 57 and in the following discussion, corresponding elements in CDAC 58 will be described by using the same reference number.

Returning now to FIG. 2, decoder 59 has an input terminal for receiving an m-bit digital input signal from a digital loop filter (not shown), and an output terminal for providing N-bit digital code D.sub.0 -D.sub.N-1. Testing circuit 60 includes a CDAC 61 labelled "TEST CDAC B", a CDAC 62 labelled "TEST CDAC A", and a scan path formed by scannable data latches 63-65. TEST CDAC 61 is connected to node TEST A, and receives signal TEST MODE, a signal labelled "TEST ENABLE B", and a signal labelled "SCAN ENABLE" at control input terminals thereof. TEST CDAC 62 is connected to node TEST B, and receives signals TEST MODE and SCAN ENABLE and a signal labelled "TEST ENABLE A" at control input terminals thereof. TEST CDACS 61 and 62 are understood with reference to FIG. 4, which illustrates in schematic form TEST CDAC 61 of FIG. 2.

TEST CDAC 61 includes a P-channel transistor 90, a capacitor 91, N-channel transistors 93--95, and a capacitor 96. Transistor 90 has a source connected to a power supply voltage terminal labelled "V.sub.DD ", a gate for receiving a complement of signal SCAN ENABLE labelled "SCAN ENABLE", and a drain connected to TEST A. V.sub.DD is a more-positive power supply voltage terminal representative of a logic high voltage and having a nominal voltage of approximately 5.0 volts. Signal SCAN ENABLE is preferably generated globally by inverting signal SCAN ENABLE and providing the resulting signal to each of TEST CDACs 61 and 62, but this function was omitted from FIG. 2 to ease discussion. Capacitor 91 has a first terminal connected to the drain of transistor 90, and a second terminal. Transistor 93 has a drain connected to the second terminal of capacitor 91, a gate for receiving signal TEST MODE, and a source connected to V.sub.SS. Transistor 94 has a drain connected to the drain of transistor 90, a gate for receiving signal TEST ENABLE B, and a source. Transistor 95 has a drain connected to the source of transistor 94, a gate for receiving signal SCAN ENABLE, and a source connected to V.sub.SS. Capacitor 96 has a first terminal connected to the source of transistor 94, and a second terminal connected to V.sub.SS.

When VCO 50 is in normal operation mode, signals TEST MODE, SCAN ENABLE, SCAN ENABLE, and TEST ENABLE B are inactive at their respective logic levels. During normal operation mode, transistors 90, 93, 94, and 95 are inactive and capacitors 91 and 96 are floating. During test mode, signal TEST MODE is active at a logic high, making transistor 93 conductive to connect the second terminal of capacitor 91 to V.sub.SS. When test data is being shifted into the scan path, signal SCAN ENABLE becomes active, precharging the first terminal of capacitor 91 to V.sub.DD. Signal SCAN ENABLE also becomes active, discharging the first terminal of capacitor 96 to V.sub.SS. Signal TEST ENABLE B remains inactive, isolating the first terminal of capacitor 96 from node TEST A. When the scan data has been properly shifted into the scan path, signals SCAN ENABLE and SCAN ENABLE again become inactive for testing, but by this time the first terminal of capacitor 91 has been precharged to approximately V.sub.DD.

During the actual test, signal TEST ENABLE B becomes active, connecting the first terminal of capacitor 96 to node TEST A, and causing capacitor 91 to discharge into capacitor 96. In the preferred embodiment, capacitors 91 and 96 are equal-valued. In the absence of any parasitic capacitance, the voltage on node TEST A should approach approximately V.sub.DD /2 or about 2.5 volts. However, the parasitic capacitance on node TEST A also stores charge during the precharge period, increasing the capacitance with which capacitor 96 shares charge. Thus, the voltage on node TEST A will be above V.sub.DD /2. This voltage is provided through MUX 54 of FIG. 2 to the negative input terminal of comparator 56 to provide a comparison voltage, or offset, against which CDAC 58 is tested. The structure and operation of TEST CDAC 62, which is used when testing CDAC 57, is identical except that TEST CDAC 62 is connected to node TEST B, and a transistor corresponding to transistor 94 receives signal TEST ENABLE A at its gate.

Returning now to FIG. 2, scannable data latch 63 has a parallel input terminal connected to the output terminal of decoder 59 for receiving signals D.sub.0 -D.sub.N-1, a parallel clock input terminal for receiving a signal labelled "PARALLEL CLOCK", a serial data input terminal for receiving a signal labelled "SCAN DATA IN", a scan clock input terminal for receiving a signal labelled "SCAN CLOCK", a control input terminal for receiving signal SCAN ENABLE, a parallel output terminal for providing signals DA.sub.0 -DA.sub.N-1 to capacitor array 54, and a serial data output terminal. Signal PARALLEL CLOCK is a high-speed clock which is some sub-multiple of CLOCK OUTPUT SIGNAL and which switches synchronously therewith. In the illustrated embodiment, PARALLEL CLOCK has a frequency equal to one-half the frequency of CLOCK OUTPUT SIGNAL.

Scannable data latch 64 has a parallel input terminal connected to the output terminal of comparator 56, a serial data input terminal connected to the serial data output terminal of scannable data latch 63, a scan clock input for receiving signal SCAN CLOCK, a control input terminal for receiving signal SCAN ENABLE, and a serial data output terminal. Scannable data latch 65 has a parallel input terminal connected to the output terminal of decoder 59 for receiving signals D.sub.0 -D.sub.N-1, a parallel clock input terminal for receiving signal PARALLEL CLOCK, a serial data input terminal connected to the serial data output terminal of scannable data latch 64, a scan clock input signal for receiving signal SCAN CLOCK, a control input terminal for receiving signal SCAN ENABLE, a parallel output terminal for providing signals DB.sub.0 -DB.sub.N-1 to capacitor array 58, and a serial data output terminal for providing a signal labelled "SCAN DATA OUT".

In overall operation, pulled-crystal oscillator 50 has two modes of operation, normal operation mode and test mode. During normal operation mode, pulled-crystal oscillator 50 provides low signal jitter and avoids difficult circuit design tradeoffs, resulting in both low power consumption and a reliable fifty percent duty cycle clock. In normal operation mode, various capacitors in CDACs 57 and 58 are switched into oscillator 50 by digital codes DA.sub.0 -DA.sub.N-1 and DB.sub.0 -DB.sub.N-1, respectively, each of which is the same as digital code D.sub.0 -D.sub.N-1. In other embodiments, digital code DA.sub.0 -DA.sub.N-1 need not be the same as digital code DB.sub.0 -DB.sub.N-1. In test mode, pulled-crystal oscillator 50 provides a scannable testing apparatus which is able to detect open capacitors in a short amount of time. In test mode, digital codes DA.sub.0 -DA.sub.N-1 and DB.sub.0 -DB.sub.N-1 enable selected capacitors in response to a sequence of data bits conducted through latches 63-65 through the SCAN DATA IN signal line and in general, DA.sub.0 -DA.sub.N-1 is not the same as DB.sub.0 -DB.sub.N-1 during test mode.

Normal operation mode is signified by signal TEST MODE being inactive at a logic low. In normal operation mode, testing circuit 60 is disabled, CDACs 57 and 58 are enabled, and MUXes 54 and 55 select the second inputs thereof. In normal operation mode, pulled-crystal oscillator 50 consumes less power than oscillator 20 of FIG. 1 because the inputs of comparator 56 do not need to be biased. Oscillator 50 also provides CLOCK OUTPUT SIGNAL with less signal jitter than oscillator 20 because comparator 56 is a comparator with high common-mode and power-supply rejection, and thus is not susceptible to being biased above or below its switchpoint or to noise-induced changes in its switchpoint.

Test mode is signified by signal TEST MODE being active at a logic high. FIG. 5 illustrates partial block diagram, partial logic diagram, and partial schematic diagram form a portion 150 of pulled-crystal oscillator 50 of FIG. 2 useful in understanding a test mode thereof. Pulled-crystal oscillator 150 includes generally portions of MUXes 54 and 55, comparator 56, CDAC 57, TEST CDAC 62, and scannable latches 63 and 64. It should be understood that while the same reference numbers are used for corresponding elements of FIG. 2, some of the circuitry used has been omitted or simplified.

FIG. 5 illustrates an "A" testing cycle in which capacitors in CDAC 57 are tested. Thus, CDAC 57 and TEST CDAC 62 are operatively connected to comparator 56. MUX 54 disconnects XTAL.sub.IN from the negative input terminal of comparator 56, represented by a switch 151 thereof being open. Another switch 155 is closed to connect XTAL.sub.IN to V.sub.SS. MUX 54 connects node TEST A to the negative input terminal of comparator 56, represented by a switch 153 thereof being closed. Likewise, MUX 55 disconnects XTAL.sub.OUT from the positive input terminal of comparator 56, represented by a switch 161 thereof being open. Another switch 165 is closed to connect XTAL.sub.OUT to V.sub.SS. MUX 55 connects node TEST B to the positive input terminal of comparator 56, represented by a switch 163 thereof being closed.

FIG. 5 shows two illustrative legs of CDAC 57, including a most significant leg with a capacitor 171, N-channel transistors 172 and 173, and an inverter 174. Capacitor 171 has a first terminal connected to XTAL.sub.IN, and a second terminal. Transistor 172 has a drain connected to the second terminal of capacitor 171, a gate for receiving signal DA.sub.N-1, and a source connected to V.sub.SS. Transistor 173 has a drain connected to node TEST A, a gate, and a source connected to the drain of transistor 172. Inverter 174 has an input terminal for receiving signal DA.sub.N-1, and an output terminal connected to the gate of transistor 173.

A capacitor is tested by setting a corresponding scan data bit to a binary zero, and setting all other scan data bits to binary ones. Signal SCAN ENABLE is activated to scan this data in through the SCAN DATA IN signal line synchronously with SCAN CLOCK. Scannable data latch 63 includes N D-type scan flip flops including illustrative flip flops 180 and 181. To test capacitor 171, a binary zero is scanned in through SCAN DATA IN during a first clock, followed by N-1 binary ones. After these N cycles of SCAN CLOCK, the binary zero has propagated into scan flip flop 181. Thus, the Q output terminal of scan flip flop 181 is set to a binary zero, making transistor 172 nonconductive and transistor 173 conductive. The result is that the first terminal of capacitor 171 is connected to V.sub.SS, through switch 152 in MUX 54, and the second terminal of capacitor 171 is connected to the negative input terminal of comparator 56 through node TEST A.

During data input scanning, signal SCAN ENABLE causes transistors 90 of TEST CDACs 61 and 62 to precharge nodes TEST A and TEST B, respectively, to V.sub.DD. These voltages are held at approximately V.sub.DD by a parasitic capacitance 175 on the negative input terminal of comparator 56, and by a parasitic capacitance 176 on the positive input terminal of comparator 56 and by capacitors 91 in TEST CDACs 61 and 62, respectively. After the data has been aligned in scannable data latch 63, signal SCAN ENABLE becomes inactive and transistors 90 of TEST CDACs 61 and 62 are nonconductive. Capacitors 91 in TEST CDAC 62 and 176 discharge into capacitor 96, through a path represented by a switch 94, assigned the same reference number as transistor 94 of FIG. 4, being closed. The first terminal of capacitor 96 had previously been discharged to V.sub.SS, through a path represented now by a switch 95, assigned the same reference number as transistor 95 of FIG. 4, being open. Capacitors 91 and 96 preferably have equal values which are smaller than the smallest capacitor in CDAC 57. Thus, the comparison voltage on the positive input terminal of comparator 56 will be somewhat above mid-supply due to parasitic capacitor 176, and is preferable made about 3.75 volts.

This comparison voltage initially causes comparator 56 to switch low, because the 5.0 volts on the negative terminal of comparator 56 is greater than the 3.75 volts on the positive terminal thereof. However, if capacitor 171 and the associated control circuitry (transistors 172 and 173 and inverter 174) are functional, capacitors 91 and 175 will discharge into capacitor 171, bringing the negative terminal of comparator 56 to a lower voltage than the comparison voltage. This discharging is guaranteed by sizing capacitor 96 to be smaller than the smallest capacitor in CDAC 57 and having approximately equal parasitic capacitances 175 and 176.

FIG. 6 illustrates a timing diagram of pulled-crystal oscillator 50 of FIG. 2 during the test mode. FIG. 6 illustrates an "A" testing cycle, in which a selected capacitor in CDAC 57 is tested. FIG. 6 illustrates four signals, SCAN CLOCK, SCAN ENABLE, TEST ENABLE A, and the output of comparator 56, versus time on the horizontal axis. FIG. 6 also illustrates four intervals relevant to the test sequence. A first interval, labelled "SCAN IN", occurs when data is being read into the scan path. During the SCAN IN interval, signal SCAN ENABLE is active, inhibiting the outputs of scannable data latches 63 and 65 of FIG. 2, and discharging capacitor 96 of FIG. 4 to V.sub.SS. Active signal SCAN ENABLE (not shown in FIG. 6) also causes the input signal lines to comparator 56 to be precharged to approximately V.sub.DD through transistors 90. Signal TEST ENABLE A is inactive, isolating capacitor 96 from node TEST A. Data is input serially and shifted through the scan path synchronously with SCAN CLOCK.

A second interval, labelled "TEST", follows the SCAN IN interval. During the TEST interval, signal SCAN ENABLE becomes inactive, causing scannable data latches 63 and 65 to provide their outputs to CDACs 57 and 58, respectively. Signal TEST ENABLE A becomes active, providing a comparison voltage to the positive input terminal of comparator 56 by causing the charge stored in capacitors 91 and 176 to discharge into capacitor 96. The connection of the selected capacitor in CDAC 57 causes the output of comparator 56 to reflect the outcome of the test during the TEST interval.

During a third interval, labelled "LATCH", the output of comparator 56 is latched into the scan path. Scannable data latch 64 latches the output of comparator 56 on the falling edge of SCAN CLOCK.

A fourth interval labelled "SCAN OUT" follows the LATCH interval. During the SCAN OUT interval, signal SCAN ENABLE is active, again inhibiting the outputs of scannable data latches 63 and 65. Data is output serially and shifted through the scan path synchronously with SCAN CLOCK until the data bit corresponding to the output of comparator 56 is output as signal SCAN DATA OUT. Preferably, a single physical latch implements the function of latches 63 and 65. This sharing of circuitry is possible because during the normal operation mode, corresponding capacitors in CDACs 57 and 58 are activated simultaneously, and during test mode signals SELECT A and SELECT B ensure that only one of CDACs 57 and 58 are tested at any one time. Thus, a single clock is required during the SCAN OUT interval to output the value of comparator 56 latched during the LATCH interval as signal SCAN DATA OUT, keeping test time to a minimum.

In a preferred test sequence, testing of capacitors in CDACs 57 and 58 is preceded by a pretest sequence which tests "A" and "B" control circuitry. For example, in order to test the "A" control circuitry, a sequence of N binary ones is scanned into scannable data latch 63. Since no capacitors in CDAC 57 are connected, capacitor 96 in TEST CDAC 62 should pull the voltage at the positive input terminal of comparator 56 below the voltage at the negative input terminal, causing comparator 56 to output a logic low.

The "A" capacitors are tested one at a time by setting signal SELECT A to a logic high and signal SELECT B to a logic low. Each capacitor is tested in turn by scanning in a binary zero in the corresponding bit position zero, and a binary one in all other bit positions. In the test of CDAC 57, if the i-th capacitor is not open, then it will pull the voltage at the negative input terminal of comparator 56 below that at the positive input terminal thereof. Thus, if the output of comparator 56 is a logic high, then the i-th capacitor is not open. The test of the "B" capacitors in CDAC 58 proceeds in a similar fashion, except that a logic low output of comparator 56 indicates a functional capacitor. It is possible to write control software to either test all elements, or to exit the test after the first defective element is detected. Note that while it is preferred to pretest control circuitry, other test sequences may test both the control circuitry and the CDAC capacitors together.

FIG. 7 illustrates in partial logic diagram and partial schematic diagram form a portion 250 of pulled-crystal oscillator 50 of FIG. 2 useful in understanding an advantage thereof in normal operation mode. In pulled-crystal oscillator 250, crystal 51, resistor 52, inverter 53, and comparator 56 are connected as in FIG. 2. Note that MUXes 54 and 55 are omitted for clarity. CDAC 57 includes capacitor 85, an active capacitor 251, and N-1 inactive capacitors 252. Since it is active, capacitor 251 has a first terminal connected to XTAL.sub.IN, and a second terminal connected to V.sub.SS. Since signal SELECT A is active, each of the N-1 inactive capacitors has a first terminal connected to XTAL.sub.IN, and a second terminal connected to node TEST A. CDAC 58 includes capacitor 85, an active capacitor 253, and N-1 inactive capacitors 254. Since it is active, capacitor 253 has a first terminal connected to XTAL.sub.OUT, and a second terminal connected to V.sub.SS. Since signal SELECT B is active, each of the N-1 inactive capacitors has a first terminal connected to XTAL.sub.OUT, and a second terminal connected to node TEST B.

In normal operation mode, the second terminals of the inactive capacitors are connected to either TEST A or TEST B. Since CDACs 57 and 58 connect these nodes to V.sub.SS through high-valued resistors 86, the inactive capacitors slowly charge up to the average voltage present on the XTAL.sub.IN and XTAL.sub.OUT nodes. The values of resistors 86 must be chosen such that the RC time constant, for a capacitance of the smallest capacitor in CDACs 57 and 58, is much longer than the period of oscillation. The result is that when an inactive capacitor is activated, it is already charged to the average voltage on a corresponding one of nodes XTAL.sub.IN and XTAL.sub.OUT. The switchpoint occurs at the zero-crossing point of the inputs of comparator 56 because new input data is latched by PARALLEL CLOCK, and thus switches synchronously with CLOCK OUTPUT SIGNAL. Since the zero-crossing point occurs near the average voltages on the XTAL.sub.IN and XTAL.sub.OUT nodes, there will be virtually no "glitch" on XTAL.sub.IN and XTAL.sub.OUT when the inactive capacitors are switched in.

FIG. 8 illustrates in schematic form a crystal oscillator 300 using an inverter 305 according to the prior art. Crystal oscillator 300 includes a crystal 301, a resistor 302, capacitors 303 and 304, and inverter 305. Crystal 301 has first and second terminals. Resistor 302 has a first terminal connected to the first terminal of crystal 301, and a second terminal connected to the second terminal of crystal 301. Capacitor 303 has a first terminal connected to the first terminal of crystal 301, and a second terminal connected to V.sub.SS. Capacitor 304 has a first terminal connected to the second terminal of crystal 301, and a second terminal connected to V.sub.SS. Inverter 305 has a first terminal connected to the first terminal of crystal 301, and a second terminal connected to the second terminal of crystal 301. Inverter 305 is a conventional CMOS inverter including a P-channel transistor 306, and an N-channel transistor 307. Transistor 306 has a source connected to V.sub.DD, a gate connected to the first terminal of crystal 301, and a drain connected to the second terminal of crystal 301. Transistor 307 has a drain connected to the drain of transistor 306, a gate connected to the first terminal of crystal 301, and a source connected to V.sub.SS.

Oscillator 300, known as a Pierce oscillator, is commonly used in CMOS integrated circuits. Crystal 301 is an off-chip crystal having a desired resonant frequency. Resistor 302 and capacitors 303 and 304 are normally off-chip because they require too much integrated circuit area in most embodiments, but may be integrated in embodiments requiring relatively-small values or in pulled-crystal oscillators. Transistors 306 and 307 are almost always integrated on-chip using conventional CMOS processes.

Although oscillator 300 is simple and reliable, it has higher power consumption in applications in which V.sub.DD is equal to approximately 5 volts and which require large values for capacitors 303 and 304, such as a pulled-crystal oscillator. Oscillator 300 is a good solution when V.sub.DD is slightly above (V.sub.TN +.vertline.V.sub.TP .vertline.), where V.sub.TN is the gate-to-source threshold voltage of transistor 307, and V.sub.TP is the gate-to-source threshold voltage of transistor 306. However, when V.sub.DD is about 5.0 volts and V.sub.TN and .vertline.V.sub.TP .vertline. are each about 0.8 volts, and transistors 306 and 307 in inverter 305 are sized strongly enough to oscillate with a large capacitive load, inverter 305 will have undesirably large overlap current. It is possible to provide a reduced positive power supply voltage at the source of transistor 306, but the reduced supply will require an additional package pin for a decoupling capacitor, which may not be available in some applications. In addition, it is always desirable to minimize integrated circuit pin count. Thus, a new inverter design is needed in order to reduce power consumption for such applications.

FIG. 9 illustrates in schematic form inverter 53 of FIG. 2. Inverter 53 includes generally an P-side source-follower portion 310, an N-side source-follower portion 320, a P-side gate control portion 330, an N-side gate control portion 340, a P-channel transistor 350, and an N-channel transistor 360.

P-side source-follower portion includes a current source 311, a P-channel transistor 312, and a current source 313. Current source 311 has a first terminal connected to V.sub.DD, and a second terminal, and conducts a current labelled "I.sub.311 ". Transistor 312 has a source connected to the second terminal of current source 311, a gate for receiving an input voltage signal labelled "V.sub.IN ", and a drain. In inverter 53, V.sub.IN is connected to the XTAL.sub.IN terminal of crystal 51, but in other embodiments may be an arbitrary input voltage such as a CMOS logic signal. Current source 313 has a first terminal connected to the drain of transistor 312, and a second terminal connected to V.sub.SS, and conducts a current labelled "I.sub.313 ". Note that as used herein, all current conducting devices are considered to be current "sources", although current source 313 may also be considered a current sink.

N-side source-follower portion 320 includes a current source 321, a N-channel transistor 322, and a current source 323. Current source 321 has a first terminal connected to V.sub.DD, and a second terminal, and conducts a current labelled "I.sub.321 ". Transistor 322 has a drain connected to the second terminal of current source 321, a gate for receiving signal V.sub.IN, and a source. Current source 323 has a first terminal connected to the source of transistor 322, and a second terminal connected to V.sub.SS, and conducts a current labelled "I.sub.323 ".

P-side gate control portion 330 includes P-channel transistors 331 and 332. Transistor 331 has a source connected to V.sub.DD, a gate connected to the source of transistor 312, and a drain connected to the source of transistor 312. Transistor 332 has a source connected to V.sub.DD, a gate connected to the second terminal of current source 321, and a drain connected to the source of transistor 312. N-side gate control portion 340 includes a single N-channel transistor also referred to as 340. Transistor 340 has a drain connected to the source of transistor 322, a gate connected to the source of transistor 322, and a drain connected to V.sub.SS. P-channel transistor 350 has a source connected to V.sub.DD, a gate connected to the source of transistor 312, and a drain for providing a signal labelled "V.sub.OUT ". In inverter 53, V.sub.OUT is connected to the XTAL.sub.OUT terminal of crystal 51, but in other embodiments may be considered a CMOS logic signal. N-channel transistor 360 has a drain connected to the drain of transistor 350, a gate connected to the source of transistor 322, and a source connected to V.sub.SS.

Inverter 53 is very effective in minimizing overlap currents. First, inverter 53 includes source follower transistor pairs 312 and 350, and 322 and 360. These source-follower pairs reduce the available headroom between V.sub.DD and the sum of the thresholds of transistors 350 and 360. Second, transistor 331 in P-side gate control portion 330 and transistor 340 in N-side gate control portion 340 are clamping transistors which limit the gate-to-source voltages seen by output transistors 350 and 360, respectively. The current in P-side gate control portion 330 is limited to the difference in current sources 313 and 311, namely a value of (I.sub.313 -I.sub.311). Likewise, the current in N-side gate control portion 340 is limited to the difference in current sources 321 and 323, namely a value of (I.sub.321 -I.sub.323). Currents I.sub.313 and I.sub.321 must be greater than currents I.sub.311 and I.sub.323, respectively, and these differences and appropriate transistor size ratios determine the maximum currents flowing through transistors 350 and 360, respectively. Specifically, the maximum current through transistor 350 is equal to

I.sub.350 (MAX)=(I.sub.313 -I.sub.311) * (W/L).sub.350 /(W/L).sub.331[ 1] where I.sub.350 (MAX) is the approximate maximum drain current of transistor 350, (W/L).sub.350 is the gate width-to-length ratio of transistor 350, and (W/L).sub.331 is the gate width-to-length ratio of transistor 331. Likewise, the maximum current through transistor 360 is equal to I.sub.360 (MAX)=(I.sub.321 -I.sub.323) * (W/L).sub.360 /(W/L).sub.340[ 2] where I.sub.360 (MAX) is the approximate maximum drain current of transistor 360, (W/L).sub.360 is the gate width-to-length ration of transistor 360, and (W/L).sub.340 is the gate width-to-length ratio of transistor 340.

As long as currents I.sub.311, I.sub.313, I.sub.321, and I.sub.323 are reasonably well-controlled, the output current will be relatively independent of changes in device characteristics due to process, temperature, and power supply variations. Current sources 311 and 321 are preferably formed by P-channel transistors biased by a stable bias voltage, and current sources 313 and 323 are preferably formed by N-channel transistors biased by another stable bias voltage. These bias voltages are preferably generated from a stable reference (not shown) such as from a V.sub.BE /R or .DELTA.V.sub.BE /R generator. As used here, "V.sub.BE " is the forward-biased base-to-emitter diode voltage drop of a reference bipolar transistor, ".DELTA.V.sub.BE " is the difference in V.sub.BE between two bipolar transistors, and "R" is the resistance of a reference resistor. The bias voltages are preferably provided by a bandgap reference voltage generator, providing a .DELTA.V.sub.BE /R reference. Because current I is very stable, transistors 350 and 360 may be made strong (i.e., have large W/L ratios) without having large overlap currents. This strength is desirable when inverter 53 is used in crystal oscillator inverters because it ensures oscillation even with high capacitive loading.

Transistor 332 is a "helper" device that helps to make transistor 350 nonconductive when V.sub.IN switches to a logic high. Being a P-channel device, transistor 350 will normally have a gate width-to-length ratio two to three times greater than that of transistor 360. However this gate sizing provides correspondingly greater capacitance on the source of transistor 312. When voltage V.sub.IN is increasing, the combined effects of clamping transistor 331 and current source 311 may slow the increase in voltage at the gate of transistor 350 so much that a large overlap current in transistors 350 and 360 results. Under these conditions, the voltage at the drain of transistor 322 is falling; thus, helper transistor 332 biased by the voltage at the drain of transistor 322 provides additional current to make transistor 350 nonconductive quicker.

Note that a similar helper device cannot be used to help make transistor 360 nonconductive quicker, because having such a helper device would form a latch structure which could make both transistor 350 and transistor 360 simultaneously nonconductive. However, the smaller capacitance on the gate of transistor 360 reduces the need for a second helper device.

Inverter 53 has greatly reduced power consumption over inverter 300 of FIG. 8 both when V.sub.DD is equal to the standard 5.0 volts and at a reduced value of 3.3 volts. Table I illustrates simulation results when inverters 300 and 53 were used in an oscillator with a 20.48 megahertz (MHz) crystal resonance frequency:

TABLE I______________________________________ Typical Power Maximum PowerCircuit V.sub.DD (V) (mW) (mW)______________________________________Inverter 300 5.0 14.1 39Inverter 300 3.3 9.0 25.6Inverter 53 5.0 6.75 14.85______________________________________ For the purposes of this comparison, "typical" means 25 degrees Celsius, typical processing parameters, and 50 picofarads (pF) on V.sub.IN and V.sub.OUT (connected to nodes XTAL.sub.IN and XTAL.sub.OUT, respectively). "Maximum" means -40 degrees Celsius, best case speed processing (lowest N- and P-channel thresholds, shortest effective channel lengths, etc.), and 100 picofarads (pF) on V.sub.IN and V.sub.OUT. In addition, for each of the three cases, the last-stage transistors were sized as small as possible while still ensuring oscillation under all operating and processing conditions. Thus, inverter 53 significantly reduces power consumption over inverter 300 at 5.0 volts. Inverter 53 also reduces power consumption even when inverter 300 is powered from a reduced-supply voltage at 3.3 volts.

While inverter 53 includes both P- and N-side source follower and clamp circuits, either the P- or the N-side circuitry may be used independently to reduce overlap current. Current sources 313 and 321, connected to the drains of their respective source follower transistors, may be omitted in other embodiments. However, current source 321 is important to the operation of helper transistor 332 and should be used when helper transistor 332 is present.

Referring to FIG. 5 in conjunction with FIG. 9, note that during test mode, both nodes XTAL.sub.IN and XTAL.sub.OUT are connected to V.sub.SS, potentially causing a short circuit through transistor 350 from V.sub.DD to V.sub.SS. This short circuit may be avoided in a variety of ways, such as by disconnecting V.sub.IN and V.sub.OUT from nodes XTAL.sub.IN and XTAL.sub.OUT, respectively, during test mode. Signal V.sub.OUT also may be placed in a high impedance state by connecting the gates of transistors 350 and 360 to V.sub.DD and V.sub.SS, respectively, during the test mode. Preferably, however, the gates of both transistors 350 and 360 are driven to V.sub.DD during test mode so that transistor 360 performs the function of switch 165 to connect node XTAL.sub.OUT to V.sub.SS.

While the invention has been described in the context of a preferred embodiment, it will be apparent to those skilled in the art that the present invention may be modified in numerous ways and may assume many embodiments other than those specifically set out and described above. For example, an inverter according to the present invention may be used with other circuits besides crystal oscillators. Also, the inverter may be used in oscillators other than pulled-crystal oscillators. Accordingly, it is intended by the appended claims to cover all modifications of the invention which fall within the true spirit and scope of the invention.

Claims

1. A low-power inverter for a circuit such as a crystal oscillator buffer, comprising:

an input node for receiving an input signal;

an output node for providing an output signal;

a first P-channel transistor having a source coupled to a first power supply voltage terminal, a gate coupled to said input node, and a drain coupled to said output node for providing said output signal;

a first N-channel transistor having a drain coupled to said output node, a gate, and a source coupled to a second power supply voltage terminal;

a first source follower stage having an input terminal coupled to said input node, and an output terminal coupled to said gate of said first N-channel transistor, said first source follower stage comprising:

a first current source having a first terminal coupled to said first power supply voltage terminal, and a second terminal, for conducting a first predetermined current;

a second N-channel transistor having a drain coupled to said second terminal of said first current source, a gate coupled to said input node, and a source coupled to said gate of said first N-channel transistor; and

a second current source having a first terminal coupled to said source of said second N-channel transistor, and a second terminal coupled to said second power supply voltage terminal, for conducting a second predetermined current; and

first clamp means coupled to said gate of said first N-channel transistor and to said second power supply voltage terminal, for clamping a voltage at said gate of said first N-channel transistor to be less than, a first predetermined voltage greater than a voltage of said second power supply voltage terminal, thereby limiting a drain-to-source current of said first N-channel transistor.

2. The low-power inverter of claim 1 further comprising a second P-channel transistor having a source coupled to said first power supply voltage terminal, a gate coupled to said second terminal of said second current source, and a drain coupled to said gate of said first P-channel transistor.

3. The low-power inverter of claim 1 wherein said first clamp means comprises a third N-channel transistor having a drain coupled to said gate of said first N-channel transistor, a gate coupled to said gate of said first N-channel transistor, and a source coupled to said second power supply voltage terminal.

4. The low-power inverter of claim 1 further comprising:

a second source follower stage having an input terminal coupled to said input node, and an output terminal coupled to said gate of said first P-channel transistor; and

second clamp means coupled to said gate of said first P-channel transistor and to said first power supply voltage terminal, for clamping a voltage at said gate of said first P-channel transistor to be greater than, a second predetermined voltage less than a voltage of said first power supply voltage terminal, thereby limiting a drain-to-source current of said first P-channel transistor.

5. The low-power inverter of claim 4 wherein said second source follower stage comprises:

a first current source having a first terminal coupled to said first power supply voltage terminal, and a second terminal coupled to said gate of said first P-channel transistor, for conducting a first predetermined current; and

a second P-channel transistor having a source coupled to said second terminal of said first current source and to said gate of said first P-channel transistor, a gate coupled to said input node, and a drain coupled to said second power supply voltage terminal.

6. The low-power inverter of claim 5 wherein said second source follower stage further comprises a second current source having a first terminal coupled to said drain of said second P-channel transistor, and a second terminal coupled to said second power supply voltage terminal, for conducting a second predetermined current.

7. The low-power inverter of claim 4 wherein said second clamp means comprises a third P-channel transistor having a source coupled to said first power supply voltage terminal, a gate coupled to said gate of said first P-channel transistor, and a drain coupled to said gate of said first P-channel transistor.

8. A low-power inverter for a circuit such as a crystal oscillator buffer, comprising:

an input node for receiving an input signal;

an output node for providing an output signal;

a first P-channel transistor having a source coupled to a first power supply voltage terminal, a gate, and a drain coupled to said output node for providing said output signal;

a first N-channel transistor having a drain coupled to said output node, a gate coupled to said input node, and a source coupled to a second power supply voltage terminal;

a source follower stage having an input terminal coupled to said input node, and an output terminal coupled to said gate of said first P-channel transistor, said source follower stage comprising:

a first current source having a first terminal coupled to said first power supply voltage terminal, and a second terminal coupled to said gate of said first P-channel transistor, for conducting a first predetermined current;

a second P-channel transistor having a source coupled to said second terminal of said first current source and to said gate of said first P-channel transistor, a gate coupled to said input node for receiving said input signal, and a drain; and

a second current source having a first terminal coupled to said drain of said second P-channel transistor, and a second terminal coupled to said second power supply voltage terminal, for conducting a second predetermined current, and

clamp means coupled to said first power supply voltage terminal and to said gate of said first P-channel transistor, for clamping a voltage at said gate of said first P-channel transistor to be greater than, a predetermined voltage less than a voltage of said first power supply voltage terminal, thereby limiting a drain-to-source current of said first P-channel transistor.

9. The low-power inverter of claim 8 wherein said clamp means comprises a third P-channel transistor having a source coupled to said first power supply voltage terminal, a gate coupled to said gate of said first P-channel transistor, and a drain coupled to said gate of said first P-channel transistor.

10. A low-power inverter for a circuit such as a crystal oscillator buffer, comprising:

an input node for receiving an input signal;

an output node for providing an output signal;

a first P-channel transistor having a source coupled to a first power supply voltage terminal, a gate, and a drain coupled to said output node for providing said output signal;

a first N-channel transistor having a drain coupled to said output node, a gate, and a source coupled to a second power supply voltage terminal;

a first current source having a first terminal coupled to said first power supply voltage terminal, and a second terminal, for conducting a first predetermined current;

a second N-channel transistor having a drain coupled to said second terminal of said first current source, a gate coupled to said input node, and a source coupled to said gate of said first N-channel transistor;

a second current source having a first terminal coupled to said gate of said first N-channel transistor and to said source of said second N-channel transistor, and a second terminal coupled to said second power supply voltage terminal, for conducting a second predetermined current;

a third N-channel transistor having a drain coupled to said gate of said first N-channel transistor, a gate coupled to said gate of said first N-channel transistor, and a source coupled to said second power supply voltage terminal;

a third current source having a first terminal coupled to said first power supply voltage terminal, and a second terminal coupled to said gate of said first P-channel transistor, for conducting a third predetermined current;

a second P-channel transistor having a source coupled to said second terminal of said second current source and to said gate of said first P-channel transistor, a gate coupled to said input node, and a drain coupled to said second power supply voltage terminal; and

a third P-channel transistor having a source coupled to said first power supply voltage terminal, a gate coupled to said gate of said first P-channel transistor, and a drain coupled to said gate of said first P-channel transistor.

11. The low-power inverter of claim 10 further comprising a fourth P-channel transistor having a source coupled to said first power supply voltage terminal, a gate coupled to said second terminal of said first current source, and a drain coupled to said gate of said first P-channel transistor.

12. The low-power inverter of claim 10 further comprising a third current source having a first terminal coupled to said drain of said second P-channel transistor, and a second terminal coupled to said second power supply voltage terminal, for conducting a fourth predetermined current.

13. An oscillator comprising:

a first node for being coupled to a first terminal of a crystal;

a second node for being coupled to a second terminal of a crystal;

first and second power supply voltage terminals;

a first P-channel transistor having a source coupled to said first power supply voltage terminal, a gate, and a drain coupled to said second node;

a first N-channel transistor having a drain coupled to said second node, a gate, and a source coupled to said second power supply voltage terminal;

a first source follower stage having an input terminal coupled to said first node, and an output terminal coupled to said gate of said first N-channel transistor, said first source follower stage comprising;

a first current source having a first terminal coupled to said first power supply voltage terminal, and a second terminal, for conducting a first predetermined current;

a second N-channel transistor having a drain coupled to said second terminal of said first current source, a gate coupled to said first node, and a source coupled to said gate of said first N-channel transistor; and

a second current source having a first terminal coupled to said gate of said first N-channel transistor and to said source of said second N-channel transistor, and a second terminal coupled to said second power supply voltage terminal, for conducting a second predetermined current;

first clamp means coupled to said gate of said first N-channel transistor and to said second power supply voltage terminal, for clamping a voltage at said gate of said first N-channel transistor to be less than, a first predetermined voltage greater than a voltage of said second power supply voltage terminal, thereby limiting a drain-to-source current of said first N-channel transistor;

a second source follower stage having an input terminal coupled to said first node, and an output terminal coupled to said gate of said first P-channel transistor; and

second clamp means coupled to said gate of said first P-channel transistor and to said first power supply voltage terminal, for clamping a voltage at said gate of said first P-channel transistor to be greater than, a second predetermined voltage less than a voltage of said first power supply voltage terminal, thereby limiting a drain-to-source current of said first P-channel transistor.

14. The oscillator of claim 13 further comprising a second P-channel transistor having a source coupled to said first power supply voltage terminal, a gate coupled to said second terminal of said first current source, and a drain coupled to said gate of said first P-channel transistor.

15. The oscillator of claim 13 wherein said first clamp means comprises a third N-channel transistor having a drain coupled to said gate of said first N-channel transistor, a gate coupled to said gate of said first N-channel transistor, and a source coupled to said second power supply voltage terminal.

16. The oscillator of claim 13 wherein said second clamp means comprises a third P-channel transistor having a source coupled to said first power supply voltage terminal, a gate coupled to said gate of said first P-channel transistor, and a drain coupled to said gate of said first P-channel transistor.

17. The oscillator of claim 13 further comprising:

a resistor having a first terminal coupled to said first node, and a second terminal coupled to said second node;

a first capacitor having a first terminal coupled to said first node, and a second terminal coupled to said second power supply voltage terminal; and

a second capacitor having a first terminal coupled to said second node, and a second terminal coupled to said second power supply voltage terminal.

18. A low-power inverter for a circuit such as a crystal oscillator buffer, comprising:

an input node for receiving an input signal;

an output node for providing an output signal;

a first MOS transistor of a first conductivity type having a first current electrode coupled to a first power supply voltage terminal, a gate coupled to said input node, and a second current electrode coupled to said output node for providing said output signal;

a second MOS transistor of a second conductivity type having a first current electrode coupled to said output node, a gate, and a second current electrode coupled to a second power supply voltage terminal;

a first source follower stage having an input terminal coupled to said input node, and an output terminal coupled to said gate of said second MOS transistor, comprising:

a first current source having a first terminal coupled to said first power supply voltage terminal, and a second terminal, for conducting a first predetermined current; and

a third MOS transistor of said second conductivity type having a first current electrode coupled to said second terminal of said first current source, a gate coupled to said input node, and a second current electrode coupled to said gate of said second MOS transistor; and

a fourth MOS transistor of said second conductivity type having a first current electrode coupled to said second current electrode of said third MOS transistor, a gate coupled to said second current electrode of said third MOS transistor, and a second current electrode coupled to said second power supply voltage terminal.

19. The low-power inverter of claim 18 wherein said first source follower stage further comprises a second current source having a first terminal coupled to said second current electrode of said third MOS transistor, and a second terminal coupled to said second power supply voltage terminal, for conducting a second predetermined current.

20. The low-power inverter of claim 18 further comprising a second source follower stage having an input terminal coupled to said input node, and an output terminal coupled to said gate of said first MOS transistor, comprising:

a fifth MOS transistor of said first conductivity type having a first current electrode coupled to said gate of said first MOS transistor, a gate coupled to said input node, and a second current electrode;

a third current source having a first terminal coupled to said second current electrode of said fifth MOS transistor, and a second terminal coupled to said second power supply voltage terminal; and

a sixth MOS transistor having a first current electrode coupled to said first power supply voltage terminal, a gate coupled to said first current electrode of said fifth MOS transistor, and a second current electrode coupled to said first current electrode of said fifth MOS transistor.

21. The low-power inverter of claim 18 wherein said first conductivity type is P-channel and said second conductivity type is N-channel.

22. The low-power inverter of claim 18 wherein said first conductivity type is N-channel and said second conductivity type is P-channel.