Method and apparatus for dynamically adjusting the spectral content of an audio signal

Abstract

An electronic circuit for dynamically adjusting the spectral content of an audio signal. The circuit includes a constant current source, a output buffer amplifier and a biased inductor for introducing controlled amplitude asymmetry. This apparatus thus can be arranged to process an audio signal so as to introduce a predictable and controllable harmonic distortion that is negligible at small signal amplitudes and increases progressively at larger signal amplitudes.

Description

DETAILED DESCRIPTION OF THE INVENTION

As shown in FIG. 5, the basic circuit consists of an input buffer, an output buffer, a constant-current source and a nonlinear element which consists of an inductor. The audio signal is AC-coupled at both ends of the nonlinear element and it is forward-biased by the constant-current source.

The circuit is intentionally unsymmetrical. As the audio signal voltage goes positive the core of the inductor begins to saturate which reduces its impedence at audio frequencies and causes an increase in the instantaneous value of the audio signal at its ouput. When the audio signal goes negative, this does not occur and the resulting asymmetry causes the generation of a monotonic harmonic spectrum.

As shown in FIG. 6, the constant current source in a preferred embodiment is a ring source. Other topologies such as a Widlar current mirror can also be used. The influence of the current source on the circuit operation has been investigated and the ring source has been found to be optimum when implemented with transistors of high beta. This is because it maintains a very high AC impedance over the required frequency range and over the voltage range for which the rest of the circuit is useful. The current value, which is supplied by the constant-current source, is a basic operating parameter of the circuit. For a given range of signal amplitudes, the onset and quantity of harmonic distortion, which is generated, can be adjusted by varying the bias current from the constant-current source. The input buffer of the present invention is shown in FIG. 7. This stage is required in order to define the source impedance, which drives the inductor. Because the operation is based upon an instantaneous signal-dependent impedance change in the inductor, it follows that if the source resistance is too high the desired nonlinearity will be proportionally less and the intended circuit function will be diminished. In a preferred embodiment a source resistance should be held to less than 10 Ohms. If a driving amplifier with sufficiently low source resistance is available then the input buffer could eliminated. The output of the buffer must be AC-coupled to the input of the inductor with the coupling capacitor value large enough to prevent restriction of low frequencies due to the input impedance of the inductor. The exact value of the input impedance depends on the bias current supplied from the constant-current source. Anyone skilled in the art of circuit design will have no difficulty determining the coupling capacitor value.

The output buffer of the present invention is shown in FIG. 8. This stage is required in order to prevent the downstream circuit from placing an undefined load on the inductor. In a preferred embodiment as shown, the buffer is a simple MOSFET source-follower, which is DC-coupled to the output of the inductor. Since the buffer will have a standing DC voltage on its source terminal it may be necessary to AC couple from the buffer to the following circuitry.

In an alternative implementation of the output buffer the signal may be returned to a ground-centered voltage by integrating the DC voltage at the output of the inductor at a sub-audio rate and subtracting it from the signal in a differential amplifier. Both embodiments are shown.

FIG. 9: The nonlinear inductor. The application of a constant-current bias to the inductor assures that it will produce the desired odd-even monotonic harmonic series as it approaches magnetic saturation. If the inductor is not biased then only odd harmonics are produced, which is not desirable. The constant-current source is shown in FIG. 6. An input buffer is as shown in FIG. 7. An output buffer is as shown in FIG. 8. Operation of the inductor is as follows: an alternating current flows through the inductor due to the application of an alternating voltage at 9.a from the buffer amplifier. The current flow is from the buffer amplifier via coupling capacitor 9.b through the inductor and through the load resistor 9.c. The resulting voltage across load resistor 9.c is taken as the output signal via the output buffer.

Current flow in an inductor produces a magnetizing force in the winding, which in turn produces a concentrated magnetic flux in the core. The total current is composed of the AC audio signal plus the DC constant-current. This causes more magnetic flux in the core when the AC signal is in the same direction as the DC bias, and less flux in the core when the AC signal is in opposition to the DC bias. Assuming the magnitudes of the currents are appropriately scaled, the core of the inductor will approach saturation more quickly for one polarity of the AC signal than for the other polarity. As the core of an inductor approaches saturation, the value of the inductance falls. Since the impedance of an inductor is directly proportional to the inductance, the series impedance of the signal path will vary asymmetrically through the signal cycle. The resulting asymmetry accomplishes the desired spectral alteration. The degree of asymmetry is directly proportional to the constant-current bias and may therefore be adjusted by changing the bias current. The rate of onset of the asymmetry is governed by the magnetic properties of the core, and by the range of AC signal amplitude. A core with a gradual magnetic saturation characteristic will provide a gradual increase in harmonic production. Such a core may be fabricated from powdered iron or Molypermalloy material. A core with an abrupt saturation characteristic will provide a more abrupt onset of harmonic production. Such a core may be fabricated from ferrite or amorphous metal.

The required inductance can be determined by considering the load resistance, R (item 9.c in FIG. 9). The impedance magnitude of an inductor varies directly with frequency. The result of this is that there will be a low-pass filter effect on the signal, i.e. the higher frequencies will be progressively attenuated. A criterion must be arbitrarily chosen for the allowable attenuation at the highest frequency of interest. In an audio application the attenuation should probably not exceed 1 dB ant 15 kHz. Given this requirement, the reactance of the inductor should be about 0.12 times the value of R. For example, if R=1000 Ohms, the inductive reactance, should be about 120 Ohms at 15 kHz. Since X_L=2πFL where:

X_L=Inductive reactance in Ohms

F=frequency in Hz

L=inductance in Henries (H)

the required inductance will be about 1.3 mH. If the inductance index A_L (in nH/n²) of the intended core is known, the number of turns (n) in the winding can be calculated as n=sqrt(L/A_L) remembering that for this equation L is expressed in nH.The required bias current can be determined by the application of the relationship H=(nI)/(0.8Le) where:

H=magnetizing force in Oersteds

n=number of turns of wire in the winding

Le=effective magnetic path length of the core in cm

I=DC bias current in Amperes

and by the relationship B=uH where:

B=magnetic flux density in Gauss

u=average magnetic permeability of the core

Likewise, the required AC audio signal current can be determined by assuming that its peak value should be about 10 to 20 times the bias current. In the derivation of the inductance value above, the reactance at most audio frequencies can be neglected as the current will be mostly determined by the load resistance, R (item 9.c). The signal voltage, which will be required, is simply the product of the required RMS AC current and the load resistance. The RMS AC current can be safely taken to be 0.71× the peak AC current.

All of the above leads to an iterative calculation to determine the core size. Since the inductive reactance is small compared to the load resistance, there will not be much voltage developed across the winding. Since one expression for AC flux density is: B=(Vrms×10E8)/(4.44 nFA_E) where:

Vrms=applied AC voltage across the winding in Volts

n=number of turns

F=frequency of the applied AC voltage in Hz

A_E=effective magnetic cross-sectional area of the core in square cm

it would appear that the cross-section of the core is important. In fact, the applied voltage across the winding is due to the AC current times X_L, and will be small. On the other hand, since B=uH as above, in this case H is due to ΔI and ΔI=the RMS value of the peak AC signal current derived above (Ipkac). H=(nIpkac)/(0.8Le). The total magnetizing force will be the sum of H due to the DC bias current and H due to the AC signal current. Thus the effective magnetic path length of the core dominates. The resulting total flux density, B, should approach the rated saturation flux density for the core material at the highest AC signal level, which is to be processed. In a preferred embodiment, the physical implementation of the inductor should employ a toroidal core in the case of Molypermalloy, powdered iron or amorphous metal, or a pot core in the case of ferrite. This construction will give the best immunity to external magnetic fields, which could otherwise induce extraneous noise.

FIG. 12 shows a circuit, which can be added to the signal path after the spectral modification circuit, described above to counteract an undesired property of either the diode string or the inductor implementation of the nonlinear element. The desired asymmetry is imparted to the audio signal by effectively slightly “squashing” or “stretching” one polarity of the signal relative to the other. The net effect is a slight loss of energy at high signal levels compared to an unprocessed signal. Although the action is electrically instantaneous in the time domain, it is perceived in listening as an average loss of dynamics in loud passages. To counteract this effect, the added item in FIG. 10 is a signal expander. In an expander, the gain is proportional to the signal, i.e. the louder it gets, the louder it gets. In the instant invention, the expansion ratio is quite small being on the same order as the compression due to the nonlinear processes described above. This expander circuit responds to the average amplitude of the signal and operates with electrical symmetry. The result is that the average dynamic compression due to the nonlinear processes is compensated, but the asymmetry is not removed. Therefore the harmonic spectrum shaping is preserved and the dynamic energy is restored.

It should be noted that this technique can also be used to compensate the dynamic compression, which occurs in some loudspeakers due to heating of the voice-coil. In this application the circuit could be used separately or combined with spectral modification circuits of FIG. 9.

In a preferred embodiment the variable gain element, 10.a, is current-controllable and consists of a co-packaged light source and light dependent resistor (LDR). The LDR resistance varies inversely to the illumination from the light source which is typically a light emitting diode (LED) but which can also be an incandescent or electroluminescent device. In the case of the LED, the resistance value of the LDR will be inversely proportional to the current through the LED. The signal detector, 10.b can detect either the average or the root-mean-square value of the input signal. Average detection is done with a precision rectifier circuit well known in the art, the output of which is averaged in a resistor-capacitor network with a time constant appropriate to the desired speed of operation. If the detector has low output impedance and a circuit with high input impedance buffers the voltage on the capacitor, then the attack and release times of the circuit will be symmetrical. Typical attack and release times are on the order 50 milliseconds. This is a sufficient arrangement for most applications. RMS (root-mean-square) detection can also be used but has been found to be subjectively less effective than average detection. Peak detection is also possible as a variation of the precision rectifier circuit using well-known circuit design techniques. It can be argued that peak detection may be more appropriate since it is the signal peaks, which need to be “uncompressed”. Whatever detection method is used, the result must be post-filtered, 10.c to achieve the desired slow time constants. The post filtered voltage from the detector circuit is buffered and scaled as required, 10.d to control the variable gain element, 10.a Where the variable gain element is current-controlled, the voltage from the detector may converted to a current, 10.e using well known techniques.

Claims

1. An electronic circuit for processing an audio signal for introducing predictable and controllable harmonic distortion that increases with increased signal amplitude, said electronic circuit comprising an input buffer, an output buffer, a constant current source and a non-linear element.

2. The electronic circuit of claim 1 wherein said non-linear element comprises semiconductors.

3. The electronic circuit of claim 1 wherein said non-linear element comprises a DC biased inductor.

4. The electronic circuit of claim 1 wherein said audio signal is AC-coupled at both ends of the non-linear element and is forward-biased by said constant-current source.

5. The electronic circuit of claim 1 wherein said constant current source comprises a ring source.

6. The electronic circuit of claim 1 wherein said constant current source comprises a Widlar current mirror.

7. The electronic circuit of claim 1 wherein the quantity of harmonic distortion generated by said circuit is adjustable by varying the bias current from said constant current source.

8. The electronic circuit of claim 3 further comprising an input buffer AC-coupled to the input of said inductor

9. The electronic circuit of claim 8 wherein said input buffer is AC-coupled to the input of said inductor with a coupling capacitor of sufficient value to substantially prevent restriction of low frequencies due to the input impedance of the inductor.

10. The electronic circuit of claim 3 further comprising an output buffer.

11. The electronic circuit of claim 10 wherein said output buffer comprises a MOSFET source-follower DC-coupled to the output of said inductor.

12. The electronic circuit of claim 1 wherein said non-linear element comprises an inductor.

13. The electronic circuit of claim 12 wherein said inductor is provided with a constant-current bias.

14. The electronic circuit of claim 12 wherein higher frequencies passing through said circuit are progressively attenuated, said attenuation not to exceed approximately 1 dB at 15 KHz.

15. The electronic circuit of claim 1 wherein a signal expander is added to said circuit

16. A method for dynamically adjusting the spectral content of an audio signal, which increases the harmonic content through the systematic introduction of amplitude asymmetry.

17. The method of claim 16 in which the amplitude asymmetry creates both even and odd order harmonics.

18. The method of claim 16 in which the asymmetry is controlled so that the resulting harmonic spectrum is low-order and monotonic.

19. An apparatus for dynamically adjusting the spectral content of an audio signal comprising a constant current source, an output buffer amplifier and a biased inductor to produce a controlled asymmetry of the transfer characteristic.

20. The apparatus as set forth in claim 19 wherein said electronic circuit further comprises an input buffer amplifier.

21. The apparatus as set forth in claim 19 wherein the constant current source is adjustable.

22. The apparatus of claim 19 wherein the output buffer amplifier is offset to eliminate DC offset of the biased inductor.

23. An apparatus for dynamically adjusting the spectral content of an audio signal comprising a constant current source, an input buffer amplifier and a biased inductor to produce a controlled asymmetry of the transfer characteristic.

24. The apparatus as set forth in claim 23 wherein said constant current source is adjustable.

25. The apparatus set forth in claim 23 incorporated within the signal path of a power amplifier

26. The apparatus as set forth in claim 23 incorporated within the signal path of a power amplifier.

27. The apparatus as set forth in claim 26 wherein said power amplifier comprises a linear amplifier.

28. The apparatus as set forth in claim 26 wherein said amplifier comprises a switching, or Class D amplifier.

29. The apparatus as set forth in claim 26 where in said amplifier comprises a tracking, or Class H amplifier.

30. An apparatus for dynamically adjusting the spectral content of an audio signal comprising a constant current source and a biased inductor to produce a controlled asymmetry of the transfer characteristic.

31. The apparatus of claim 30 wherein said constant current source is adjustable.

32. The apparatus of claim 30 further comprising an input buffer amplifier.

33. The apparatus of claim 30 further comprising an output buffer amplifier offset to eliminate the DC offset of said biased inductor.

34. The apparatus of claim 30 further comprising an output buffer amplifier.

35. The apparatus of claim 30 incorporated within the signal path of the power amplifier.

36. The apparatus of claim 35 wherein said power amplifier is a linear amplifier.

37. The apparatus of claim 35 wherein said power amplifier is a switching or class D amplifier.

38. The apparatus of claim 35 wherein said power amplifier is a tracking or class H amplifier.

39. An apparatus for adjusting the average amplitude of an audio signal by expansion comprising a variable gain amplifier or a variable attenuator, a signal detector and a control conditioning circuit.

40. The apparatus as set forth in claim 39 wherein variable gain amplifier is characterized as having its gain controlled by a voltage or current control signal.

41. The apparatus as set forth in claim 39 wherein said variable attenuator is controlled by a voltage or current control signal.

42. The apparatus as set forth in claim 39 wherein said signal detector is responsive to the average or peak value of the input signal.

43. The apparatus as set forth in claim 39 wherein said signal detector is responsive to the RMS value of the input signal.

44. The apparatus as set forth in claim 39 characterized as having an expansion ratio that is numerically low.

45. The apparatus as set forth in claim 39 wherein said conditioning circuit is characterized as having adjustable parameters with respect to integration time and expansion ratio.

46. The apparatus as set forth in claim 39 incorporated within the signal path of a spectral content processor.

47. The system as set forth in claim 39 incorporated within the signal path of a power amplifier.

48. The system as set forth in claim 47 wherein said power amplifier is a linear amplifier.

49. The system as set forth in claim 47 wherein said power amplifier is a switching, or Class D amplifier.

50. The system as set forth in claim 47 wherein said power amplifier is a tracking, or Class H amplifier.

51. The system as set forth in claim 39 incorporated as an integral part of a system which comprises a spectral content processor and n audio power amplifier.