Patent Grant
6940233
1. Field of the Invention
The invention relates to driving a CCFL (cold cathode fluorescent lamp) with a high voltage sine wave to produce an efficient and cost effective light source. This light source can be used for backlighting in applications including, but not limited to, a notebook computer, a flat panel display, and a personal digital assistant (PDA).
2. Discussion of the Related Art
Fluorescent lamps are being used in an increasing number of applications. These applications include backlighting for many consumer products including, for example, notebook computers, flat panel displays, and personal digital assistants (PDAs). One common type of fluorescent lamp is a cold cathode fluorescent lamp (CCFL). A CCFL tube contains a gas, which is ionized to generate the desired light for the application.
During standard operation, CCFL tubes typically require a sine wave of 600 V and run at a current of several milliamps. However, the starting (or striking) voltage of the CCFL tube, which is used to ionize its contained gas, can be as high as 2000 V. At start up, the CCFL tube looks like an open circuit, i.e. the impedance of the CCFL tube prevents any current. However, after the gas is ionized, the impedance drops and current starts to flow in the CCFL tube.
In typical embodiments, the CCFL tube is driven by a high Q circuit, wherein Q refers to the quality of the circuit and is measured by the inductive or capacitive reactance of the circuit at resonance divided by the resistance. This high Q circuit generally includes additional capacitors and inductors, which undesirably increase the number of components in the system. Therefore, a need arises for a CCFL circuit that minimizes the number of additional components while still achieving an efficiency of at least 85%.
In accordance with one feature of the invention, a CCFL circuit can include a PMOS transistor, first and second NMOS transistors, and a high turns ratio transformer. The transformer can include a primary coil having a center tap, thereby forming first and second primary windings, as well as a single secondary coil. The drain of the PMOS transistor can be connected to the center tap and the source of the PMOS transistor can be connected to a battery. The drains of the first and second NMOS transistors can be connected to the ends of the first and second primary windings, respectively. The sources of the first and second NMOS transistors can be connected to a voltage source VSS,
Of importance, the first primary winding is tightly coupled to the second primary winding. However, the first and second primary windings are loosely coupled to the secondary coil, thereby resulting in significant leakage inductance. Specifically, this loose coupling results in significant leakage inductance, which can be modeled as a series inductance in the secondary coil. In one embodiment, the primary to secondary turns ratio is approximately 100 and the primary inductance is approximately 200 uH.
Due to the leakage inductances of the transformer, voltages at the drains of the first and second NMOS transistors can potentially ring to values substantially higher than the ideal value (e.g. twice times the battery voltage). To limit the extent of the ringing voltage, the CCFL system can include a snubbing circuit connected to the drains of the NMOS transistors, the source of the PMOS transistor, and the first and second primary windings.
The snubbing circuit can include first and second diodes, a capacitor, and a resistor. In one embodiment, an input terminal of the first diode can be connected to an end of the first primary winding, an input terminal of the second diode can be connected to the end of the second primary winding, and output terminals of the first and second diodes can be connected to a common node. A resistor and a capacitor can be connected in parallel between the common node and the battery.
In the snubbing circuit, the capacitor, resistor, and diodes are configured to maintain a nominal voltage at the common node. In one embodiment, this nominal voltage is approximately twice the battery voltage. However, if either of the drains of the first and second NMOS transistors have a voltage above that nominal voltage, then the first and second diodes forward bias and allow the ringing energy to charge the capacitor. The resistor can bleed off the extra ringing energy, thereby preventing the voltage at the common node from increasing substantially higher than the nominal voltage.
In accordance with another feature of the invention, a detection circuit for detecting over-voltages in a CCFL circuit is provided. Of importance, the resistive and capacitive components of the detection circuit are isolated from the high voltage terminal of CCFL tube. Having resistive and capacitive components exposed to such a high voltage can undesirably reduce current and energy through such components, thereby reducing efficiency.
The detection circuit can include an integrator receiving an output signal of the CCFL circuit. The integrator generates a DC signal COMP such that a time-averaged voltage of the output signal from the CCFL circuit is substantially equal to a reference voltage. Advantageously, the COMP signal does not experience high voltages and typically does not vary significantly during normal circuit operation. For example, even during dimming cycles, the rise and fall of the COMP signal are smooth and relatively noise free. However, if arcing occurs, then the COMP signal becomes erratic as the circuit fights to stay in regulation.
The detector circuit can further include a first capacitor having a first terminal connected to an output of the integrator, a first diode having an input terminal connected to a second terminal of the first capacitor, and a second diode having an output terminal connected to the second terminal of the first capacitor. The detector circuit can further include a pnp transistor having a base connected to an output terminal of the first diode, an emitter connected to an input terminal of the second diode, and a collector connected to a voltage source VSS. A first resistor can be connected between the output terminal of the first diode and the voltage source VSS. A second capacitor can be connected between the output terminal of the first diode and the voltage source VSS. A second resistor can be connected between the source of the NPN transistor and a voltage source VDD. In this configuration, the emitter of the pnp transistor can provide a signal indicating whether an over-voltage condition occurs in the CCFL circuit. In one embodiment, the second capacitor and the second resistor establish a time constant for a trigger transition period of the output signal of the CCFL circuit.
In accordance with another feature of the invention, a method of detecting an over-voltage condition in a CCFL circuit is provided. The method can include providing a transistor configured for generating a detect signal indicative of the over-voltage condition. The transistor can be isolated from the CCFL circuit using an integrator. A first circuit can be provided for pumping up a voltage at a base of the pnp transistor. A second circuit can be provided for leaking the voltage at the base of the pnp transistor. If an output signal of the integrator is moving erratically, then the pumping can overcome the leaking, thereby increasing a voltage at a drive terminal of the transistor as well as the detect signal. In one embodiment, the method can further include establishing a time constant for a trigger transition period of the output signal of the CCFL circuit.
In accordance with another feature of the invention, another detection circuit for detecting an over-voltage condition in a CCFL circuit is provided. The detection circuit can include a PCB trace formed within 7 to 15 mils (thousandths of an inch) of a high voltage connector of the CCFL circuit. The PCB trace provides a detect signal that indicates whether the over-voltage condition exists.
In accordance with another feature of the invention, a CCFL system for driving first and second CCFL tubes is provided. The CCFL system can include a PMOS transistor, first and second NMOS transistors, and a high turns ratio transformer. The transformer includes a primary coil having a center tap forming a first primary winding and a second primary winding, and a secondary coil having a first secondary winding and a second secondary winding. In one embodiment, the drain of the PMOS transistor is connected to the center tap and the source of the PMOS transistor is connected to a battery. The drain of the first NMOS transistor is connected to an end of the first primary winding, the drain of the second NMOS transistor is connected to an end of the second primary winding, and the sources of the first and second NMOS transistors are connected to a voltage source VSS.
Of importance, the first primary winding is tightly coupled to the second primary winding, and the first and second primary windings are loosely coupled to the secondary coil, thereby resulting in significant leakage inductance. The first CCFL tube can be coupled between the first secondary winding and the voltage source VSS, whereas the second CCFL tube can be coupled between the second secondary winding and the voltage source VSS.
Advantageously, because the current through the first and second CCFL tubes is substantially equal (as long as the parasitic capacitive paths are approximately equal for both tubes), only one feedback loop connected to the first CCFL tube is necessary for determining the current through either CCFL tube.
In one embodiment, the CCFL system further includes at least a first resistor connected between the first CCFL tube and the voltage source VSS and a second resistor connected between the second CCFL tube and the voltage source VSS. The first resistor and the second resistor are sized to provide substantially equal resistances, thereby ensuring that impedances of the first and second CCFL tubes are substantially equal.
In another embodiment where the application is used to drive two CCFLs with one transformer, the secondary coil of the CCFL system includes a connection located between the first and second secondary windings. The connection is placed at approximately halfway between the first and second secondary windings. The connection provides a voltage substantially at the voltage source VSS. In contrast, the ends of the first and second secondary windings provide a large positive voltage and a large negative voltage, respectively. This connection provides a convenient method for detecting over-voltages since during normal operation it remains near VSS. If one of the CCFLs becomes open (or somewhat open) then the voltages in the secondary windings are no longer balanced and the midpoint of the two secondary windings will differ significantly from ground. This condition can be easily detected with a resistive voltage divider and a comparator. It dissipates little power since the midpoint of the two secondary windings is normally near VSS.
A CCFL system for driving first, second, third, and fourth CCFL tubes is provided. The CCFL system includes a PMOS transistor as well as first and second NMOS transistors. The CCFL system also includes a first high turns ratio transformer, which can have a first primary coil with a first center tap, thereby forming a first primary winding and a second primary winding. The first high turns transformer can also have a first secondary coil, which includes a first secondary winding and a second secondary winding. The CCFL system can further include a second high turns ratio transformer, which can have a second primary coil with a second center tap, thereby forming a third primary winding and a fourth primary winding. The second high turns ratio transformer can have a second secondary coil, which includes a third secondary winding and a fourth secondary winding.
The drain of the PMOS transistor is connected to the first and second center taps and the source of the PMOS transistor is connected to a battery. The drain of the first NMOS transistor is connected to an end of the first primary winding and an end of the third primary winding. The drain of the second NMOS transistor is connected to an end of the second primary winding and an end of the fourth primary winding. The sources of the first and second NMOS transistors are connected to a voltage source VSS.
The first primary winding is tightly coupled to the second primary winding. The third primary winding is tightly coupled to the fourth primary winding. The first and second primary windings are loosely coupled to the first secondary coil. The third and fourth primary windings are loosely coupled to the second secondary coil. The first CCFL tube is coupled between the first secondary winding and the voltage source VSS. The second CCFL tube is coupled between the second secondary winding and the voltage source VSS. The third CCFL tube is coupled between the third secondary winding and the voltage source VSS. The fourth CCFL tube is coupled between the fourth secondary winding and the voltage source VSS. The first and fourth secondary windings are connected. The second and third secondary winding are connected.
In one embodiment, the CCFL system can further include a current sensing network coupled to one of the first, second, third, and fourth CCFL tubes. In another embodiment, the CCFL system can further including a fault circuit coupled to the second secondary winding and the third secondary winding. The fault circuit can include a first resistor divider, a second resistor divider, a first diode coupled to the first resistor divider, and a second diode coupled to the second resistor divider. The first and second diodes can be connected to provide a logic OR function to fault detection circuitry.
A method of determining a fault condition for a system is provided. The system can include a transformer having a primary coil and a secondary coil, a first CCFL tube, and a second CCFL tube. The method can include creating a tap in the secondary coil, thereby forming a first secondary winding and a second secondary winding. The first CCFL tube can be connected to an end of the first secondary winding. The second CCFL tube can be connected to an end of the second secondary winding. The fault condition can be determined by sensing the voltage at the tap.
In one embodiment, determining the voltage at the tap includes dividing and rectifying the voltage. Dividing the voltage can include sizing a resistor divider so that under normal operating conditions, the rectified voltage is less than a first predetermined threshold voltage, and during a fault condition, the rectified voltage is higher than a second predetermined threshold voltage.
Another CCFL system for driving first, second, third, and fourth CCFL tubes is provided. This CCFL system also includes a PMOS transistor as well as first and second NMOS transistors. The CCFL system further includes a single high turns ratio transformer. The transformer includes a primary coil having a center tap forming a first primary winding and a second primary winding. The transformer further includes a secondary coil having a first secondary winding, a second secondary winding, a third secondary winding, and a fourth secondary winding.
The drain of the PMOS transistor is connected to the center tap and the source of the PMOS transistor is connected to a battery. The drain of the first NMOS transistor is connected to an end of the first primary winding, a drain of the second NMOS transistor is connected to an end of the second primary winding, and sources of the first and second NMOS transistors are connected to a voltage source VSS. The first primary winding is tightly coupled to the second primary winding, and the first and second primary windings are loosely coupled to the first, second, third, and fourth secondary coils.
The first CCFL tube is coupled between one end of the first secondary winding and the voltage source VSS. The second CCFL tube is coupled between one end of the second secondary winding and the voltage source VSS. The third CCFL tube is coupled between one end of the third secondary winding and the voltage source VSS. The fourth CCFL tube is coupled between one end of the fourth secondary winding and the voltage source VSS. Note that the other ends of the first and second secondary windings are connected. Similarly, the other ends of the third and fourth secondary winding are connected. As was the case with two separate transformers, the connections of the secondary windings to each other provide a convenient method for detecting over-voltage faults. In one embodiment, a current sensing network can be coupled to one of the first, second, third, and fourth CCFL tubes.
Yet another CCFL system for driving first, second, third, and fourth CCFL tubes is provided. This CCFL system also includes a PMOS transistor as well as first and second NMOS transistors. The CCFL system further includes a single high turns ratio transformer. The transformer includes a primary coil having a first center tap, thereby forming a first primary winding and a second primary winding. The transformer also includes a second center tap, thereby forming a third primary winding and a fourth primary winding. The transformer also includes a secondary coil having a first secondary winding, a second secondary winding, a third secondary winding, and a fourth secondary winding.
The drain of the PMOS transistor is connected to the first and second center taps and the source of the PMOS transistor is connected to a battery. The drain of the first NMOS transistor is connected to an end of the first primary winding and an end of the third primary winding. The drain of the second NMOS transistor is connected to an end of the second primary winding and an end of the fourth primary winding. The sources of the first and second NMOS transistors are connected to a voltage source VSS. The first primary winding is tightly coupled to the second primary winding, the third primary winding is tightly coupled to the fourth primary winding, the first and second primary windings are loosely coupled to the first and second secondary coils, and the third and fourth primary windings are loosely coupled to the third and fourth secondary windings.
In this CCFL system, the first CCFL tube is coupled between one end of the first secondary winding and the voltage source VSS, the second CCFL tube is coupled between one end of the second secondary winding and the voltage source VSS, the third CCFL tube is coupled between one end of the third secondary winding and the voltage source VSS, and the fourth CCFL tube is coupled between one end of the fourth secondary winding and the voltage source VSS. The other ends of the first and second secondary windings are connected. Similarly, the other ends of the third and fourth secondary winding are connected. As was the case earlier, the ends of the secondary windings that are connected together provide convenient means to detect over-voltage faults. The method of determining over-voltage faults in the single transformer case (4 tubes) is substantially analogous to the fault determining method of the 2 transformer case (also 4 tubes). In one embodiment, a current sensing network can be coupled to one of the first, second, third, and fourth CCFL tubes.
A method of implementing a transformer is also provided. The transformer has a middle area, a first end, and a second end. The method includes providing a low AC voltage in the middle area, a first high AC voltage having a first phase at the first end, and providing a second high AC voltage having a second phase at the second end. In one embodiment, the low AC voltage is VSS. In another embodiment, the first phase is positive and the second phase is negative. The first end can include a first winding and a second winding providing first in-phase outputs, whereas the second end can include a third winding and a fourth winding providing second in-phase outputs. Of importance, the phase of the first in-phase outputs is out of phase with the second in-phase outputs.
In accordance with one feature of the invention, the high voltage required for CCFL operation can be generated using a transformer-LC tank circuit combination driven by several small power mosfets. For example,
When PMOS transistor 101 is turned off, the voltage of midpoint 109 returns to ground as does the drain of NMOS transistor 103 that was at twice the battery voltage. Halfway through one cycle, NMOS transistor 102 (that was on) turns off and NMOS transistor 103 (that was off) turns on. At this point, PMOS transistor 101 turns on again, thereby allowing current to ramp up in side 108 of the primary winding. Energy in the primary winding is transferred to the secondary winding and stored again in the leakage inductance Lleak, but this time with the opposite polarity.
Thus, the duty cycle of PMOS transistor 101 controls the amount of power transferred from the primary winding to the secondary winding in transformer 104. Note that CCFL circuit 100 can work with PMOS transistor 101 on constantly (i.e. a duty cycle of 100%), although the power would be unregulated in this case.
The efficiency of CCFL circuit 100 is still high even with the addition of the second MOS transistor (i.e. either NMOS transistor 102 or NMOS transistor 103) in the current path. The I-squared losses of the extra MOS transistor are especially negligible. For instance, consider a 6W application running at 10V battery voltage. The power (P) loss for a transistor with 50 mohm resistance (R) and a drain current (I) of 600 mA is:
P=I×I×R=600×600×0.05=18 mW
The switching losses of NMOS transistors 102 and 103 must also be taken into account to determine the efficiency of CCFL circuit 100. However, these switching losses are hardly more significant than the I-squared losses. For example, the power loss for a transistor with a change in drain voltage (V) of 10V, a rise time (tau) of the gate drive signal of 50 nS, and a period (T) of 10 μs is:
P=⅓×I×V×(tau/T)=⅓×600×10×(50/10)=10 mW
Note that because there is no primary side capacitor, no capacitor ESR losses are incurred. Thus, when considering both I-squared and switching losses in the NMOS transistors, CCFL circuit 100 could easily achieve an efficiency of approximately 85%. However, the losses associated with transformer 104 can be significantly more than the I-squared and switching losses. Therefore, the transformer losses, discussed in more detail in reference to
Traces 401, 402, and 403 in each figure show the gate drive waveforms for transistors 101, 102, and 103, respectively. In one embodiment, the gate drive waveform for transistor 101 drives up to the battery voltage but down only to approximately 7.5 V below the battery voltage. Note that in the preferred embodiment trace 401 would be driving a PMOS transistor so that the PMOS device would be “on” when trace 404 is low and “off” when trace 404 is high. The NMOS case is exactly the opposite of the PMOS case such that when trace 402 is high then its NMOS transistor is “on” and when trace 402 is low then its transistor is “off”. Trace 404 (in
Trace 405 shows the voltage at the drain of NMOS transistor 102 (i.e. the voltage on the line connected to the primary winding of transformer 104) (note that the trace for NMOS transistor 103 would be identical, but shifted in time). Trace 407 shows the current through the NMOS transistor, which is equal to the current in PMOS transistor 101 for the portion of time that PMOS transistor 101 is conducting (see region I, for example). As the current ramps up in the primary winding, energy is transferred to the secondary winding and stored in the leakage inductance Lleak (and any parasitic capacitance on the secondary winding). Note that the current in the NMOS transistor is close to zero when that NMOS transistor is turned off, thereby indicating that CCFL circuit 100 is being driven close to its resonant frequency. Although this embodiment does not sense the zero current point directly, the switching frequency could be modified such that the zero current condition was met.
Once PMOS transistor 101 completes one on/off cycle, it is repeated again with the alternate NMOS transistor conducting. This complementary operation produces a symmetric, approximately sinusoidal waveform at the input to the load (e.g. the CCFL tube 105), as shown by trace 408.
The operation of CCFL circuit 100 can be divided into 4 regions (I, II, III, and IV) as shown in
As the duty cycle changes with battery voltage, the overall resonant frequency might also change. For example, referring to trace 407 on
System Overview
In accordance with the present invention, the current through CCFL 801 is controlled by a combination of the duty cycle of the driving waveform (i.e. the waveform that drives a transistor 803) and the frequency of that driving waveform. In one embodiment, system 800 includes a first control block connected to a node N3 that provides a DC signal COMP to a positive terminal of a comparator 853. The first control block controls the duty cycle of the driving waveform. Specifically, the first control block senses the CCFL current, integrates it against an internal reference and adjusts the duty cycle to obtain the desired power.
System 800 further includes a second control block that provides a signal RAMP (sawtooth waveform) to a negative terminal of comparator 853. The output signal of comparator 853, i.e. a PWM signal (a pulse width modulated waveform), is provided to an output driver 880, which in turn provides the clock signals OUTA, OUTAB, and OUTC to transistors 803, 804, and 816, respectively (i.e. the driving waveforms to CCFL circuit 801). The second control block can be used to change the frequency of the driving waveforms as a function of battery voltage. As the voltage of battery 802 increases, the oscillator frequency also increases. This tends to keep the circuit operating near its resonant frequency as the battery voltage changes.
System 800 further includes a third control block that adjusts the brightness of CCFL tube 805 by turning the lamp on/off at varying duty cycles. In this embodiment, a user-provided BRIGHT voltage can be compared with a slow ramp signal to generate a CHOP signal. This CHOP signal is provided to fault and control logic 870, which in turn generates a NORM signal input to output driver 880.
First Control Block
As described above, the current through CCFL 805 can be sensed on a line 813, which is coupled to node N3. In accordance with one feature of the present invention, that voltage on line 813 can drive an input of an integrator 820. Specifically, integrator 820 receives the voltage on line 813 through a resistor 821, wherein resistor 821 is coupled to the negative terminal of an error amplifier 823. In one embodiment, resistor 821 provides a resistance of 10 kOhm. Error amplifier 823 compares this voltage with a reference voltage VR1 received on its non-inverting terminal.
In one embodiment, reference voltage VR1 is derived from a temperature and supply stable reference (such as a bandgap reference) through a resistor divider. Other known techniques for providing reference voltage VR1 can also be used. In one embodiment, reference voltage VR1 can be between 0.5 V and 3.0 V. Note that the larger the reference voltage VR1, the larger the average voltage across resistor 821. In contrast, if reference voltage VR1 is too small, then error amplifier offsets and other non-idealities may become significant. Therefore, in one embodiment, reference voltage VR1 can be 2.5 V.
A capacitor 822, in one embodiment providing a capacitance of 1 uF, is coupled to the negative terminal and the output terminal of error amplifier 823, thereby completing the formation of integrator 820. The purpose of integrator 820 is to generate a DC signal COMP such that the time-averaged voltage at node N4 is substantially equal to reference voltage VR1.
Clamping circuit 840 can limit the increase of the COMP signal. In one embodiment, clamping circuit 840 includes an error amplifier 842 providing an output signal to the gate of a transistor 841. Transistor 841, an n-type transistor, has its source coupled to VSS and its drain coupled to the positive input terminal of error amplifier 842 as well as to the output of integrator 820. Error amplifier 842 further includes a negative input terminal coupled to a current source 843 and one terminal of a capacitor 844 (the other terminal being coupled to VSS). In this configuration, clamping circuit 840 allows the COMP signal to increase at a rate that is no faster than current source 843 can charge capacitor 844. Thus, clamping circuit 840 prevents the COMP signal (and thus the PWM signal) from immediately going to its full power mode, thereby allowing CCFL 805 to start up slowly. Having a gradual increase of the power to CCFL 805 advantageously prolongs its life as well as the life of other components of CCFL circuit 801.
Second Control Block
The oscillator frequency of VCO 850 determines the frequency of the drive signal at the gate of PMOS transistor 803. In this embodiment, the user can set the minimum oscillator frequency with resistor 852, wherein
Oscillator Frequency (Hz)=2.8E9/Resistance 852 (ohms)
A detail of VCO 850 is shown in FIG. 10. In this embodiment, VCO 850 includes a user-adjustable current source including an error amplifier 1001, resistor 852, and NMOS transistor 1002. Error amplifier 1001 is configured to receive a reference voltage VR3 and the signal at the source of NMOS transistor 1002. Error amplifier 1001 provides its output signal to the gate of NMOS transistor 1002. In this configuration, the current is equal to the reference voltage VR3 divided by the resistance of resistor 852. In one embodiment, reference voltage VR3 is approximately 1.5 V.
This current is then mirrored using PMOS transistors 1003 and 1004 onto a capacitor 1005. That current charges capacitor 1005, thereby increasing the voltage at node N11. Specifically, the voltage ramps up to a predetermined voltage determined by an error amplifier 1007, which receives the ramp voltage on node N11 and a reference voltage VR4. In one embodiment, the reference voltage VR4 can be approximately 3.0 V, thereby also setting the predetermined ramp voltage on node N11 to 3.0 V. When the voltage on node N4 reaches the predetermined voltage, error amplifier 1007 outputs a signal to close a switch 1006, thereby discharging capacitor 1005 to VSS (e.g. ground). Therefore, in this configuration, capacitor 1005, error amplifier 1007, and switch 1006 form a standard relaxation oscillator. Note that the output of error amplifier 1007 is also buffered using inverters 1009 and 1010 to provide the clock signal CLK. Further note that the ramping signal generated at node N11, i.e. signal RAMP, can be used to create the PWM signal (see comparator 853 in FIG. 8A).
In one embodiment, a current divider 1008, a PMOS transistor 1011, and error amplifier 873 can be used to add some current to node N11, thereby increasing the frequency of the RAMP signal. In this embodiment, error amplifier 873 is connected in unity gain, which will output a constant voltage substantially equal to reference voltage VR2. In one embodiment, reference voltage VR2 is approximately 1.25 V.
As the voltage Vbatt increases, more current flows across resistor 851 into current divider 1008. Resistor 851, which is coupled to battery 802, controls how much the oscillator frequency increases as a function of battery voltage (Vbatt). In one embodiment, resistor 851 has a resistance of 200 kohm. The relationship is:
ΔFrequency (Hz)=3.44E8*(Vbatt−VR2)/Resistance 851
In one embodiment, current divider 1008 divides the current by a factor of 50, thereby ensuring the amount of current added to that already present on node N11 is quite small. Because the oscillator frequency can be adjusted upwards as the battery voltage increases, harmonic distortion of the output waveform can be advantageously minimized.
Third Control Block
The third control block adjusts the brightness by turning the lamp on and off at varying duty cycles. In this description, “dimming cycle” refers to the complete period including both an “on” and “off” states. At the end of each dimming cycle, the COMP pin is pulled low. At the beginning of a new dimming cycle, the COMP signal tries to increase quickly but it is clamped to the voltage at the SSV (soft-start voltage) pin. Capacitor 844, which is discharged at the end of every dimming cycle, sets the slew rate of the voltage at the SSV pin, and hence also the maximum positive slew rate of the COMP pin.
In one embodiment, a ramp generator 860 can generate a slow ramp voltage (i.e. a sawtooth waveform) that is limited by a small capacitor 861. In one embodiment, capacitor 861 has a capacitance of approximately 0.015 uF. A comparator 862 can compare this ramp voltage with a BRIGHT signal, e.g. a DC voltage provided by a user, which is proportional to the desired brightness. Based on this comparison, comparator 862 outputs a variable duty factor signal CHOP. Of importance, the CHOP signal can stop output driver 880 from switching, thereby stopping the OUTA signal by pulling it high. Signals OUTAPB and OUTC continue switching in order to allow the energy in the LC tank circuit to dissipate slowly without producing large voltages. As the voltage at the BRIGHT pin increases, the duty cycle of the dimming cycle (and the brightness of CCFL tube 805) increases.
The frequency of the dimming cycles is set by the value of capacitor 861 and is proportional to the current set by resistor 852 (which sets the minimum operating frequency of VCO 850). Setting capacitor 861 to 0.01 uF, resistor 852 to 47.5 kOhm, and VSS to ground yields a dimming cycle frequency of approximately 100 Hz. This frequency should vary inversely with the value of capacitor 861.
The brightness may also be controlled by using a variable resistor in place of resistor 807 (and 808). In this case, the BRIGHT pin should be pulled to VDD so that CCFL 811 runs at 100% duty cycle. Note that this configuration can result in some flicker at low intensities, but is otherwise functionally equivalent to the embodiment using resistor 807.
Start-Up Operations
In one embodiment, an SSC signal can be generated by alternative current sources. Specifically, two current sources, one at 1 uA and another at 150 uA, can be selectively connected to the SSC terminal of fault and control logic 870 as well as to one terminal of capacitor 871. Capacitor 871 has its other terminal connected to VSS. In one embodiment, capacitor 871 has a low capacitance of 0.022 uF.
During a “cold” start-up operation of CCFL 805, i.e. a start-up following a predetermined period of time in which CCFL 805 has been off, fault and control logic 870 generates an active signal FIRST, thereby selecting the lower value current source (i.e. 1 uA, in this embodiment). In contrast, during subsequent “warm” starts, i.e. a start-up following a time period less than the predetermined period of time, fault and control logic 870 generates an inactive signal FIRST, thereby selecting the higher value current source (i.e. 150 uA). In this manner, capacitor 871 takes longer to charge during a cold start-up than a warm start-up. The ramp generated by the SSC pin is used to define a time period when the fault detection circuity is disabled. Without this “blanking” interval the circuit would permanently shut down during every dimming cycle because of a misperceived fault. This operation is more fully explained in the fault circuitry description.
Exemplary Layout
Capacitor 828, in this embodiment can serve as a bypass capacitor, which effectively supplies driver section 880 with large peaks of AC current necessary for switching the external mosfets 803, 804, and 816. In one embodiment, capacitor 829 can provide a capacitance of 4.7 uF. A dashed box 825 indicates that the components therein can be fabricated on one chip.
CCFL Circuit Operation
Referring to
Alternative Embodiments
In the embodiment of
The embodiment of
Supply Voltages
In accordance with one embodiment, battery 802 can provide a voltage source between 7-24V (typical for 3 lithium ion cells provided in a notebook computer application). Most of the circuitry in system 800 can operate at a conventional voltage, e.g. 5 V. To this end, PNP transistor 827 can be used to provide a regulated VDD voltage from battery 802. Specifically, the PNP pin (see
When the chip enable signal (CE) is low (e.g. less than 0.4 V), the chip goes into a zero current state. In one embodiment, the PNP pin can be put into a high impedance state, thereby reducing the VDD voltage to zero volts. The VDD voltage can be sensed internally so that the switching circuitry will not turn on unless the VDD voltage is larger than a first predetermined threshold voltage (e.g. 4.5 V) and the internal reference (e.g. 3.3V) is valid. Circuitry within the reference block is used to determine if the reference is close to regulation. Once it has been determined that the reference is close to regulation, the reference voltage can be used to determine if VDD is above a certain threshold voltage, e.g. 4.5V. In one embodiment, once the predetermined threshold has been reached, the switching circuitry will run until the VDD voltage is less than a second predetermined threshold voltage (e.g. 3.5 V).
Output Drivers
In one embodiment, the OUTAPB and OUTC pins are standard CMOS driver outputs. In contrast, in a preferred embodiment, the OUTA driver pulls up to the battery voltage, e.g. a maximum of 24 V, but is internally clamped to within 8 V of the battery voltage. On each signal transition for PMOS transistor 803, the OUTA pad will sink/source current (e.g. approximately 500 mA) for a short period of time (e.g. approximately 100 nS). After the initial burst of current, the current is scaled back (e.g. 1 mA for sinking and 12 mA for sourcing). This technique allows for fast edge transitions, yet minimizes overall power dissipation.
Fault Protection
In accordance with another feature of the present invention, fault condition checks can identify undesirable voltages provided associated with CCFL tube 805. When any one of the fault conditions is met, then CCFL circuit 801 is latched off. At this point, a power on reset or cycling the CE pin can restore CCFL circuit 801 to normal operation.
A first fault condition check identifies an over-voltage provided to CCFL tube 805. In this embodiment of system 800, resistors 811 and 810 are coupled between node N6 and VSS, thereby forming a voltage divider. In this configuration, a node N5 between resistors 811 and 810 provides an OVP signal proportional to the voltage across CCFL 805. Node N5 is connected to fault and control logic 870 via line 812. If the OVP signal (and thus CCFL voltage) is too high, then a long active CHOP signal generated by fault and control logic 870 can actually shut down CCFL circuit 801 to prevent potentially dangerous conditions from developing. In other words, if the voltage at node N6 is too high (e.g. 3 V), then fault and control logic 870 will turn off the chip regardless of the current operating mode.
A second fault condition check identifies an under-voltage provided to CCFL tube 805. Specifically, fault and control logic 870 can also check to see that there are no under-voltages at node N6. The second fault condition check can be used to ensure that the input voltage to CCFL tube 805 is above a predetermined voltage level on a cycle-by-cycle basis. In one embodiment, fault and control logic 870 is semi-disabled for a predetermined period of time after either a cold or warm start-up. Alternatively, this protection is disabled while the SSC ramp is below 3 V (which typically occurs during start-up and at the beginning of every dimming cycle). (Note that the first SSC ramp after power on reset (or CE enabled) can be 150 times slower than subsequent start up ramps.) After start-up, if the OVP pin does not cross a predetermined (e.g. 250 mV) threshold once during a certain number of (e.g. four) successive clock periods, then this fault will be identified. In this manner, fault and control logic 870 prevents an unwanted shutdown down due to a single spurious under-voltage event. After the semi-disabled time, fault and control logic 870 can again be fully enabled.
A third fault condition check can be used to monitor the current through CCFL tube 805. Specifically, to monitor the current, the voltage at node N4 can be checked. In one embodiment, the trigger voltage at node N4 is 250 mV. Fault and control logic 870 receives a CSDET signal from node N4. Thus, fault and control logic 870 can look for under-voltage conditions (tube under-current) at node N4. Once again, this fault check can be disabled for a certain period after each dimming cycle (similar to the under-voltage check of node N6). In one embodiment, fault and control logic 870 must receive four consecutive periods of under-voltage operation at node N4 before fault and control logic 870 generates a fault and shuts the chip down. Alternatively, this protection can be disabled while the SSC ramp is below 3 V.
Note that in one embodiment the resistor divider comprising resistors 810 and 811 (see also, resistors 922 and 923 in
The CLK signal is the clock output from VCO 850. The CLK signal provides the time base for the gate drive of the external FETs (like PMOS transistor 803). The OVP signal, which is generated at node N5 of CCFL circuit 801 (see FIG. 8A), is provided to two comparators, i.e. comparator 1102 for determining an over-voltage and comparator 1103 for determining an under-voltage. The CSDET signal, which is generated at node N4, is provided to a comparator 1104 for monitoring of the CCFL current. As previously mentioned, the under-voltage and the under-current conditions can trigger a fault if these conditions are presented a predetermined number of times. Hence, 2-bit counters can be coupled to the outputs of comparators 1103 and 1104, thereby facilitating the counting of successive under-voltage and under-current conditions.
The SSC signal, which is a capacitor-controlled voltage ramp available in system 800, and a reference voltage (in this case 3.3 V) are provided to a comparator 1105. In this configuration, the BLANK signal output by comparator 1105 is low while the SSC signal is below 3.3V, thereby effectively disabling the two fault checks associated with the 2 bit counters. Thus, the SSC signal can be used to provide a time delay during which two of the fault detection checks are disabled. Note that an output signal FIRST of fault and control logic 870 is high during the first dimming cycle after power is turned on, thereby causing the SSC pin to source significantly less current than on subsequent burst cycles. At the beginning of every dimming cycle, SSC starts at 0V and ramps linearly up to the VDD supply, however the first of these ramps after power up is 150 times slower than subsequent ramps.
Fault and control logic 870 also receives a chip enable CE signal (on line 872 in FIG. 8A), which can generate a power on reset condition as well as turning the CCFL on and off.
Arc Detection Circuitry
Typically, an over-voltage condition results when the impedance of the load increases above a predetermined level. Specifically, if the impedance goes too high, then the current sensed at the CSDET pin will fall below its threshold and circuit 801 will shut down. However, another problem occurs when CCFL tube 805 has poor contact to the rest of the circuit, i.e when a connector of CCFL tube 805 is not plugged in all the way.
In this case, the voltage generated by transformer 814 is so high that it can easily jump a 1 mm gap in air. Unfortunately, CCFL tube 805 will still operate in this condition, arcing across the open connector. If the connector is disconnected from CCFL tube 805 by a substantial distance (1 cm), then it is unlikely arcing will be a problem. If the connector is connected correctly there will also be no problem. However with a small gap in the connector (or anywhere in the high voltage power path) arcing can occur, thereby causing undesirable high temperatures in CCFL circuit 801. Therefore, over-voltage conditions caused by arcing should be detected as quickly as possible, and when detected, the circuit should be shut down.
As described above, an over-voltage condition can be sensed using a voltage (or capacitor) divider, which is coupled to the secondary winding of transformer 814 as well as CCFL tube 805. Unfortunately, this divider can change the AC characteristics of CCFL tube 805 and thus its resonant frequency. Moreover, the divider, by adding components, complicates the PC board layout.
Therefore, in accordance with one embodiment of the invention shown in
Advantageously, circuit 1200 can generate the OVP signal, thereby eliminating the need for resistors 810 and 811 (FIG. 8A). Of importance, the resistive and capacitive components of circuit 1200 are isolated from the high voltage terminal of CCFL tube 805 (i.e. node N6). Having resistive and capacitive components exposed to such a high voltage can undesirably reduce current and energy through such components, thereby reducing efficiency. Moreover, the high voltage at node N6 can affect the impedance, thereby making voltage detection difficult.
In contrast to node N6, the COMP signal does not experience high voltages and typically does not vary significantly during normal circuit operation. For example, even during dimming cycles, the rise and fall of the COMP signal are smooth and relatively noise free. However, if arcing occurs, then the COMP signal becomes erratic as the circuit fights to stay in regulation.
Therefore, detection of this erratic behavior of the COMP signal can be used to shut the circuit down. In
Components of circuit 1200 operate in the following manner. Fast transitions (e.g. on the order of milliseconds) of the COMP signal are received by a capacitor 1202. A positive transition is passed through diode 1207 to the base of pnp transistor 1205. When the voltage at the base of pnp transistor 1205 increases so does the voltage on its emitter (which is coupled to a voltage VDD via a resistor 1208). A negative transition is blocked by diode 1207, but during this transition, diode 1206 conducts current from VDD through resistor 1208 into capacitor 1202. On the next positive transition, capacitor 1202 is charged up and is ready to supply current into the base of pnp transistor 1205. In this embodiment, resistor 1203 and a capacitor 1204 establish the time constant for the “fast” transition period. During a fast transition, the voltage at the emitter of pnp transistor 1205 will eventually increase to a point where it will trip the OVP threshold of the chip, thereby shutting down CFFL circuit 801 (FIG. 8A).
Another method of detecting and shutting down the circuit during arcing events is to use a preferential arcing path. For example, in one embodiment shown in
Different operational characteristics could be achieved by modifying gap 1320 on the PC board and by opening the solder mask on the area between the preferential arcing node 1310 and connector 1301. As the preferential arcing gap 1320 between node 1310 and connector 1301 is made smaller the voltage at which arcing will occur is also smaller because the electric field between the two electrodes of the arcing path increases as the distance between those two electrodes decreases (assuming a constant potential difference between the two electrodes). Note that because gap 1320 to connector 1301 would have air as its dielectric, it is advantageous to use air as the dielectric for the preferential arcing path as well.
Circuitry For Minimizing Ringing
Due to the leakage inductances of transformer 814 (FIG. 8A), voltages at the drains of NMOS transistors 804 and 816 can potentially ring to values substantially higher than the ideal value (e.g. twice times the battery voltage). To limit the extent of the ringing voltage, the CCFL system can include a snubbing circuit 913, as shown in FIG. 9. In snubbing circuit 913, capacitor 902, resistor 903, and diodes 904 and 905 are configured to maintain a nominal voltage at their common node N10. In one embodiment, this nominal voltage is approximately twice the battery voltage. However, if either of the drains of NMOS transistors 804/816 ring above that voltage, then diodes 904 and 905 forward bias and allow the ringing energy to charge capacitor 902. Resistor 903 bleeds off the extra ringing energy, thereby preventing the voltage at common node N10 from increasing substantially higher than the nominal voltage. The extra power dissipation is:
P(dissipated)=Vbatt2/Resistance (903)
For example, assuming that resistor 903 has a resistance of 3.9 kOhm and the battery voltage is 15 V, then the power dissipation of snubbing circuit 913 would be 58 mW or approximately 1% of the total input power. Thus, the value of resistor 903 can be optimized for a particular application to minimize dissipated power.
Note that the amount of ringing is a strong function of the operating frequency. Therefore, a user can advantageously select an appropriate resistance for resistor 852 such that the oscillator frequency is near the resonant frequency of the transformer LC network.
Multiple Tube Drive Circuit
Current LCD monitors may require multiple CCFL tubes to provide the high intensity light necessary for their intended application. Unfortunately, simply paralleling tubes with a single larger transformer is not advisable because differences in the load characteristics of the tubes may cause large mismatches in tube current and subsequent early tube failure. Alternatively, a single controller, single transformer can be used for each CCFL tube in the application; however, the cost of this type of application would soon become prohibitive.
In circuit 1400, the topology is substantially the same as for CCFL system 800 (see FIG. 8A). For example, the configuration and operation of PMOS transistor 803 and NMOS transistors 804 and 816 are identical to that in CCFL system 800. Moreover, the feedback loop for determining the current through CCFL tube 805 is identical to that in CCFL system 800. Note that the feedback loop need only be coupled to CCFL tube 805 because, as previously noted, the current in CCFL tube 1401 should be substantially identical to the current in the regulated tube, i.e. CCFL tube 805, as long as the parasitic capacitive paths are approximately equal for both tubes. A resistor 1402 can be sized to be substantially equal to the sum of the resistances of resistors 807 and 808, thereby ensuring that the impedances of CCFL tubes 805 and 1401 are equal.
The geometry of modified transformer 1410 is shown in greater detail in FIG. 15. In this geometry, a connection 1504, which is located between the two secondary windings 1501 and 1503, remains at a low voltage, e.g. ground. In contrast, the voltage outputs from secondary windings 1501 and 1502 are alternately a large positive voltage and a large negative voltage (e.g. +600 V and −600 V).
In one embodiment, connection 1504 is placed at approximately halfway between secondary windings 1501 and 1503. As long as the loads on the outputs of secondary windings 1501 and 1503 are substantially the same, this configuration eliminates the potential for arcing to occur between primary winding 1502 and secondary windings 1501 and 1503. Moreover, the highest voltages on the secondary windings occur as far away from each other as is possible, thereby also reducing the risk of arcing within the transformer.
Node 1504 is an ideal place to sense potential fault conditions caused by a missing tube or a marginal connection in the high voltage path where arcing may occur. For normal operation where the CCFL loads are approximately electrically equal the voltage at node 1504 remains close to ground. When a fault occurs in one of the secondary paths (such as a missing or broken CCFL) the voltage at node 1504 will deviate greatly from ground. By sensing the voltage at node 1504 through an appropriate resistor divider 1410 and rectifying diode 1411 (both shown in
Note that some components illustrated in various figures have been described as having exemplary resistances or capacitances. However, those skilled in the art will recognize that in other embodiments such components can have other values to modify performance outputs. Therefore, the present invention is not limited to the values of the disclosed embodiments.
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| Number | Date | Country | |
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| 20040066149 A1 | Apr 2004 | US |
