Patent Grant
10674283
This application claims the benefit, under 35 U.S.C. § 119, of German patent application DE 10 2017 203 630.3, filed Mar. 6, 2017; the prior application is herewith incorporated by reference in its entirety.
The invention relates to a method according to the preamble of the main method claim for distorting the frequency of an audio signal. Furthermore, the invention relates to a hearing apparatus according to the preamble of the main apparatus claim which operates according to this method.
In general, a “hearing apparatus” denotes an appliance which outputs a supplied audio signal or audio signal produced by recording ambient sound (also referred to as “input signal” below)—in an amplified manner and/or modified in any other way—as a sound signal in a form that is perceivable by the user (e.g. as air-borne sound fed into the auditory canal or as a body-borne sound). Hearing apparatuses also include, in particular, hearing aid appliances in addition to headphones. In turn, a “hearing aid appliance” denotes, in general, a portable hearing apparatus which serves to improve the perception of the ambient sound surging at the ear of a user. A subclass of the hearing aid appliances that is conventionally referred to “hearing aids” is configured to treat the hard of hearing who, in the medical sense, suffer from a loss of hearing.
In order to accommodate the numerous individual requirements of users, different constructions of hearing aid appliances, such as behind-the-ear (BTE) hearing aid appliances, hearing aid appliances with an external receiver (RIC, receiver in canal), in-the-ear (ITE) hearing aid appliances, or else concha hearing aid appliances or canal hearing aid appliances (ITE, CIC), are offered. The hearing aid appliances listed in an exemplary manner are worn on the outer ear or in the auditory canal. Moreover, bone conduction hearing aids, implantable or vibro-tactile hearing aids are commercially available. In these, the damaged hearing is stimulated either mechanically or electrically.
Of late, there are also hearing aid appliances for assisting humans with a normal sense of hearing in addition to the above-described conventional hearing aids. Such hearing aid appliances are also referred to as “personal sound amplification products” or “personal sound amplification devices” (abbreviated: “PSAD”). These PSADs serve to improve the normal human sense of hearing and are usually specialized for specific hearing situations (e.g. for the improved perception of animal noises, for an improved speech understanding in complex sound surroundings or for the targeted suppression of ambient noise).
In hearing apparatuses of the types described above, the supplied input signal is often reproduced in a frequency-distorted, in particular frequency-shifted and/or frequency-compressed, manner. In this case, the frequency distortion is often used, first, within the scope of feedback suppression and it facilitates an improved estimation of the feedback signal in this context and consequently facilitates better feedback suppression and reduced artifacts in the reproduced signal. Second, frequency distortion is often used in hearing aids to facilitate improved sound perception (in particular speech sound) for the hard of hearing by virtue of high-frequency noise components, which can often be perceived particularly badly by the hard of hearing, being mapped to lower frequencies.
However, the frequency distortion in both cases is not, as a rule, applied to the entire audio spectrum but it is only applied to a high-frequency signal component, which exceeds a predetermined cut-off frequency, of same.
A method according to the preamble of the main method claim and the hearing apparatus claim are known from European patent EP 2 244 491 B2, corresponding to U.S. Pat. No. 8,411,885. Here, an input signal is divided into a high-frequency signal component and a low-frequency signal component by a crossover filter, wherein the high-frequency signal component is frequency-distorted. The low-frequency signal component and the frequency-distorted high-frequency signal component are subsequently overlaid to form an output signal. European patent EP 2 244 491 B2 discusses the problem of the two signal components always having a certain spectral overlap on account of the inaccuracy of real crossover filters in the region of the cut-off frequency. On account of this overlap, it is known that the frequency distortion may lead to characteristic artifacts, particularly if the input signal has dominant frequencies (i.e. spectral peaks, in particular loud sinusoidal sounds) in the overlap region. This is because, in this case, one part of the dominant frequency with the high-frequency signal component is frequency-distorted while another part of the dominant frequency with the low-frequency signal component remains undistorted. Consequently, the dominant frequency of the input signal is mapped to two closely adjacent frequencies of the output signal, causing an audible beat that is often perceived as bothersome. According to European patent EP 2 244 491 B2, this problem is reduced by virtue of the cut-off frequency being displaced in such a way that artifacts in the output signal are reduced.
Moreover, international patent disclosure WO 00/02418 A1 has disclosed a hearing apparatus which divides an input signal into a low-frequency frequency band and a high-frequency frequency band by a crossover filter and which regulates the amplitude of the signals of the two frequency bands by two AGCs. The compression rate of the AGCs is set by way of a control signal, with an increase in the one compression rate causing a simultaneous decrease in the other compression rate. Subsequently, the two amplified frequency bands are overlaid using a summation device.
Finally, published European patent application EP 2 988 529 A1, corresponding to U.S. patent publication No. 2016/057548, discloses a method for suppressing acoustic feedback in a hearing aid appliance. In the method, a frequency range to be transferred by the hearing aid appliance is subdivided into two frequency ranges that are separated by a dividing frequency. A transfer function of a feedback path is estimated in one frequency range and evaluated in terms of its behavior at the dividing frequency. Depending on the result of the evaluation, the dividing frequency is lowered or raised and a phase and/or frequency modification is applied in the upper frequency range for the purposes of suppressing feedback.
The invention is based on the object of specifying a method for distorting the frequency of an audio signal, by which artifacts of the above-described type can be suppressed particularly effectively. Furthermore, the invention is based on the object of specifying a hearing apparatus, in which artifacts of the above-described type are suppressed particularly effectively.
According to the invention, the object is achieved by a method having the features of the main method claim. Furthermore, the object is achieved, according to the invention, by the main hearing apparatus claim. Advantageous configurations and developments, which in part are considered to be inventive in their own right, are specified in the dependent claims and the following description.
The method according to the invention serves to distort the frequency of an audio signal, in particular during the operation of a hearing apparatus. This audio signal, which is referred to as “input signal” below, is divided into a low-frequency signal component (referred to as “LF component” below) and a high-frequency signal component (referred to as “HF component” below). The frequency at which these two signal components adjoin one another is referred to as “cut-off frequency” below. Here, the terms “low-frequency signal component” (“LF component”) and “high-frequency signal component” (“HF component”) only describe the spectral position of these signal components relative to one another within the sense that the spectral center of the high-frequency signal component lies at a higher frequency than the spectral center of the low-frequency signal component.
Preferably, the LF component and the HF component completely cover the spectrum of the input signal. In this case, the input signal is therefore only subdivided into the two aforementioned signal components. However, in principle, further signal components, in addition to the LF component and the HF component, may be derived from the input signal within the scope of the invention, the further signal components lying above the HF component and/or below the LF component in the audio spectrum and the further signal components differing from the adjacent signal components in terms of the type of frequency distortion in each case.
According to the method, the HF component is frequency-distorted, in particular frequency-shifted or frequency-compressed. Here, the term “frequency shift” denotes a mapping of the HF component of the input signal to another spectral range with the same spectral extent. By contrast, the term “compression” denotes a mapping of the HF component to a spectral range with a smaller spectral extent. In principle, the frequency distortion within the scope of the invention may alternatively also consist of a “stretch”, i.e. a mapping of the HF component to a spectral range with a greater spectral extent, even if such a frequency distortion in hearing apparatuses is currently unusual.
The LF component is preferably not frequency-distorted, i.e. it is left unchanged in respect of its spectral position and extent. However, deviating therefrom, the LF component may also be subjected to frequency distortion within the scope of the invention, the frequency distortion, however, having a different characteristic in this case than the frequency distortion of the HF component.
According to the method, the LF component and the frequency-distorted HF component are overlaid to form an output signal.
In this case, optionally, one or more further signal processing steps, such as e.g. analog-to-digital conversion, frequency-dependent amplification, feedback suppression, etc., are also performed on the input signal prior to the frequency division into the LF component and the HF component, or between the frequency division and the overlay of the LF component and the frequency-distorted HF component (and in this case optionally before or after the frequency distortion). Likewise, the output signal may be subjected to further signal processing (e.g. digital-to-analog conversion and/or amplification) within the scope of the invention.
According to the invention, an associated gain factor is modified, i.e. increased or reduced, at least for a spectral edge region, containing the cut-off frequency, of the HF component and/or of the LF component, such that a level difference between a signal level of the LF component and a signal level of the frequency-distorted HF component is increased. To the extent that the change of the gain factor does not relate to the entire LF or HF component, but only to the edge region of same, use should be made of a signal level from this edge region when determining the level difference. In particular, the signal levels of the LF component and of the HF component at a dominant frequency are compared to one another for determining the regulating difference. The modification of the gain factor is expediently undertaken in such a way that audible beats in an overlap region between the HF component and the LF component are eliminated or at least reduced.
The invention is based on the discovery that the artifacts described at the outset become more clearly perceivable as the similarity of the signal levels of a dominant frequency of the input signal in the LF component and in the frequency-distorted HF component increases. As a result of the increase according to the invention in the level difference between the LF component and the HF component, in any case in the edge region of these signal components, the perceivability of artifacts is recognized to be reduced particularly effectively.
In principle, it is conceivable within the scope of the invention for the input signal to be divided into exactly two signal components (which themselves are not subdivided any further), namely the LF component and the HF component—for example, by means of a crossover filter as described in European patent EP 2 244 491 B2. In a preferred embodiment of the invention, however, a filter bank is used for dividing the input signal, the filter bank dividing the input signal into a multiplicity of frequency bands (i.e. substantially more than two frequency bands, and at least four frequency bands). In a typical embodiment of such a filter bank, the input signal is subdivided into, for example, 48 frequency bands.
According to the method, a number of high-frequency frequency bands thereof carry the HF component. Accordingly, these high-frequency frequency bands are frequency-distorted in the above-described manner. By contrast, a number of low-frequency frequency bands carry the LF component. Accordingly, these frequency bands are either not frequency-distorted or are frequency-distorted in a different way when compared to the HF component. Here, once again, the terms “high-frequency” (“HF”) and “low-frequency” (“LF”) should be understood to be relative specifications. Moreover, within the meaning of the explanations made above, there may be further frequency bands with frequencies above the “high-frequency” frequency bands or below the low-frequency frequency bands, the further frequency bands being assigned neither to the HF component nor to the LF component and being distinguished as further signal components as a consequence of a different type of frequency distortion.
Optionally, the edge region of the high-frequency signal component is formed by a subset of the high-frequency frequency bands which adjoin the low-frequency frequency bands. Additionally, or as an alternative thereto, the edge region of the low-frequency signal component is formed by a subset of the low-frequency frequency bands which adjoin the high-frequency frequency bands.
Here, the phrase “subset of frequency bands” denotes a number of frequency bands which is smaller than the overall number of frequency bands of the associated signal component and which may also contain only a single frequency band in a limiting case. In fact, this limiting case, in which the respective edge region of the HF or LF component is formed by a single frequency band, constitutes a preferred configuration of the invention. Within this sense, the plural “frequency bands” should be understood to the effect of comprising the case of a single frequency band.
The respective edge region and the frequency bands assigned thereto are distinguished by virtue of—in contrast to the remaining frequency bands of the HF or LF component—the gain factor for increasing the level difference relative to the signal level of the respective other signal component only being modified in the frequency bands of the respective edge region.
In particular, the edge region of the LF component and/or of the HF component is selected in such a way that its spectral extent contains the spectral overlap region of the LF component and of the HF component. To the extent that the input signal is divided into a multiplicity of frequency bands, the respective edge region is formed, in particular, by those frequency bands which contain the overlap region.
In expedient embodiments of the invention, the edge region, in which the gain factor is modified for increasing the level difference, is only defined for one of the two signal components (i.e., only for the HF component or only for the LF component), while the gain factor is kept constant in the respective other signal component. However, deviating therefrom, an edge region is defined in each case for both the LF component and the HF component in a particularly advantageous embodiment of the invention. Here, the gain factor in these two edge regions is always modified in the opposite sense. Consequently, the gain factor is increased in the edge region of a first of the two signal components (i.e., the HF component or the LF component), while the gain factor in the edge region of the second signal component (i.e., the LF component or the HF component) is reduced.
In a particularly advantageous variant of the invention, the gain factor for the second signal component is reduced in such a way here that this compensates the increase of the gain factor for the first signal component. Thus, expressed differently, the gain factors in the two edge regions are modified in the opposite sense so that the signal level, averaged over the two edge regions, or the signal power, averaged over the two edge regions, remains constant (i.e., uninfluenced by the modification of the gain factor). This leads—particularly in the case of a very tonal nature of the input signal in the overlap region of the HF component and of the LF component (i.e. if a very dominant frequency is present in this overlap region)—to the change as gain factor according to the invention in the output signal not being perceivable or being perceivable only to a very small extent, especially since the perceivable loudness of the dominant frequency is not influenced, or is only influenced to a very small extent, by the level change.
Consequently, the change of the gain factor leads to significant reduction or even elimination of artifacts of the frequency distortion without having a negative influence, in turn, on the reproduction quality of the input signal. Specifically, sinusoidal tones in the surroundings of the cut-off frequency are reproduced with virtually the same loudness as in conventional methods, wherein, however, the beats of these sinusoidal tones are completely, or at least largely, eliminated as a result of the frequency distortion.
In an advantageous development of the invention, the increase in the level difference according to the invention is not undertaken without conditions but only if this is really expedient (or only to the extent that this is really expedient), namely if audible artifacts are to be expected in the output signal (or in correspondence with the strength of the artifacts to be expected). It is recognized that audible artifacts are to be expected when the input signal in the spectral overlap region of the HF component and of the LF component has a high tonality; i.e., if dominant frequencies (in particular loud sinusoidal tones) are present in this overlap region. Therefore, a characteristic is captured in this development of the method, the characteristic being characteristic for the tonality of the input signal in the overlap region (which therefore, expressed differently, forms an estimate or comparison value for the tonality of the input signal in the overlap region).
The change according to the invention in the gain factor and hence the increase in the level difference between HF component and LF component are undertaken here according to the method in a manner depending on this characteristic. In particular, the increase in the level difference is only undertaken when this characteristic satisfies a predetermined criterion, in particular if it exceeds a predetermined threshold. In an alternative embodiment of the invention, the increase in the level difference is weighted depending on this characteristic (in linear or nonlinear fashion). The characteristic that is characteristic for the tonality of the input signal in the overlap region is preferably ascertained by auto-correlating the input signal in the overlap region in this case. In particular, the characteristic is formed by the absolute value of the autocorrelation function (which has complex values in the mathematical sense).
In general, the hearing apparatus according to the invention is configured to automatically carry out the method according to the invention described above. The embodiments and developments of the method described above correspondingly conform to associated embodiments and developments of the apparatus, wherein advantages of these method variants may also be transferred to the corresponding embodiments of the hearing apparatus. Specifically, the hearing apparatus according to the invention contains a frequency splitter which is configured to divide a reception signal into a low-frequency signal component (LF component) and a high-frequency signal component (HF component), wherein these two signal components adjoin one another at a cut-off frequency. Furthermore, the hearing apparatus contains a signal processor, which is configured to distort the frequency of the high-frequency signal component, and a synthesizer which is configured to overlay the low-frequency signal component and the frequency-distorted high-frequency signal component for forming an output signal.
According to the invention, the signal processor is configured to modify a gain factor, at least for a spectral edge region, containing the cut-off frequency, of the HF component and/or of the LF component, such that a level difference between a signal level of the LF component and a signal level of the frequency-distorted HF component is increased.
Preferably, the frequency splitter is formed by an (analysis) filter bank which is configured to split the input signal into a multiplicity of frequency bands. In this embodiment, the synthesizer is correspondingly formed by a (synthesis) filter bank which then combines the frequency bands after the frequency distortion (and optional further signal processing steps) to form the output signal. In view of embodiment variants of the signal processor, reference is otherwise made in an analogous manner to the explanations, made above, in respect of the method according to the invention.
The hearing apparatus according to the invention is, in particular, a hearing aid appliance and, once again in this case, preferably a hearing aid embodied to treat the hard of hearing.
Other features which are considered as characteristic for the invention are set forth in the appended claims.
Although the invention is illustrated and described herein as embodied in a method for distorting the frequency of an audio signal and a hearing apparatus operating according to this method, it is nevertheless not intended to be limited to the details shown, since various modifications and structural changes may be made therein without departing from the spirit of the invention and within the scope and range of equivalents of the claims.
The construction and method of operation of the invention, however, together with additional objects and advantages thereof will be best understood from the following description of specific embodiments when read in connection with the accompanying drawings.
Parts and variables that correspond to one another are provided in each case with the same reference sign in all figures.
Referring now to the figures of the drawings in detail and first, particularly to
The input transducer 4 (formed in an exemplary manner by a microphone in the present case) converts an incoming sound signal Si from the surroundings into an (original) input signal Ei.
For the purposes of suppressing acoustic feedback, an electrical compensation signal K is subtracted from the original input signal Ei in the subtraction device 6, the compensation signal being produced in the electrical feedback path 16. A (compensated) input signal Ek, which is supplied to the (analysis) filter bank 8, emerges from the subtraction of the input signal Ei and the compensation signal K.
In the filter bank 8, the input signal Ek is divided spectrally into a multiplicity of frequency bands Fj. Here, the parameter j is a counter, by which the frequency bands Fj are numbered in sequence. In the simplified example according to
In the signal processor 10, the input signal Ek that has been split into the frequency bands Fj is processed in a frequency-band-specific manner. A signal P that has been processed by the signal processor 10 is supplied—once again spectrally divided into frequency bands Fj′ (j=1, 2, . . . , 6)—to the (synthesis) filter bank 12, which combines (overlays) the frequency bands Fj′ to form an electrical output signal A.
The output signal A is supplied first to the output transducer 14 (formed e.g. by a loudspeaker or a “receiver”), which converts the output signal A into an outgoing sound signal Sa.
Second, the output signal A is supplied via the electrical feedback path 16 to the adaptive filter 18 which ascertains the compensation signal K therefrom. The compensated input signal Ek is additionally supplied to the adaptive filter 18 as a reference variable.
During the operation of the hearing aid 2, the sound signal Sa is either output directly into the auditory canal of a hearing aid wearer or supplied to the auditory canal via a sound tube. Particularly in the case of embodiments of the hearing aid 2 in which the hearing aid 2 itself is arranged in the auditory canal, some of the output sound signal Sa is unavoidably coupled back to the input transducer 4 as a feedback signal R via an acoustic feedback path 20 (e.g. via a vent channel of the hearing aid 2 or via body-borne sound), the feedback signal R overlaying the ambient sound at the input transducer to form the incoming sound signal Si.
Here, the sound signals Si, Sa and the feedback signal R are genuine sound signals, in particular air-borne sound and/or body-borne sound. By contrast, the input signals Ei, Ek, the processed signal P, the output signal A and the compensation signal K are audio signals, i.e. electrical signals that transport sound information.
As mentioned, the relevant audio signals, namely the input signal Ek and the processed signal P, are guided in spectrally split fashion in the frequency bands Fj and Fj′ in the region between the analysis filter bank 8 and the synthesis filter bank 12.
The hearing aid 2 is a digital hearing aid in particular, in which the signal processing in the signal processor 10 is effectuated by digital technology. In this case, the audio signal is digitized prior to the signal processing by an analog-to-digital converter 22 and converted back into an electrical analog signal after the signal processing by a digital-to-analog converter 24. In the illustrated example, the analog-to-digital converter 22 is disposed immediately upstream of the filter bank 8 and consequently acts on the compensated input signal Ek, while the digital-to-analog converter 24 is disposed downstream of the filter bank 12. In this case, the electrical feedback path 16 guides the output signal A and the compensation signal K in the form of analog signals.
As an alternative thereto, the analog-to-digital converter 22 is connected between the input transducer 4 and the subtraction device 6, and consequently acts on the original input signal Ei (not illustrated). In this case, the electrical feedback path 16 expediently guides the output signal A and the compensation signal K in the form of digital signals.
In a further embodiment (likewise not illustrated) of the hearing aid 2, the subtraction device 6 is disposed downstream of the analysis filter bank 8. Here, the frequency bands Fj′ or the output signal A that was spectrally split by further frequency analysis are supplied to the adaptive filter 18. The adaptive filter 18 comprises an appropriate number of channels.
The signal processor 10 subjects the input signal Ek supplied in the frequency bands Fj to multifaceted signal processing processes, as is typical for hearing aids, in particular a frequency-band-specific varying gain in order to adapt the reproduction of the input signal Ei to the individual requirements of a hearing aid user who is hard of hearing and consequently make the reproduction audible to the best possible extent for the user. Moreover, the signal processor 10 carries out a frequency distortion which decorrelates the output signal A from the input signal Ei to obtain improved feedback suppression.
In order to clarify the effect of the frequency distortion,
In
In addition to the frequency bands Fj supplied to the signal processor 10,
Signal processing processes which modify the respective gain factors for the individual frequency bands F1′-F6′ relative to one another were not imaged in the schematic illustration according to
The bandwidth of the frequency bands F1-F6 and of the corresponding frequency bands F1′-F6′ is given, in particular, by the full width at half maximum. The level of the half maximum in the illustration according to
It is furthermore clear from
In order to avoid artifacts of the type set forth at the outset in the output signal A during the operation of the hearing aid 2, and hence when the frequency distortion according to
In a first step 30 of the aforementioned method (which constitutes a part of a method for operating the hearing aid 2), the signal processor 10 receives the input signal Ek, which, as described above, was divided by the filter bank 8 into the frequency bands Fj and hence, implicitly, also into the signal components LF and HF.
In a subsequent step 32, the signal processor 10 in each case forms the autocorrelation function over the cut-off-near frequency bands F3′ and F4′ (and consequently over the respective edge regions RL and RH) in order to obtain a characteristic which represents a quantitative measure for the tonality of the input signal Ek in the edge regions RL and RH.
As mentioned above, the term “tonality” denotes a property of the input signal Ek, which characterizes the dominance of an individual frequency f0 (
Here, the method makes use of the discovery that the autocorrelation function represents a good measure for the tonality. Particularly in preferred embodiments of the invention, in which the filter bank 8 is a DFT modulated filter bank (i.e. a filter bank based on a discrete Fourier transform) or a similar implementation, a sinusoidal signal in the frequency bands F3 and F4 corresponds to a rotating complex phasor, which, in the case of a constant frequency, rotates with constant angular jumps between successive time steps. In a one-tap-autocorrelation, as is preferably determined in step 32 of the method, this rotating phasor is mapped onto a complex phasor which has a constant phase angle corresponding to the angular step.
The absolute value of this complex-valued autocorrelation function is used here by the signal processor 10 as a measure for the tonality. Alternatively, the variance of the complex phasor or the phase angle is used as a measure for the tonality, wherein the fact is exploited that a small variance indicates a stable frequency, and consequently a high tonality. The signal processor 10 derives the absolute value of the dominant frequency f0 from the phase angle of the complex-valued autocorrelation function by virtue of the signal processor dividing this phase angle by the absolute value of the time interval between two time steps (specifically: f0=φ/(π·TS), where φ denotes the phase angle and TS denotes the aforementioned time interval; here, the dominant frequency f0 is related to the band center of the respective frequency band T3 or T4).
In a step 34, the frequency distortion is carried out by the signal processor 10 by virtue of—as illustrated in
In a step 36, the signal processor 10 checks whether the measure ascertained previously for the tonality, i.e., for example, the absolute value of the ascertained autocorrelation function in the frequency bands F3 and F4, is below a predetermined threshold.
For as long as this is the case (Y), the signal processor 10 identifies this as a sign that no bothersome artifacts are to be expected by the frequency distortion. Accordingly, the signal processor 10 jumps to a step 38 of the method procedure in this case by virtue of outputting the frequency-distorted signal P (optionally after performing further signal processing steps) in frequency bands Fj′ to the filter bank 12 for the purposes of synthesizing the output signal A.
If, otherwise, the check carried out in step 36 yields that the measure for the tonality is not below the predetermined threshold (N), the signal processor 10, in a step 40, estimates the level difference ΔL (
In a subsequent step 42, the signal processor 10 checks whether the predetermined level difference ΔL exceeds a predetermined limit value.
For as long as this is the case (Y), the signal processor 10 recognizes this as a sign that bothersome artifacts as a consequence of the frequency distortion should not be expected on account of the already innately high level difference ΔL. Accordingly, the signal processor 10 once again jumps to step 38 in the method procedure in this case.
Otherwise (N), i.e. if the check carried out in step 42 turns out negative and, accordingly, the level difference ΔL does not exceed the threshold, the signal processor 10 adapts the gain factors of the cut-off-near frequency bands F3′ and F4′ in the opposite sense in a step 44 such that an increased level difference ΔL′ (ΔL′=|L1′−L2′|; see
Here, in particular, the signal processor 10 calculates this change of the gain factors in such a way that the level increase and the level reduction in the cut-off-near frequency bands F3′ and F4′ compensate one another, i.e. in such a way that the adapted signal levels L1′ and L2′ of the frequency bands F3′ and F4′ at the dominant frequency f0 or f0′ in sum (or on average) corresponding to the corresponding levels L1 and L2, respectively, before the level adaptation (L1′+L2′=L1+L2). Deviating from the simple formation of the sum or average, the amplitude frequency response of the affected frequency bands is also taken into account in a more developed embodiment of the method.
Subsequently, the signal processor 10 once again jumps to step 38 in the method procedure.
What is achieved by the change, in the opposite sense, in the gain factors in the cut-off-near frequency bands F3′ and F4′ performed in step 44 is that the dominant tone in the output signal A can be heard at approximately the same strength as if the level adaptation had not been carried out in step 44. Depending on in which one of the signal components LF and HF the dominant frequency f0 is more strongly pronounced, the dominant tone is heard in this case either with the non-displaced frequency f0 or the displaced frequency f0′. However, as a consequence of the increased level difference ΔL′, bothersome artifacts in the form of beats between the frequencies f0 and f0′ are suppressed in this case.
Numerous alternative embodiments of the method are possible within the scope of the invention. By way of example, the frequency distortion (step 34) may also be performed at a different point in the method procedure, e.g. after the level change (step 42). Furthermore, multifaceted further signal processing steps may be undertaken between steps 30 and 38 within the scope of the invention, in particular steps for the frequency-selective amplification of the input signal Ek, for noise suppression, etc.
The effect of the level change, according to the invention, in the cut-off-near frequency bands F3′ and F4′ is once again clarified on the basis of
The invention becomes particularly clear on the basis of the exemplary embodiments described above. However, it is equally not restricted to these exemplary embodiments. Rather, numerous further embodiments of the invention may be derived from the claims and the above description.
The following is a summary list of reference numerals and the corresponding structure used in the above description of the invention:
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