Other particularities and benefits of this invention will appear in the following description of non-limiting examples of implementation, referring to the attached drawings, in which:
Consider a receiver capable of receiving digital signals carrying multiplexed symbols according to OFDM technology. Said receiver may, for example, be part of a communication station of a wired communication system. In another embodiment illustrated in
The stream of data received at the receiver in
For each OFDM symbol gm received having a duration Ts=Tu+Tg, a first fast Fourier transform, called LFFT 8 (for Left FFT), is performed on a portion of the signal carrying that symbol. The calculation window for LFFT 8, i.e., the time span having a duration Tu over which the LFFT 8 is performed, can be positioned in various ways. Said positioning may, for example, be the result of one of the methods described in the prior art. An advantageous example of positioning such a window will be described in greater detail below.
In addition, a second fast Fourier transform, called RFFT 9 (for Right FFT), is performed on the portion of the signal carrying the same OFDM symbol. The position of the calculation window of the RFFT 9 is established relative to the previously determined position of the LFFT 8. In effect, a time lag 10 is selected, to be observed between the LFFT 8 and the RFFT 9.
This time lag 10 may be fixed. In this case, it will be advantageously selected based on the characteristics of the environment containing the communication channel over which the signal is sent. In effect, some environments, for example relatively dense urban environments, foster multiple reflections and diffractions of radio signals, which favors the dispersal of the paths of propagation over time. However, dispersal of paths of propagation is a factor of inter-symbol interference (ISI), as explained above. In order to limit ISI, a relatively long time lag 10 is selected in this type of environment so that the LFFT 8 and the RFFT 9 calculation windows are spread over a sufficient period of time to cover the principal paths of propagation followed by the signal portion involved.
Inversely, other environments do not foster the dispersal of paths of propagation. These include, for example, rural type environments, where signals are most often propagated in a line of sight. This is also the case in wired systems, where the propagation times generally vary little. For such environments, a shorter time lag 10 can be selected, for example on the order of the duration of just a few binary elements.
It is also possible to perform other FFTs in addition to the LFFT 8 and the RFFT 9. In this case, these FFTs are also differently positioned in time, each exhibiting a certain time lag relative to the LFFT 8, for example.
In one advantageous embodiment of the invention, the LFFT 8 and the RFFT 9, as well as any other FFTs, are dynamically positioned in time based on an analysis of the pulse response 11 of the communication channel traveled by the signal sent. This embodiment is particularly interesting when the communication channel is a radio channel and the communication system to which the receiver belongs is mobile.
The pulse response 11 of the communication channel is calculated by the receiver. Advantageously, such a calculation is performed several times, for example periodically, in order to have an up-to-date estimate that takes account of the changes in the profile of propagation. In this case, the calculation can be performed for each OFDM symbol transmitted, i.e., with a period approximately equal to T,. However, a longer period may also be used, when the characteristics of the channel vary less.
The pulse response 11 of the communication channel is calculated according to a conventional method. For example, the receiver involved is a rake receiver. Such a receiver estimates the pulse response of the radio channel by a series of peaks, each peak appearing with a delay corresponding to the propagation time along a specific path and having a complex amplitude corresponding to the attenuation and dephasing of the signal along that path (instantaneous fading).
A pilot channel can be provided to estimate the pulse response in the form of a succession of peaks. The pulse response is estimated by means of an adapted filter. For example, when a coded distribution access technique is used, such as the CDMA (Code Division Multiple Access) used in the UMTS system, the filter is adapted to a pilot spread code with which the transmitter modulates a sequence of known symbols, for example 1 symbols. The positions of the maximums of the output of this adapted filter give the delays used in the fingers of the rake receiver, and the associated complex amplitudes correspond to the values of said maximums.
The output of the adapted filter is then analyzed. This analysis consists of statistical calculations on the output of the adapted filter 11 to determine the delays (τi)0≦i≦M-1 associated with the M paths or echoes observed, as well as the average receiving energies (Ei)0≦i≦M-1 associated with said paths.
The highest-energy paths detected correspond to principal paths dependent upon the environment. The highest-energy path will correspond, for example, to a direct path between the transmitter and the receiver if they are in direct visibility to one another. The other principal echoes will schematically be those that yield the fewest reflections and diffractions between the transmitter and the receiver.
The energy Ei associated with a path is the mathematical expectation of the square of the module of the instantaneous amplitude Ai(t) of reception on the corresponding path of propagation.
In this embodiment, the estimate of the pulse response 11 is used to position the LFFT 8 and RFFT 9 calculation windows. For example, the LFFT 8 window is positioned relative to the first detected path. For example, it begins with a delay approximately equal to To relative to a time reference, so as to cover the energy peak relating to the first detected path. This is what is shown in
In addition, as was described above, the calculation window of the RFFT 9 is positioned relative to that of the LFFT 8 by a time lag 10. Advantageously, this time lag 10 is adjusted based on the estimated pulse response 11, as illustrated in
In the example illustrated in
Of course, other positionings of the fast Fourier transform calculation windows can be applied. For example, the RFFT 9 can be calculated on a window that ends after τ2, so as to ensure that the echo of the signal portion involved corresponding to the last path on the radio channel will actually be received this window.
Referring to
Furthermore, for each FFT calculated, an estimate is made of the radio channel over which the signal is sent. So an estimate of channel 12 is made at the output of the calculation module of the LFFT 8 and an estimate of channel 13 is made at the output of the calculation module of the RFFT 9. Numerous known methods can be used to estimate the channel. For example, the least squares method can be used.
According to this last method, the estimate module of channel 12 estimates parameters r0, r1, . . . , rn corresponding to disturbances appearing on the radio channel, such that {circumflex over (X)}k=r0Xk+r1Xk-1+. . . rnXk-n+wk, where k is an integer greater than n, Xi is the data corresponding to the ith OFDM symbol transmitted (Xi may, for example, be the Xn of
Similarly, the estimate module of channel 13 estimates parameters r′0, r′1, . . . , r′n, such that {circumflex over (X)}k=r′0Xk+r′1Xk-1+. . . r′nXk-n+w′k, where k is an integer greater than n, {circumflex over (X)}{circumflex over (′)}k is the estimate corresponding to the kth OFDM symbol received, which follows the calculation of the RFFT 9, and w′k is a second estimate of the noise affecting the radio channel over which the OFDM symbols are being sent.
A combination module 14 is then used to make an estimate of the data corresponding to the OFDM symbols sent, on the basis of the elements sent by the channel 12 and 13 estimate modules. If the least squares method has been used to estimate the channel, the module 14 can then proceed as follows. A vector {circumflex over (X)} is defined as a concatenation of the 2k+2 estimates {circumflex over (X)}0, {circumflex over (X)}1, . . . , {circumflex over (X)}k and {circumflex over (X)}′0, {circumflex over (X)}′1, . . . , {circumflex over (X)}′k made after the channel estimates made respectively by the modules 12 and 13. In addition, a first convolution matrix R is defined based on the parameters r0, r1, . . . , rn estimated by the channel 12 estimate module. This matrix R, whose dimensions are (k+1)·(k+1), has the following structure:
Similarly, a second convolution matrix R′ is defined based on the parameters r′0, r′1, . . . , r′n estimated by the channel 13 estimate module. This matrix R′, whose dimensions are (k+1)·(k+1), has the following structure:
Lastly, a vector X is defined, which includes all the k+1 quantities of data X0, X1, . . . , Xk corresponding to the OFDM symbols sent, which need to be determined.
It is then possible to write the following equation: {circumflex over (X)}=M·X, where M represents a column concatenation matrix of the block matrices R and R′, i.e.,
The noises wk and w′k introduced above are delayed from one another due to the lag 10 existing between the two FFTs performed, so they are independent and do not enter into the equation system to detect the value of the symbols sent.
The vector X can then be estimated as the product of (MHM)−1·MH·{circumflex over (X)} where the operator (·)H is the conjugated transpose.
Of course, many methods other than the least squares method can be used to estimate the channel. In this case, the estimate of the vector X will be adapted to the method used (for example, one of the following methods: a posteriori maximum, MLSE (Maximum Likelihood Sequence Estimator), Viterbi algorithm, etc.).
Once the combination is performed by the module 14, an estimate is obtained of the data X0, X1, . . . , Xk corresponding to the signals sent. The reliability of this estimate is greater than that of those obtained by conventional methods because it relies on a combination of estimates made as a result of two different observations. Furthermore, as was indicated above, the time lag between the FFTs makes it possible to achieve some data redundancy which is taken advantage of in the combination 14, all the while ensuring that the final estimate is based on the highest-energy, and therefore the most significant, echoes.
It will be noted that the principles discussed above can be implemented by using a computer program containing the corresponding instructions and run, for example, by a processing unit in the receiver involved.
Number | Date | Country | Kind |
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0406128 | Jun 2004 | FR | national |
Filing Document | Filing Date | Country | Kind | 371c Date |
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PCT/EP05/05336 | 5/17/2005 | WO | 00 | 8/13/2007 |