Microcontroller, switched-mode power supply, ballast for operating at least one electric lamp, and method of operating at least one electric lamp

Abstract

The invention relates to a microcontroller (MC) having at least one device (G) for generating pulse-width modulated or frequency modulated control signals for a switched-mode power supply. The device (G) has a further device (SQ1, SS1) for the alternate charging and discharging an electric charge store (C27) that can be connected to the microcontroller (MC), control means for this device (SQ1, SS1) for controlling the charging and discharging operations, and an evalutor for evaluating the time periods which are needed for the individual charging and discharging operations to generate pulse-width modulated or frequency modulated control signals. The microcontroller (MC) generates finely graduated, frequency modulated or pulse-width modulated control signals which are independent of the operating cycle frequency of the microcontroller (MC).

Description

The invention relates to a microcontroller according to the preamble of patent claim 1 , a switched-mode power supply according to patent claim 10 , a ballast for at least one electric lamp according to the preamble of patent claim 11 , and a method of operating at least one electric lamp according to the preamble of patent claim 25 .

I. TECHNICAL FIELD

In particular, the invention relates to a microcontroller which is provided to drive the switching transistors of a switched-mode power supply, to be specific preferably of a switched-mode power supply for operating electric lamps. In the switched-mode power supplies normally used to operate electric lamps, there are normally inverters, in particular half-bridge, full-bridge and push-pull inverters, and also step-up converters and step-down converters. Modern electronic ballasts for operating electric lamps generally have an inverter to produce a high-frequency alternating current for lamp operation and often also have a step-up converter as a DC supply for the inverter. The switching transistors of the inverter and of the step-up converter are driven by means of driver circuits, which are constructed as integrated circuits designed using analog techniques. In addition, modern electronic ballasts for electric lamps also contain a microcontroller, which is generally used for communication with a control unit arranged outside the ballast and for evaluating the control commands from this control unit for lamp operation and also for monitoring the lamp operation.

II. PRIOR ART

The European publication EP 0 708 579 A1 discloses a circuit arrangement for operating a high-pressure discharge lamp using an inverter, whose switching transistors have pulse-width modulated control signals applied to them by means of a microcontroller and a downstream integrated driver circuit. The pulse-width modulated control signals are generated with the aid of the auto-reload timer implemented in the microcontroller. In principle, this is a counting mechanism which operates at the operating cycle frequency of the microcontroller. During the counting operation, the reaching of a reference value and the overflow of the counting mechanism are monitored. During the time period which is needed to reach the reference value, the output of the auto-reload timer is at the “high” logic level, and during the time period which the counting mechanism needs to count up from the reference value until the counter overflows, the output of the auto-reload timer is at the “low” logic level. In this way, with the aid of the microcontroller, pulse-width modulated control signals for the inverter are generated, in order to permit lamp operation with a frequency-modulated voltage in a small frequency range and with a comparatively low number of discrete frequencies.

However, in this way, using cost-effective microcontrollers, no finely graduated pulse-width modulation control nor any finely graduated frequency control of the inverter can be carried out, since the smallest possible, adjustable change in the pulse width or in the frequency of the control signal which can be generated by the counting mechanism explained above is limited by the operating cycle frequency of the microcontroller and by the memory size of the counting mechanism. In order, for example, to permit dimming operation of fluorescent lamps on an electronic ballast by means of frequency modulation of the lamp current, frequency changes in steps of approximately 50 Hz are required in the frequency range from about 30 kHz to 100 kHz. If this frequency modulation is to be generated with the aid of the auto-reload timer, a microcontroller with an operating cycle frequency of more than 100 MHz is needed for this purpose. However, for cost reasons, such microcontrollers cannot be used in electronic ballasts for lamp operation.

III. SUMMARY OF THE INVENTION

It is an object of the invention to provide a microcontroller with an improved device for pulse-width modulation control and/or frequency control of a switched-mode power supply.

A further object of the invention is to provide a switched-mode power supply provided with a microcontroller with improved driving of the switching means of the switched-mode power supply.

In addition, it is an object of the invention to provide a ballast equipped with an inverter for operating at least one electric lamp which, with the aid of a microcontroller, permits finely graduated frequency control and/or pulse-width modulation control of the switching means of the inverter.

Furthermore, it is an object of the invention to specify an improved method for generating frequency control signals and/or pulse-width modulation control signals for the switching means of an inverter of a ballast for operating electric lamps by means of a microcontroller.

The aforementioned objects of the invention are achieved by the features of the independent patent claims 1 , 10 , 11 and 25 , respectively. Advantageous refinements of the invention are described in the dependent patent claims.

The microcontroller according to the invention has at least one device for pulse-width modulation control and/or frequency control of a switched-mode power supply, this device having

a device for the alternate charging and discharging of a charge store that can be connected to the microcontroller or integrated into the microcontroller,

control means for the device for controlling the charging operations and/or the discharging operations, and

evaluation means which are used to evaluate the time periods required for recharging the charge store between different charge states and, on this basis, to generate a pulse-width modulation control signal and/or frequency control signal.

The device for the alternate charging and discharging of a charge store, and its control means, permit controlled charging operations and discharging operations to be carried out alternately with each other on a charge store and, with the aid of the evaluation means, the evaluation of the time periods which are needed for the partial charging and discharging of the charge store and on this basis the generation of a pulse-width modulation control signal and/or frequency control signal. Even if the microcontroller according to the invention has only a low operating cycle frequency, it can be used to implement finely graduated pulse-width modulation control and/or frequency control of a switched-mode power supply, since the device for the alternate charging and discharging of a charge store operates independently of the operating cycle frequency of the microcontroller.

The device for the alternate charging and discharging of a charge store advantageously comprises a controllable current source for applying an adjustable charging current to the charge store, and a controllable current sink for applying an adjustable discharging current to the charge store. As a result, the individual charging and discharging operations can be controlled independently of one another. In addition, the controllable current source and current sink can be produced in a known way by means of semiconductor technology and integrated into the microcontroller. In order to permit very fine graduation of the pulse-width modulation control signals and/or frequency control signals, the controllable current source and the controllable current sink are formed in such a way that their settings can be varied in relation to a reference current level, in each case with a resolution of at least 8 bits. The reference current level for the charging and the discharging current is in this case advantageously predefined with the aid of a nonreactive resistor. The control means provided for the device for the alternate charging and discharging of a charge store is advantageously at least one read/write memory. The content of the read/write memory can be updated continuously, for example under program control, and can be read in order to control the device for the alternate charging and discharging of a charge store. The control means advantageously comprise a switching means which are used to switch over the device for the alternate charging and discharging of a charge store from charging to discharging of the charge store when a first voltage value is reached, and to switch over the device for the alternate charging and discharging of a charge store from discharging to charging of the charge store when a second, lower voltage value is reached. With the aid of the switching means, the device for the alternate charging and discharging of a charge store is simply forced into mutually alternating charging and discharging operations, so that the charge state of the charge store is subjected to an incessant oscillation, which can be evaluated in order to generate frequency control signals and/or pulse-width modulation control signals. The first or the second voltage value can advantageously be adjusted by means of a read/write memory. As a result, the aforementioned oscillation of the charge state of the charge store can be influenced under program control.

The microcontroller according to the invention advantageously has a frequency divider or a pulse divider which, at its input, detects the changeover of the device for the alternate charging and discharging of a charge store from discharging to charging or from charging to discharging and divides the input signal into signals for the alternating control of alternately switching means of the switched-mode power supply. With the aid of the frequency divider or pulse divider, the oscillation of the charge state of the charge store can be evaluated in order to generate frequency control signals and/or pulse-width modulation control signals for the switching means of a switched-mode power supply with alternately switching means.

The microcontroller according to the invention additionally advantageously has interfaces for registering external signals or data and has a device for evaluating the external signals or data and for the program-controlled determination of actuating values for controlling the device for the alternate charging and discharging of a charge store. As a result, a control loop for the oscillation of the charge state of the charge store can be implemented on the basis of external operating parameters and the actuating values derived therefrom.

The switched-mode power supply according to the invention is distinguished by a microcontroller as claimed in one or more of claims 1 to 9. As distinguished from the previously conventional switched-mode power supplies, in the case of the switched-mode power supply according to the invention, the signals for pulse-width modulation or for frequency control of the switching transistors of the switched-mode power supply are generated by the microcontroller. The corresponding control signals are forwarded to the control electrodes of the switching transistors of the switched-mode power supply from the microcontroller, directly or if appropriate via driver circuits. As has already been mentioned above, these control signals are independent of the operating cycle frequency of the microcontroller.

The ballast according to the invention for operating at least one electric lamp has an inverter, at least one load circuit coupled to the inverter and having terminals for the at least one electric lamp, a control circuit for controlling the switching means of the inverter and a DC supply circuit for the inverter, the control circuit comprising a microcontroller having a device for pulse-width modulation control and/or frequency control of the switching means of the inverter. According to the invention, the device for pulse-width modulation control and/or frequency control of the switching means of the inverter has

a device for the alternate charging and discharging of a charge store,

control means for the device for the alternate charging and discharging of the charge store, which are used to control the charging operations and/or the discharging operations, and

evaluation means, which are used to evaluate the duration of the alternate charging and discharging operations of the charge store and on this basis to generate a frequency control signal and/or a pulse-width modulation control signal for controlling the switching means of the inverter.

The device for the alternate charging and discharging of a charge store, the charge store and the control means for the device for the alternate charging and discharging of a charge store form an oscillator, which operates independently of the operating cycle frequency of the microcontroller. The oscillations of the charge state of the charge store are evaluated with the aid of the evaluation means in order to generate frequency control signals and/or pulse-width modulation control signals for the inverter.

As a result of the aforementioned features of the ballast according to the invention, it becomes possible, with the aid of a relatively simple and cost-effective microcontroller, to implement all the essential control functions of a modern, dimmable ballast. In particular, these are the power factor correction, the control of the inverter, the control of the lamp electrode heating, the regulation of the load circuit, the brightness control of the lamps and monitoring of the lamp operation. As compared with previously conventional ballasts, which either have a freely oscillating inverter or an inverter controlled externally by means of an integrated circuit, and are able to ensure monitoring of the lamp operation only with numerous additional components, the ballast according to the invention manages with comparatively few additional components. Most functions in the ballast according to the invention are performed by the microcontroller. For example, end-of-life monitoring of the lamp can be implemented particularly simply with the ballast according to the invention, but is very complicated and expensive in ballasts according to the prior art.

For the alternate control of the switching means of the inverter, the device for pulse-width modulation control and/or frequency control advantageously has a frequency divider or a pulse divider which, at its input, detects the changeover of the device for the alternate charging and discharging of a charge store from discharging to charging or from charging to discharging of the charge store, and divides the input signal into signals for the alternating control of the switching means of the inverter.

In order to apply a heating current to the lamp electrodes, the ballast according to the invention advantageously has a heating device equipped with a controllable switching means, and the microcontroller has a comparator, which compares the charge state of the charge store with a reference value for the lamp electrode heating and which is used to generate a control signal for the pulse-width modulation of the controllable switching means of the heating device. As a result, the oscillation of the oscillator mentioned above can be evaluated not only for the purpose of controlling the inverter but additionally for the regulation of the heating current for the lamp electrodes. The reference value for the lamp electrode heating is advantageously adjustable by means of a read/write memory, in order to be able to adapt the heating current for the lamp electrodes to the different operating states of the lamp. The microcontroller additionally advantageously has synchronization means for synchronizing the controllable switching means of the heating device with a switching means of the inverter. As a result, driving the switching means of the heating device is simplified. In addition, the oscillatory behavior of the inverter is influenced positively as a result.

In the ballast according to the invention, the DC supply circuit of the inverter advantageously has a step-up converter for power factor correction and/or to achieve a most sinusoidal mains current consumption, and the microcontroller is equipped with a second device for the alternate charging and discharging of a second charge store, and also with second control means for this second device for controlling the charging and/or discharging operations. The second device for the alternate charging and discharging of a charge store, the second charge store and the second control means for this second device form a second oscillator, which likewise operates independently of the operating cycle frequency of the microcontroller. The microcontroller is additionally equipped with second evaluation means, which are used to evaluate the oscillations of the charge state of the second charge store in order to produce pulse-width modulation control signals and/or frequency control signals for the controllable switching means of the step-up converter. In particular, the time periods required for recharging the second charge store between different charge states are evaluated for this purpose. The microcontroller therefore additionally performs the control of the step-up converter as well.

In order to evaluate the oscillations of the charge state of the second charge store to produce pulse-width modulation control signals and/or frequency control signals, the second evaluation means advantageously have a first comparator to compare the charge state of the second charge store with a first voltage value, and a second comparator to compare the charge state of the second charge store with a second, lower voltage value, and the second control means advantageously have switching means which are used to switch over the second device for the alternate charging and discharging of a charge store from charging to discharging of the second charge store when the first voltage value is reached, and to switch over the second device for the alternate charging and discharging of a charge store from discharging to charging of the second charge store when the second, lower voltage value is reached. The first or the second voltage value can advantageously be adjusted by means of a read/write memory. As a result, the first or second voltage value can be varied, for example by means of a program executed by the microcontroller, and can be stored in order to control the second device for the alternate charging and discharging of a charge store.

The two devices for the alternate charging and discharging of a charge store advantageously in each case have a controllable current source for applying an adjustable charging current to the charge store and the second charge store, and in each case a controllable current sink for applying an adjustable discharging current to the charge store and, respectively the second charge store. The controllable current sources and current sinks may be produced in a known manner with the aid of semiconductor technology and integrated into the microcontroller. As a result, the two devices for the alternate charging and discharging of a charge store can be produced with simple means as a constituent part of the microcontroller. In order to ensure fine graduation of the frequency control signals or the pulse-width modulation control signals, the settings of the controllable current sources and current sinks can be varied in relation to a reference current level, in each case with a resolution of at least 8 bits. The aforementioned reference current level for the charging current and the discharging current can advantageously be predefined by means of a nonreactive resistor. This makes it possible to adapt the control of the inverter to different mains voltages by means of appropriate dimensioning of the nonreactive resistor. In order to save components, it is additionally preferable for only a single nonreactive resistor to be used to predefine the same reference current level for the charging and discharging currents of the two charge stores.

The microcontroller of the ballast according to the invention advantageously has at least one status bit which can be set and reset and via which the at least one controllable switching means of the inverter can be activated and deactivated. With the aid of this status bit, the inverter can be switched off in a simple way in the event of a defective lamp or during end-of-life monitoring of the lamp. Instead, of course, it is also possible for the controllable switching means of the step-converter and therefore the voltage supply to the inverter to be deactivated by means of the status bit, in order in a simple way to implement safety shutdown of the ballast. The microcontroller advantageously has one or more further status bits which can be set and reset, in order to be able to switch the pulse-width modulation control of the step-up converter or of the inverter off or on as desired. As a result, it is possible to apply only frequency control signals or pulse-width modulation control signals or frequency signals and pulse-width modulation control signals as desired to the controllable switching means of the step-up converter and of the inverter.

The microcontroller of the ballast according to the invention is advantageously provided with interfaces for registering operating parameters of the step-up converter or of the inverter or of the at least one electric lamp, in order, by means of a program-controlled device belonging to the microcontroller, to evaluate the operating parameters and to generate actuating values for controlling the devices for the alternate charging and discharging of a charge store, or to determine the reference value for the lamp electrode heating or the first or second reference value for the control of the step-up converter. The microcontroller is preferably provided with interfaces for registering at least one operating parameter of the step-up converter, of the inverter and of the load circuit or the at least one electric lamp. As a result, control loops can be built up for the step-up converter, the inverter and the load circuit with the lamp.

The ballast according to the invention advantageously has terminals and means for communication with an externally arranged control device, which are in turn connected to interfaces of the microcontroller. As a result, the ballast according to the invention is prepared to receive and process control commands from an external control device and to emit status messages to the external control device. These processes are likewise monitored by the microcontroller of the ballast according to the invention.

According to the invention, the method according to the invention of operating at least one electric lamp on a ballast which has an inverter with a control circuit containing a microcontroller for the switching means of the inverter and at least one load circuit coupled to the inverter and having terminals for the at least one lamp, is distinguished by the fact that, with the aid of the microcontroller, a charge store has a charging current and a discharging current alternately applied to it, and the duration of the alternate charging and discharging operations of the charge store is evaluated and on this basis a frequency control signal and/or a pulse-width modulation control signal for the alternating control of the switching means of the inverter is generated. The method according to the invention makes it possible, irrespective of the operating cycle frequency of the microcontroller, to generate control signals for frequency control and/or for pulse-width modulation of the inverter, with the aid of the microcontroller. As a result, a comparatively cost-effective microcontroller, that is to say a microcontroller with a low operating cycle frequency, can be used in the ballast according to the invention in order to implement all the essential control functions.

In order to drive the switching means of the inverter alternately, use is advantageously made of a frequency divider or a pulse divider, which detects the changeover of the device for the alternate charging and discharging of a charge store from discharging to charging of the charge store or from charging to discharging of the charge store.

The method according to the invention also permits heating of the lamp electrodes, by the heating current for the lamp electrodes being regulated by means of a controllable switching means. The signals for the pulse-width modulated control of the controllable switching means of the heating device are advantageously generated with the aid of a comparator, which compares the charge state of the charge store with a reference value for the lamp electrode heating. In this way, frequency control signals and/or pulse-width modulation control signals can be generated both for the switching means of the inverter and for the controllable switching means of the heating device, by the duration of the charging and discharging operations of the charge store being evaluated. The reference value for the lamp electrode heating is advantageously set on the basis of the desired heating power and stored in a read/write memory of the microcontroller. As a result, the heating power can be set under program control by means of the microcontroller. In addition, the controllable switching means for regulating the heating current are advantageously switched on synchronously with a switching means of the inverter. This simplifies the driving of the controllable switching means of the heating device. The duty cycle of the controllable switching means for regulating the heating current is preferably smaller than or equal to the duty cycle of the corresponding switching means of the inverter.

The DC supply to the inverter is regulated with the aid of a step-up converter, in order to ensure power factor correction and/or a sinusoidal mains current consumption. The pulse-width modulation control signals and/or the frequency control signals for the controllable switching means of the step-up converter are likewise advantageously generated with the aid of the microcontroller, by a second charge store being recharged between different charge states, and the time periods for recharging the second charge store being evaluated in order to generate the pulse-width modulation control signals and/or the frequency control signals for the controllable switching means of the step-up converter. The same microcontroller as is used to control the inverter can in this way also be used to control the step-up converter. The recharging of the second charge store can be detected and evaluated in a simple way by means of two comparators, by the first comparator comparing the charge state of the second charge store with a first voltage value, and the second comparator comparing the charge state of the second charge store with a second, lower voltage value. When the first voltage value is reached, the charging operation is terminated and the discharging operation of the second charge store is started, while when the second, lower voltage value is reached, the discharging operation is terminated and the charging operation of the second charge store is restarted anew. The first or second voltage value is advantageously set by means of a read/write memory. As a result, the corresponding voltage value can be varied under program control.

Advantageously, with the aid of the microcontroller, actual values of operating parameters of the inverter and/or of the DC supply circuit of the inverter and/or of the at least one electric lamp are monitored and evaluated in order to control the charging or discharging operations of the charge store and/or to determine the reference value for the lamp electrode heating and/or to determine the first and/or second voltage value. As a result, control loops for controlling the inverter and its DC supply and also for the lamp electrode heating can be implemented.

IV. BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a schematic illustration of the first half of the circuit arrangement according to the preferred exemplary embodiment of the ballast according to the invention.

FIG. 2 shows a schematic illustration of the second half of the circuit arrangement according to the preferred exemplary embodiment of the ballast according to the invention.

FIG. 3 shows a block diagram of the microcontroller.

FIG. 4 shows a block diagram of the second control module G for controlling the half-bridge inverter and the heating device.

FIG. 5 shows a graph of the control signals for the inverter and the heating device.

FIG. 6 shows a block diagram of the first control module E for controlling the step-up converter.

FIG. 7 shows a graph of the control signals for the step-up converter.

V. DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS

The circuit arrangement of the preferred exemplary embodiment of the ballast according to the invention is illustrated schematically in FIGS. 1 and 2 . Because of its size, the circuit arrangement has had to be illustrated on two sheets. The two halves of the circuit arrangement, depicted in FIGS. 1 and 2 , are linked to each other at the connecting points designated by J 10 to J 26 . This ballast is an electronic ballast, as it is known, for operating fluorescent lamps. The ballast has two mains voltage terminals J 1 , J 2 , to which a filter circuit, comprising the capacitor C 1 and the transformer L 1 , is connected to suppress radio interference from the ballast. This filter circuit is connected to a bridge rectifier, which is formed by four rectifier diodes D 1 , D 2 , D 3 and D 4 . Connected downstream of the bridge rectifier D 1 -D 4 is the capacitor C 2 , which forms the DC output of the bridge rectifier D 1 -D 4 . Connected to the capacitor C 2 is a step-up converter, which comprises the field effect transistor V 1 , the inductor L 2 , the diode D 5 and the resistor R 13 . The DC voltage present across the capacitor C 2 is used as a supply voltage for the step-up converter. The gate electrode of the transistor V 1 is connected via the resistor R 4 to the pin 4 of the microcontroller MC, which performs the control of the transistor V 1 . The voltage output from the step-up converter is formed by the intermediate circuit capacitor C 3 . The voltage across the intermediate circuit capacitor C 3 is monitored by means of the voltage divider resistors R 2 , R 5 on pin 21 of the microcontroller MC. In addition, in order to control the transistor V 1 , the voltage across the capacitor C 2 is also detected with the aid of the voltage divider resistors R 1 , R 18 on pin 20 of the microcontroller MC.

Across the intermediate circuit capacitor C 3 , a smoothed DC voltage is provided in order to supply the half-bridge inverter connected downstream. The half-bridge converter substantially comprises the field effect transistors V 2 , V 3 , the snubber capacitors C 10 , C 11 , the inductor L 4 , the coupling capacitors C 15 , C 16 and the firing capacitor C 12 . Connected to the center tap between the two transistors V 2 , V 3 of the inverter is a load circuit, which comprises the inductor L 4 , the firing capacitor C 12 , the terminals X 1 to X 8 for the electrode filaments E 1 , E 2 and E 3 , E 4 of the two parallel connected fluorescent lamps LP 1 , LP 2 , the transformer L 5 and the coupling capacitors C 15 , C 16 . The firing capacitor C 12 is connected in parallel with the two lamps LP 1 , LP 2 . The coupling capacitors C 15 , C 16 are in each case arranged in series with one of the lamps LP 1 , LP 2 . The transformer L 5 is used to balance the currents in the lamp circuits. For this purpose, in each case one of the transformer windings is arranged in one of the lamp circuits, that is to say in series with one of the lamps LP 1 , LP 2 . The two lamp circuits are combined again at the terminal X 8 and at the two terminals of the coupling capacitors C 15 , C 16 connected to the internal circuit ground GRD. The gate electrodes of the transistors V 2 , V 3 are controlled by the microcontroller MC, via the resistors R 6 and R 7 , with the aid of the integrated circuit IC, which substantially only has driver circuits for driving the inverter transistors and circuits for generating auxiliary voltages for the microcontroller MC. In the load circuit for the lamps LP 1 , LP 2 , the half-bridge inverter generates a high-frequency current at a frequency between about 30 kHz and 100 kHz. After the gas discharge in the lamps LP 1 , LP 2 has been fired, high-frequency lamp currents flow in the two lamp circuits, via the terminal X 8 , the discharge path of the lamp LP 1 and LP 2 , the terminal X 5 and X 7 and via the coupling capacitors C 16 and C 15 . The inductor L 4 and the firing capacitor C 12 are designed as a series resonant circuit. The firing voltage required to fire the gas discharge in the fluorescent lamps is provided by the resonant peak method on the firing capacitor C 12 , by the switching frequency of the transistors V 2 , V 3 of the half-bridge inverter being brought close to the resonant frequency of the series resonant circuit during the firing phase. The center tap between the inductor L 4 and the firing capacitor C 12 is connected to pin 18 of the microcontroller MC via the capacitor C 22 , the resistor R 24 and the diode D 12 polarized in the forward direction. A half wave belonging to the alternating current component of the load current is monitored on pin 18 by means of the resistors R 24 , R 25 , the diodes D 12 , D 13 and the capacitors C 22 , C 23 . The other half wave of the alternating current component of the current flowing in the load circuit is clamped to the internal circuit ground potential GRD by the diode D 13 . Pin 19 of the microcontroller MC is connected via the resistor R 27 to the source electrode of the transistor V 3 and, via the capacitor C 24 , is coupled to the internal circuit ground potential GRD. The resistor R 9 connects the source electrode of the transistor V 3 to the internal circuit ground potential GRD. The current through the transistor V 3 is monitored at pin 19 .

The ballast also has a heating device for the electrodes E 1 -E 4 of the two fluorescent lamps, which is connected to the center tap between the two field effect transistors V 2 , V 3 of the half-bridge inverter. This heating device substantially comprises the field effect transistor V 4 and the transformer L 3 . The primary winding of the transformer L 3 is connected on one side to the center tap between the transistors V 2 , V 3 and on the other side to the drain terminal of the transistor V 4 and also, in the DC forward direction, via the diode D 8 to the positive pole of the intermediate circuit capacitor C 3 . The source electrode of the transistor V 4 is connected via the resistor R 17 to the internal circuit ground potential GRD. The three secondary windings of the transformer L 3 , when lamps LP 1 , LP 2 are connected, are in each case arranged together with a rectifier diode D 9 or D 10 or D 11 in a closed circuit for heating the electrode filaments E 1 and E 3 and the electrode filament E 2 or E 4 . The heating current in the three heating circuits fitted with the secondary windings of the transformer L 3 is regulated by the switching cycle of the transistor V 4 . In order to control the switching cycle of the transistor V 4 , its gate electrode is connected via the resistor R 26 to pin 10 of the microcontroller MC. The heating device is used firstly to preheat the electrode filaments E 1 -E 4 before the gas discharge in the lamps LP 1 , LP 2 is fired, and secondly for heating the electrode filaments E 1 -E 4 during dimming operation of the lamps LP 1 , LP 2 . The heating current, that is to say the current through the primary winding of the transformer L 3 and through the transistor V 4 , is monitored with the aid of the RC element R 23 , C 18 on pin 17 of the microcontroller MC. For this purpose, pin 17 is connected to the source electrode of the transistor V 4 via the resistor R 23 .

With the aid of the resistor R 10 and the diode D 9 , a direct current path is implemented which, starting from the positive pole of the capacitor C 3 , is led via the resistor R 10 , the terminal X 3 , the electrode filament E 1 , the terminal X 8 , the electrode filament E 3 , the terminal X 2 and via the resistors R 14 , R 22 to the internal circuit ground potential GRD. This direct current path is interrupted if one of the lamps LP 1 or LP 2 is missing or one of the electrode filaments E 1 or E 3 is defective. The center tap between the resistors R 14 , R 22 is connected to pin 25 of the microcontroller MC, in order to monitor the direct current path. Two further direct current paths are implemented with the aid of the resistor R 11 and R 12 and the diodes D 10 and D 11 and also the resistors R 16 , R 20 and R 15 , R 21 , in order to monitor the electrode filaments E 2 and E 4 . A rupture of the electrode filament E 2 or E 4 is detected at pin 16 or 15 by the microcontroller MC via the corresponding winding of the transformer L 5 and via a resistor R 16 or R 15 . In addition, by means of the voltage divider resistors R 15 , R 21 and R 16 , R 20 , the current through the lamp LP 1 and LP 2 or the voltage drop across the coupling capacitor C 15 or C 16 is also monitored on pin 15 , 16 of the microcontroller MC, in order to detect the rectifying effect of the lamp LP 1 or LP 2 that occurs at the end of the lifetime of the lamp LP 1 or LP 2 .

The ballast additionally has a communication device DS for communication with an external control device (not depicted). This device DS has two terminals J 3 , J 4 , which can be connected to the external control device. The terminals J 3 , J 4 are used to receive digital or analog control signals from the external control device and to transmit information, for example about the operating state of the lamps, from the ballast to the external control device. A bidirectional connection to the external control device is possible via the terminals J 3 , J 4 . One output of the communication device DS is connected to the internal circuit ground potential GRD. Pin 6 of the microcontroller MC is connected to the input to the communication device DS in order to transmit data to the external control unit, and pin 5 of the microcontroller MC is connected to the output from the communication device DS in order to receive and to evaluate control commands from the external control device.

The integrated circuit IC contains driver circuits for the transistors V 2 , V 3 , in particular a bootstrap circuit for the transistor V 2 and level-shift circuits for controlling the transistors V 2 , V 3 . The capacitor C 9 and the pins 1 , 2 , 3 and 14 of the integrated circuit IC are assigned to these driver circuits of the transistors V 2 , V 3 . The control signals for regulating the switching cycle of the transistors V 2 , V 3 and for the frequency control of the half-bridge inverter are generated by the microcontroller MC and supplied to pin 9 and 10 of the integrated circuit IC via pin 24 and 23 , respectively. With the aid of the resistor R 8 , which connects pin 13 of the integrated circuit IC to the source terminal of the transistor V 3 , and the capacitor C 8 , via which pin 13 of the integrated circuit IC is coupled to the ground potential GRD, a detector is implemented which prevents excessively high current loading of the transistors V 2 , V 3 . Via the resistor R 3 , pin 5 of the integrated circuit IC is connected to the positive pole of the capacitor C 2 . During the starting phase, that is to say before the half-bridge inverter has started its oscillation, a voltage supply for the integrated circuit IC is ensured via pin 5 . On pins 8 and 11 of the integrated circuit IC, with the aid of the capacitors C 14 and C 25 , auxiliary voltages of 5 V and 15 V for the microcontroller MC are provided. As long as the half-bridge inverter oscillates, the voltage for supplying the integrated circuit IC and the microcontroller MC is derived from the load circuit by means of the capacitor C 13 connected to pin 7 of the integrated circuit IC and to the center tap between the firing capacitor C 12 and the inductor L 4 , and by means of a two-point regulator integrated in the integrated circuit IC.

In the following text, the construction of the microcontroller MC and the generation of the control signals for the transistors V 1 -V 4 with the aid of the microcontroller MC will be explained in more detail.

The construction of the microcontroller MC is shown schematically in FIG. 3 . The microcontroller MC has a clock generator, which determines the operating cycle of the microcontroller, a central processor unit, a program memory, a data memory and a mathematic unit for carrying out simple mathematical operations. The aforementioned parts of the microcontroller MC are represented by the module A in the block diagram of FIG. 3 . Pins 1 and 2 , 15 to 22 and 23 to 28 are associated with the module A. The quartz crystal B 2 for controlling the clock generator is connected to pins 1 to 2 . The operating clock frequency of the microcontroller is 8 MHz. The module B is an interface, which is used to condition the digital or analog data for the communication with the communication device DS. Pins 5 and 6 of the microcontroller MC are assigned to the module B. Module C is a 5 V voltage supply, which is connected to the capacitor C 14 via pins 11 and 12 of the microcontroller MC and to the ground potential GRD. All the components of the microcontroller MC are connected to one another by the address and data bus D. The first control module E and the pins 3 , 4 and 9 of the microcontroller MC which are assigned co it are used to control the transistor V 1 of the step-up converter. The second control module G and the pins 7 , 8 and 10 of the microcontroller MC which are assigned to it are used to control the transistors V 2 and V 3 of the half-bridge inverter and to control the transistor V 4 of the heating device. The two control modules E, G are connected to each other via the data bus F. Module H is a 15 V voltage source, which is connected to the ground potential GRD and to the capacitor C 25 via pins 13 , 14 of the microcontroller MC.

The construction of the control module G is shown schematically in the block diagram of FIG. 4 . In order to control the transistors V 2 , V 3 of the half-bridge inverter, the control module G has the controllable current source SQ 1 , the controllable current sink SS 1 , the read/write memory DR 1 , DR 2 , the switch US 1 for switching the controllable current source and current sink on and off, the frequency divider FT 1 for halving the frequency of the changeover signal of the switch US 1 , the data memory DR 3 for storing the control signals for the transistors V 2 , V 3 , the reference current source IR for predefining the most constant possible reference current I Ref for the controllable current source SQ 1 and current sink SS 1 , and logic circuit components O 1 -O 3 , U 1 -U 6 .

A constant output voltage of 2 V is provided on pin 7 of the microcontroller MC and, in accordance with Ohm's law, permits a constant reference current I Ref to flow through the resistor R 30 . The value of this reference current I Ref can be predefined by selecting the resistance value of the resistor R 30 . The linear working range of the reference current I Ref extends from 5 μA to 50 μA. The capacitor C 27 , which is used as an electric charge store, is connected to pin 8 of the microcontroller MC. The capacitor C 27 is charged up with the aid of the controllable current source SQ 1 . Once the voltage drop across the capacitor C 27 reaches a value of 3 V, the controllable current source SQ 1 is switched off by the switch US 1 and the controllable current sink is switched on, which discharges the capacitor C 27 . Once the voltage drop across the capacitor C 27 reaches the value of 1.5 V, the controllable current sink SS 1 is switched off by the switch US 1 and the controllable current source SQ 1 is switched on again, which charges the capacitor up again to a voltage value of 3 V. In this way, the capacitor C 27 is alternately charged up and discharged. The voltage drop across the capacitor C 27 therefore oscillates incessantly between the values 1.5 V and 3 V. The controllable current source SQ 1 and the controllable current sink SS 1 and also the switch US 1 form a device for the alternate charging and discharging of the capacitor C 27 . The charging current for the capacitor C 27 , generated by the controllable current source SQ 1 , can be adjusted by means of the read/write memory DR 1 . The read/write memory DR 1 is a 16-bit data register, of which 12 bits are used to control the current source SQ 1 . The charging current for the capacitor C 27 can therefore be adjusted with a resolution of 12 bits between the values I Ref /256 and 32 I Ref , the abbreviation I Ref representing the reference current intensity of the reference current source IR. The entry in the data register DR 1 determines the charging current for the actual and following charging operation on the capacitor C 27 , and therefore the time period which is required for this charging operation. In an analogous way, the discharging current of the capacitor C 27 , generated by the controllable current sink SS 1 , can be adjusted by means of the read/write memory DR 2 . The read/write memory DR 2 is an 8-bit data register. The discharging current of the capacitor C 27 can therefore be adjusted with a resolution of 8 bits between the values 0.25 I Ref and 128 I Ref . The entry in the data register DR 2 determines the discharging current for the actual and following discharging operation on the capacitor C 27 , and therefore the time period which is required for this discharging operation. The oscillations in the charge state of the capacitor C 27 and the voltage drop across the capacitor C 27 are therefore independent of the operating cycle frequency of the microcontroller MC. The changeover signals from the switch US 1 are evaluated by the frequency divider FT 1 and the AND gates U 1 , U 2 in order to generate control signals for the transistors V 2 , V 3 of the half-bridge inverter. The frequency divider FT 1 detects only the switching pulses from the switch US 1 which start a new charging operation of the capacitor C 27 , and alternately switches its two outputs, which are respectively connected to the input of an AND gate U 1 and U 2 , alternately to “high” and “low” at each such switching pulse. The changeover signals from the switch US 1 are also supplied, however, directly to the input of the AND gates U 1 , U 2 . In addition, the status register SR 1 comprises a status bit to activate and deactivate the control signals for the transistor V 2 , and also a status bit to activate and deactivate the control signals for the transistor V 3 . The state of the status bit to activate and deactivate the control signals for the transistor V 2 is monitored by the AND gate U 2 , while the state of U 1 . The output states of the AND gate U 1 and U 2 are in each case stored in one bit in the data register DR 3 and can be called up via the address and data bus D on the pins 23 and 24 of the microcontroller MC. Via pin 23 and 24 of the microcontroller MC, which is connected to pin 10 and 9 of the integrated circuit IC, the output states of the AND gate U 1 and U 2 are communicated to the driver circuits for driving the gate electrode of the transistor V 3 and V 2 . The frequency of the half-bridge inverter, that is to say the switching cycle of its transistors V 2 , V 3 , is controlled by the duration of the individual charging and discharging operations of the capacitor C 27 . This fact is to be explained in more detail below by using the graphs a) to e) of FIG. 5 .

The triangular curve of graph a) shows the time variation of the voltage drop across the capacitor C 27 . The voltage drop across the capacitor C 27 varies linearly with time between the values 1.5 V and 3 V. Graph b) shows the time variation of the charging current for the capacitor C 27 . The charging current can assume 4096 different discrete values, according to the above explanations relating to the controllable current source SQ 1 . The time variation of the discharging current for the capacitor C 27 is shown in graph c). According to the above explanations relating to the controllable current sink SS 1 , the discharging current can assume 256 different discrete values. Graph d) shows the time variation of the control signal LG, which can be called up on pin 23 of the microcontroller MC, for the driver circuit of the transistor V 3 . Graph e) shows the time variation of the control signal HG, which can be called up on pin 24 of the microcontroller MC, for the driver circuit of the transistor V 2 . The duration of the individual charging operations on the capacitor C 27 is determined by the level of the charging current IL 1 . The higher the charging current IL 1 , the lower is the time period which is required for charging the capacitor from 1.5 V to 3 V. Analogously to this, the duration of the individual discharging operations on the capacitor C 27 is determined by the level of the discharging current IE 1 . The higher the discharging current IE 1 , the lower the time period which is required for discharging the capacitor from 3 V to 1.5 V. By comparing the voltage variation across the capacitor C 27 from graph a) with the curves from graphs d) and e), it becomes clear that during the duration of the first, third, fifth, etc. charging operation of the capacitor C 27 from 1.5 V to 3 V, the control signal LG for the transistor V 3 assumes the logic state “high” and the control signal HG for the transistor V 2 carries the logic state “low”. During the duration of the second, fourth, sixth, etc. charging operation of the capacitor C 27 from 1.5 to 3 V, on the other hand, the control signal HG for the transistor V 2 assumes the logic state “high” and the control signal LG for the transistor V 3 carries the logic state “low”. During the duration of the discharging operations of the capacitor C 27 from 3 V to 1.5 V, both control signals LG and HG assume the logic state “low”. This means that the transistor V 2 or V 3 is switched on as long as the control signal HG or LG associated with it carries the state “high”. In this way, the transistors V 2 , V 3 of the half-bridge inverter are switched on and off alternately. During the duration of the discharging operations of the capacitor C 27 , both transistors V 2 , V 3 are switched off. The evaluation of the voltage variation across the capacitor C 27 in this way permits frequency-modulated control of the half-bridge inverter.

The values for the charging current IL 1 and the discharging current IE 1 are defined by the data stored in the data register DR 1 and DR 2 . These data are determined under program control, with the aid of the module A, on the basis of the half wave of the alternating current component of the current in the load circuit, detected at pin 18 of the microcontroller MC, and on the current through the transistor V 3 , detected at pin 19 . Module A of the microcontroller MC uses the comparison between the aforementioned operating parameters and predefined set points to calculate, under program control, actuating values for controlling the controllable current source SQ 1 and the controllable current sink SS 1 , said set points being stored in the data registers DR 1 and DR 2 . In this way, a control loop is implemented for the frequency-modulated control of the half-bridge inverter on the basis of its operating parameters and the predefined set points. The set points for the frequency-modulated control of the half-bridge inverter are determined under program control by module A of the microcontroller MC, for example on the basis of external control commands to dim the lamps LP 1 , LP 2 , which are communicated via the interfaces J 3 , J 4 of the communication device DS and supplied to pin 5 of the microcontroller MC. The data registers DR 1 to DR 4 and the status register SR 1 are connected to the address and data bus D.

The voltage variation across the capacitor C 27 , illustrated in graph a) of FIG. 5 , is additionally also evaluated in order to generate pulse-width modulated control signals for the transistor V 4 of the heating device for the electrode filaments E 1 -E 4 of the lamps LP 1 , LP 2 . For this purpose, use is made of the read/write memory DR 4 , designed as an 8-bit data register, the comparator K 1 , whose inverting input detects the voltage drop across the capacitor C 27 and whose non-inverting input is controlled by the data register DR 4 , the status register SR 1 and the logic circuit components O 1 , O 2 , U 3 , U 4 , U 5 , O 3 and the driver circuit TR 1 for the transistor V 4 . The comparator K 1 compares the voltage variation on the capacitor C 27 with the actuating value stored in the data register DR 4 for the regulation of the heating current. The aforementioned actuating value can be varied with a resolution of 8 bits. Accordingly, the voltage on the non-inverting input of the comparator K 1 can also be varied with the same resolution in the range from 1.5 V to 3 V. The output signal from the comparator K 1 is supplied via the OR gate O 1 and the AND gate U 3 to the OR gate O 3 , whose output is connected to the input of the driver circuit TR 1 , which drives the gate electrode of the transistor V 4 via pin 10 of the microcontroller MC and the resistor R 26 . The output signal from the comparator K 1 is additionally also supplied to the OR gate O 2 , whose output is connected to the AND gates U 1 and U 2 . The output of the AND gate U 1 is connected via the AND gate U 3 to the OR gate O 3 . The output of the AND gate U 2 is connected via the AND gate U 4 to the OR gate O 3 . The 8-bit status register SR 1 has a first status bit to activate and deactivate a maximum heating current, said bit being connected via the OR gate O 1 and the AND gate U 3 to the OR gate O 3 . The term maximum heating current means that the duty cycle of the transistor V 4 is equal to the duty cycle of the transistor V 2 or V 3 . The second status bit of the status register SR 1 , which is connected via the AND gate U 3 to the OR gate O 3 , is used to activate and deactivate the synchronous switching-on of the transistors V 3 and V 4 . The third status bit of the status register SR 1 , which is connected via the AND gate U 4 to the OR gate O 3 , is used to activate and deactivate the synchronous switching-on of the transistors V 2 and V 4 . The fourth status bit of the status register SR 1 is connected to the AND gate U 6 , whose output is connected via the data bus F to the control module E. Since the output of the AND gate U 1 is connected to the AND gate U 6 , the connection between the control signal LG and the control module E is activated and deactivated via the fourth status bit. The fifth status bit of the status register SR 1 is connected via the AND gate U 5 to the OR gate O 3 . The AND gate U 5 additionally receives an input signal from the control module E via the data bus F. By means of the fifth status bit, the synchronization of the control signals for the transistors V 1 and V 4 can be activated and deactivated. The sixth status bit of the status register SR 1 , which is connected to the OR gate O 2 , is used to activate and deactivate the pulse-width modulation of the control signals LG and HG. The seventh and eighth status bit, which is connected to the AND gate U 1 and U 2 , is used to activate and deactivate the control signals LG and HG for the transistors V 3 and V 2 , and also for the transistor V 4 .

By means of the seventh or eighth status bit, the half-bridge inverter and the heating device can be switched off in a simple way in the event of defective lamps LP 1 , LP 2 . As has already been mentioned above, a direct current path is implemented by means of the resistor R 10 , the diode D 9 and the appropriate secondary winding of the transformer L 3 , into which path the electrode filaments E 1 and E 3 are connected in series. If one of the lamps LP 1 , LP 2 is missing, then this direct current path is interrupted. The current in this direct current path is monitored at pin 25 of the microcontroller MC via the resistor R 14 . If the abovementioned direct current path has been interrupted, then the control signal LG and HG can be switched off by resetting the seventh or eighth status bit of the status register SR 1 , and the half-bridge inverter can be stopped as a result.

As has already been mentioned above, rupture of the electrode filaments E 2 and E 4 is detected at pin 16 and 15 of the microcontroller MC via the appropriate winding of the transformer L 5 and the resistor R 16 or R 15 . In addition, the current through the lamp LP 1 and LP 2 or the voltage drop across the coupling capacitor C 15 and C 16 is monitored at pins 15 and 16 of the microcontroller MC by means of the voltage divider resistors R 15 , R 21 and R 16 , R 20 , in order to detect the rectifying effect of the lamp LP 1 or LP 2 that occurs at the end of the lifetime of the lamp LP 1 or LP 2 . The information is evaluated by the microcontroller MC and can be transmitted to an external control device via pin 6 and the communication device DS, or used to control the transistors V 2 , V 3 and V 4 .

The generation of pulse-width modulated control signals for the gate electrode of the transistor V 4 will be explained below using graphs a) and f) from FIG. 5 . In addition to the triangular time variation of the voltage across the capacitor C 27 , graph a) of FIG. 5 also illustrates a staircase function, decreasing in three steps over time, which represents the actuating value for controlling the heating current that is stored in the 8-bit data register DR 4 . This actuating value is supplied to the non-inverting input of the comparator K 1 . Since, in the present exemplary embodiment, the third status bit of the status register SR 1 is set, the control signals HTG and HG for the transistors V 4 and V 2 change simultaneously from the “low” state to the “high” state. This means that the transistor V 4 is always switched on synchronously with the transistor V 2 of the half-bridge inverter. The duty cycle or the turn-off time of the transistor T 4 , and therefore also the pulse width of the control signal HTG, depend on the output signal of the comparator K 1 , which compares the actuating value stored in the data register DR 4 for regulating the heating current with the instantaneous voltage drop across the capacitor C 27 . If, during the charging operations of the capacitor C 27 , during which the control signal HTG is in the “high” state, the voltage across the capacitor C 27 reaches the value stored in the data register DR 4 , then the control signal HTG changes from the “high” state to the “low” state. Since the signal present on the non-inverting input of the comparator K 1 can assume only values between 1.5 V and 3 V, the pulse width of the control signal HTG is less than or equal to the pulse width of the control signal HG. This means that the duty cycle of the transistor V 4 is at most exactly as long as the duty cycle of the transistor V 2 . In this case, the greatest possible heating current flows through the electrode filaments E 1 -E 4 . In order to build up a control loop for the heating current, the current through the transistor T 4 and through the primary winding of the transformer L 3 is monitored at pin 17 via the RC element R 23 , C 18 and, under program control, is compared with a set point by means of the module A and, on the basis of the comparison, an actuating value for generating the control signal HTG is stored in the data register DR 4 . The requisite heating current depends on the operating state of the lamps LP 1 , LP 2 . During the preheating phase, a relatively high heating current is needed in order to permit gentle firing of the gas discharge. In addition, a heating current for the electrode filaments is also needed in the case of highly dimmed lamps LP 1 , LP 2 .

FIG. 6 illustrates schematically the construction of the control module E for controlling the transistor V 1 of the step-up converter, which is used to supply DC to the half-bridge inverter connected downstream. The control module E has the controllable current source SQ 2 , the controllable current sink SS 2 , the read/write memories DR 5 , DR 6 , DR 7 , the status registers SR 1 , SR 2 , SR 3 , the comparators K 2 , K 3 , K 4 , K 5 and the driver circuit TR 2 for the transistor V 1 . The aforementioned components of the control module E are linked to one another by logic circuit components. The status register SR 1 is the same status register which has already been described in connection with the control module G. The controllable current source SQ 2 is used to charge the capacitor C 26 connected to pin 9 of the microcontroller MC, and the controllable current sink SS 2 is used to discharge the capacitor C 26 . The controllable current source SQ 2 and the controllable current sink SS 2 are each coupled to the reference current source IR. The charging current and the discharging current for the capacitor C 26 are each adjustable with a resolution of 8 bits between the values 0.25 I Ref and 128 I Ref . For this purpose, use is made of the read/write memories DR 5 and DR 6 each designed as 8-bit data registers. The charging current is adjusted by means of the data register DR 6 and the discharging current is adjusted by means of DR 5 .

With the aid of the controllable current source SQ 2 , the capacitor C 26 connected to pin 9 of the microcontroller MC is charged up to a predefinable upper voltage value, which lies in the range from 1.5 V to 3 V. When the upper voltage value is reached, the charging operation is broken off and the discharging operation of the capacitor C 26 with the aid of the controllable current sink SS 2 is started. When the voltage across the capacitor reaches the lower voltage value of 1.5 V, the discharging operation is broken off and a new charging operation on the capacitor C 26 is started. The activation and deactivation of the controllable current source SQ 2 and of the controllable current sink SS 2 for the alternate charging and discharging of the capacitor C 26 is carried out with the aid of the RS flip-flop FL 1 and by means of the comparators K 2 and K 4 or, alternatively, by means of the comparators K 3 and K 4 . The comparator K 2 compares the voltage across the capacitor C 26 with the upper voltage value, while the comparator K 4 compares the voltage across the capacitor C 26 with the lower voltage value of 1.5 V. The upper voltage value can be adjusted by means of the 8-bit data register DR 7 , which is connected to the inverting input of the comparator K 2 . Instead of the comparator K 2 , however, the comparator K 3 can also be selected, in order to compare the voltage across the capacitor C 26 with the upper voltage value. However, when the comparator K 3 is used, the upper voltage value is 3 V and cannot be varied. In order to control the controllable current source SQ 2 and the controllable current sink SS 2 for the alternating charging and discharging operations on the capacitor C 26 , the outputs of the comparators K 2 and K 3 are connected to the set input of the RS flip-flop FL 1 via the positive flank generator SG 1 , the AND gate U 7 and the OR gate O 4 or via the positive flank generator FG 2 , the AND gate U 8 and the OR gate O 4 . The output of the comparator K 4 is connected to the reset input of the RS flip-flop FL 1 via the positive flank generator FG 3 . The two outputs of the RS flip-flop FL 1 are connected to the controllable current source SQ 2 and to the controllable current sink SS 2 . The controllable current source SQ 2 , the controllable current sink SS 2 , the comparators K 2 (or K 3 ) and K 2 and also the RS flip-flop FL 1 form a device for the alternate charging and discharging of a charge store, which alternately applies a charging current and a discharging current to the capacitor C 26 . The voltage across the capacitor C 26 therefore oscillates incessantly between the upper and lower voltage values. This oscillation is independent of the operating cycle frequency of the microcontroller MC. The time periods which are required for charging and discharging the capacitor C 26 between the upper and the lower voltage values are used, by means of the comparators K 2 (or K 3 ), K 4 , the positive flank generators FG 1 -FG 3 , the RS flip-flop FL 2 and the logical circuit components U 9 -U 11 , O 5 , O 6 , to generate a frequency modulated and pulse-width modulated control signal PG for the input to the driver circuit TR 2 , which is supplied to the gate electrode of the transistor V 1 via pin 4 of the microcontroller MC and the resistor R 4 . In addition, the control module E also comprises the comparator K 5 , the RS flip-flop FL 3 , FL 4 , the OR gate O 7 and the status registers SR 2 , SR 3 . The status registers SR 1 -SR 3 and the data registers DR 5 -DR 7 are connected to the address and data bus D. With the aid of the RC element R 32 , C 28 , the current through the transistor V 1 is monitored at pin 3 of the microcontroller MC. By means of the comparator K 5 , the OR gate O 7 and the RS flip-flop FL 4 , the transistor V 1 is protected against excessively high currents, by the control signal PG for the transistor V 1 switching off if an excessively high current occurs. For this purpose, pin 3 of the microcontroller MC is connected to the non-inverting input of the comparator K 5 , while on the inverting input of the comparator K 5 there is a reference value which, by means of the status register SR 3 , can be adjusted with a resolution of four bits between the values 0 V and 2 V and which defines the turn-off threshold for the control signal PG. In the event that the control signal PG is switched off by the comparator K 5 and the RS flip-flop FL 4 , the first status bit in the status register SR 2 is set by means of the RS flip-flop FL 3 . The second status bit in the status register SR 2 is set and reset on the basis of the output signal from the OR gate O 6 and indicates whether a control signal PG is present or not. The remaining six bits in the status register SR 2 are unused. Of the status register SR 3 , the first four bits are used to drive the inverting input of the comparator K 5 . The fifth bit of the status register SR 3 permits additional control of the reference current source IR. The sixth bit of status register SR 3 is unused. With the aid of the seventh bit of the status register SR 3 and of the AND gate U 9 , the control signal for the driver circuit TR 2 and the transistor V 1 can be activated and deactivated. With the aid of the eighth bit of the status register SR 3 and of the AND gate U 7 , U 8 , the output signal from the comparator K 2 or the comparator K 3 can be activated as desired. As a result, two different operating modes for the step-up converter are made possible. If the output signal from the comparator K 2 is active, the step-up converter regulates not only the supply voltage of the half-bridge inverter but is also used for power factor correction. This operating mode is preferred for the operation of discharge lamps, in particular fluorescent lamps. The other operating mode of the step-up converter is suitable for the operation of low-voltage incandescent halogen lamps on an electronic transformer which has a step-up converter to regulate the supply voltage of the inverter connected downstream. In the case of the present exemplary embodiment, the output signal from the comparator K 2 is active. The control signal PG can also be made available on pin 10 of the microcontroller MC, via the AND gate U 12 , the data bus F and the AND gate U 5 , by means of the fifth status bit in the status register SR 1 , in order to control the transistor V 4 . On the other hand, the control signal LG of the control module G for controlling the transistor V 3 can also be made available on pin 4 of the microcontroller MC, via the AND gate U 6 , the data bus F and the OR gate O 7 , by means of the fourth status bit in the status register SR 1 , in order to control the transistor V 1 .

The generation of the control signal PG for the transistor V 1 will be explained in more detail below by using FIG. 7 . The triangular curve in graph a) of FIG. 7 represents the variation over time of the voltage across the capacitor C 26 . The step-like curve in graph a) of FIG. 7 represents the time variation of the memory content of the data register DR 7 , which can assume values between 1.5 V and 3 V with a resolution of eight bits. Graph b) illustrates the variation over time of the control signal PG for the gate electrode of the transistor V 1 , which can be called up on pin 4 of the microcontroller MC. Graph c) of FIG. 7 shows the time variation of the signal generated on pin 3 of the microcontroller MC by means of the RC element R 32 , C 28 in order to monitor the current through the transistor V 1 . Graph d) shows the time variation of the charging current for the capacitor C 26 , generated by the controllable current source SQ 2 , and graph e) of FIG. 7 shows the time variation of the discharging current for the capacitor C 26 , generated by the controllable current sink SS 2 . The capacitor C 26 is alternately charged up to an upper voltage value, which is determined by the memory content of the data register DR 7 , and discharged down to a lower voltage value of 1.5 V. The duration of the individual charging operations of the capacitor C 26 is therefore defined by the upper voltage value and by the charging current IL 2 , which can be adjusted by means of the data register DR 6 . In a corresponding way, the duration of the individual discharging operations is determined by the upper voltage value and the discharging current IE 2 , which can be adjusted by means of the data register DR 5 . The time periods which are required for the alternate charging and discharging of the capacitor C 26 are evaluated, by means of the above-described logical circuit components of the control module E, in order to generate the frequency modulated and pulse-width modulated control signal PG. The comparison of the voltage variation across the capacitor C 26 , illustrated in graph a), with the control signal PG depicted in graph b) shows that the transistor V 1 is switched off during the charging operations on the capacitor C 26 and is switched on during the discharging operations on the capacitor C 26 . Once the signal IV 1 (graph c) of FIG. 7 ), detected on pin 3 of the microcontroller, reaches the threshold set at the inverting input of the comparator K 5 , the control signal PG is deactivated.

As has already been described above, the voltage across the capacitor C 2 is monitored on pin 20 of the microcontroller MC, and the voltage across the capacitor C 3 is monitored on pin 21 of the microcontroller MC. From these values, by means of the module A of the microcontroller MC, the current through the step-up converter inductor L 2 can be calculated and, on the basis of these operating parameters, with the aid of the program implemented in module A, the memory contents of the data registers DR 5 , DR 6 and DR 7 can be determined in order to generate the control signal PG for the transistor V 1 . In this way, a control loop for controlling the transistor V 1 is implemented.

The invention is not restricted to the exemplary embodiments described in detail above. For example, the invention can also be used to control the switching transistors of ballasts for the operation of high-pressure discharge lamps, and also of electronic transformers for the operation of low-voltage incandescent halogen lamps. In particular, it is also possible, by means of the device according to the invention for the alternate charging and discharging of a charge store, designed as a constituent part of a microcontroller, to generate the frequency modulated or pulse-width modulated control signals for the switching transistors of a full-bridge inverter or of a push-pull inverter.

Claims

1. A microcontroller having at least one device (E, G) for controlling a switched-mode power supply, characterized in that the at least one device (E, G) hasa device (SQ1, SS1; SQ2, SS2) for the alternate charging and discharging of a charge store (C27; C26) that can be connected to the microcontroller (MC) or integrated into the microcontroller (MC), wherein the device (SQ1, SS1; SQ2, SS2) has: (i) a controllable current source (SQ1; SQ2) for applying an adjustable charging current to the charge storage capacitance (C27; C26); and (ii) a controllable current sink (SS1; SS2) for applying an adjustable discharging current to the charge storage capacitance (C27; C26), control means for the device (SQ1, SS1; SQ2, SS2) for controlling the charging operations and the discharging operations, and evaluation means which are used to evaluate the time periods required for recharging the charge storage capacitance (C27; C26) between different charge states and, on this basis, to generate at least one of: (i) a pulse-width modulation control signal; and (ii) a frequency control signal.

2. The microcontroller as claimed in claim 1, characterized in that the adjustments of the controllable current source (SQ1; SQ2) and of the controllable current sink (SS1; SS2) can be varied in relation to a reference current level that can be predefined by means of a reference current source (IR), in each case with a resolution of at least 8 bits.

3. The microcontroller as claimed in claim 2, characterized in that the reference current level for the charging and the discharging current can be predefined by means of a nonreactive resistor (R30).

4. The microcontroller as claimed in claim 1, characterized in that the control means for the device (SQ1, SS1; SQ2, SS2) for the alternate charging and discharging of the charge storage capacitance have at least one read/write memory (DR1, DR2; DR5, DR6).

5. The microcontroller as claimed in claim 1, characterized in that the control means of the device (SQ1, SS1; SQ2, SS2) for the alternate charging and discharging of the charge storage capacitance have switching means (US1; FL1) which are used to switch over the device (SQ1, SS1; SQ2, SS2) from charging to discharging of the charge storage capacitance (C27; C26) when a first voltage value is reached, and to switch over this device (SQ1, SS1; SQ2, SS2) from discharging to charging of the charge storage capacitance (C27; C26) when a second, lower voltage value is reached.

6. The microcontroller as claimed in claim 5, characterized in that the first voltage value or second voltage value can be adjusted by means of a read/write memory (DR7).

7. The microcontroller as claimed in claim 1, characterized in that a frequency divider (FT1) or a pulse divider is provided which, at its input, detects the changeover of the device (SQ1 SS1; SQ2, SS2) for the alternate charging and discharging of a charge storage capacitance from discharging to charging or from charging to discharging, and divides the input signal into signals for the alternating control of alternately switching means (V2, V3) of the switched-mode power supply.

8. The microcontroller as claimed in claim 1, characterized in that the microcontroller (MC) has interfaces (1-28) for registering external signals or data and a device (A) for evaluating the external signals or data and for the program-controlled determination of actuating values for controlling the device (SQ1, SS1; SQ2, SS2) for the alternate charging and discharging of the charge storage capacitance.

9. A ballast for operating at least one electric lamp (LP1, LP2), which has an inverter, at least one load circuit coupled to the inverter and having terminals (X1-X8) for the at least one electric lamp (LP1, LP2), a control circuit for controlling the switching means (V2, V3) of the inverter and a DC supply circuit for the inverter, the control circuit comprising a microcontroller (MC) having a device (G) for controlling the switching means (V2, V3) of the inverter, characterized in that the device (G) for controlling the switching means of the inverter hasa device (SQ1, SS1) for the alternate charging and discharging of a first charge storage capacitor (C27), control means for this device (SQ1, SS1) for controlling the charging operations and the discharging operations, and evaluation means which are used to evaluate the duration of the alternate charging and discharging operations of the first charge storage capacitor (C27) and on this basis to generate one of: (i) a frequency control signal; and (ii) a pulse-width modulation control signal for controlling the switching means (V2, V3) of the inverter, and wherein the ballast further comprises a frequency divider (FT1) or a pulse divider which: (i) at its input, detects the changeover of the device (SQ1, SS1) for the alternate charging and discharging of the first charge storage capacitor from discharging to charging or from charging to discharging; and (ii) divides the input signal into signals for the alternating control of the switching means (V2, V3) of the inverter.

10. The ballast as claimed in claim 9, characterized in that the ballast has a hearing device equipped with a controllable switching means (V4) to apply a hearing current to the lamp electrodes (E1-E4) of the at least one electric lamp (LP1, LP2) and the microcontroller (MC) has a comparator (K1), which compares the charge state of the first charge storage capacitor (C27) with a reference value for the lamp electrode heating and which is used to generate a control signal for the pulse-width modulation of the controllable switching means (V4) of the heating device.

11. The ballast as claimed in claim 10, characterized in that the reference value can be adjusted by means of a read/write memory (DR4).

12. The ballast as claimed in claim 10, characterized in that the microcontroller (MC) has synchronization means (SR1) for synchronizing the controllable switching means (V4) of the heating device with a switching means (V2) of the inverter.

13. The ballast as claimed in claim 9, characterized in thatthe DC supply circuit has a step-up converter for power factor the microcontroller (MC) has a second device (SQ2, SS2) for the alternate charging and discharging of a second charge storage capacitor (C26), the microcontroller (MC) has second control means for this second device (SQ2, SS2) for controlling the charging operations and the discharging operations, and the microcontroller (MC) has second evaluation means which are used to evaluate the time periods required for recharging the second charge storage capacitor between different charge states and, on this basis, to generate at least one of: (i) a pulse-width modulation control signal; and (ii) a frequency control signal for the controllable switching means (V1) of the step-up converter.

14. The ballast as claimed in claim 13, characterized in that the second evaluation means have a first comparator (K2, K3) to compare the charge state of the second charge storage capacitor (C26) with a first voltage value, and a second comparator (K4) to compare the charge state of the second charge storage capacitor (C26) with a second, lower voltage value, and in that the second control means of the second device (SQ2, SS2) have switching means (FL1) which are used to switch over the second device (SQ1, SS1; SQ2, SS2) from charging to discharging of the second charge storage capacitor (C26) when the first voltage value is reached, and to switch over the second device (SQ2, SS2) from discharging to charging of the second charge storage capacitor (C26) when the second, lower voltage value is reached.

15. The ballast as claimed in claim 14, characterized in that the first voltage value or the second voltage value can be adjusted by means of a read/write memory (DR7).

16. The ballast as claimed in claim 13, characterized in that the devices (SQ1, SS1; SQ2, SS2) for the alternate charging and discharging of the first and second charge storage capacitors each have a controllable current source (SQ1; SQ2) for applying an adjustable charging current to the first charge storage capacitor (C27) and, respectively, the second charge storage capacitor (C26), and in each case a controllable current sink (SS1; SS2) for applying an adjustable discharging current to the first charge storage capacitor (C27) and, respectively, the second charge storage capacitor (C26).

17. The ballast as claimed in claim 16, characterized in that the settings of the controllable current sources (SQ1; SQ2) and of the controllable current sinks (SS1; SS2) can be varied in relation to a reference current level (IR), in each case with a resolution of at least 8 bits.

18. The ballast as claimed in claim 17, characterized in that the reference current level (IR) for the charging current and the discharging current can be predefined by means of a nonreactive resistor (R30).

19. The ballast as claimed in claim 9, characterized in that the microcontroller (MC) has interfaces (18, 19; 15, 16; 20, 21, 3) for registering operating parameters of at least one of: (i) the inverter; (ii) the at least one electric lamp (LP1, LP2); and (iii) the step-up converter, wherein the microcontroller further has a program-controlled device (A) which is used to evaluate the operating parameters and to determine at least one of: (i) actuating values for controlling the devices (SQ1, SS1; SQ2, SS2) for the alternate charging and discharging of the first and second charge storage capacitors; (ii) the reference value for the lamp electrode heating; and (iii) the first or second voltage value.

20. The ballast according to claim 9, characterized in that the ballast has terminals (J3, J4) and means (DS) for communication with an external control device, and the microcontroller (MC) has interfaces (5, 6) which are coupled to the terminals (J3, J4).

21. A method of operating at least one electric lamp (LP1, LP2) with the aid of a ballast which has an inverter with a control circuit containing a microcontroller (MC) for the switching means (V2, V3) of the inverter and has at least one load circuit coupled to the inverter and having terminals (X1-X8) for the at least one electric lamp (LP1, LP2), characterized in that, with the aid of the microcontroller (MC)a charge storage capacitor (C27) has a charging current and a discharging current alternately applied to it, the duration of the alternate charging and discharging operations of the charge storage capacitor (C27) is evaluated and on this basis a control signal for the alternating control of the switching means (V2, V3) of the inverter is generated, the lamp electrodes (E1-E4) of the at least one electric lamp (LP1, LP2) have a heating current applied to them, the heating current being regulated by means of a controllable switching means (V4), by pulse-width modulated control signals being generated for the controllable switching means (V4) with the aid of a comparator (K1), which compares the charge state of the charge storage capacitor (C27) with a reference value for the lamp electrode heating.

22. The method as claimed in claim 21, characterized in that the reference value is adjusted on the basis of the desired heating power and stored in a read/write memory (DR4) of the microcontroller (MC).

23. The method as claimed in claim 21, characterized in that the controllable switching means (V4) for regulating the heating current are switched on synchronously with a switching means (V2) of the inverter, and the duty cycle of the controllable switching means (V4) for regulating the heating current is smaller than or equal to the duty cycle of the switching means (V2) of the inverter.

24. A method of operating at least one electric lamp (LP1, LP2) with the aid of a ballast which has an inverter with a control circuit containing a microcontroller (MC) for the switching means (V2, V3) of the inverter and has at least one load circuit coupled to the inverter and having terminals (X1-X8) for the at least one electric lamp (LP1, LP2), characterized in that, with the aid of the microcontroller (MC)a charge storage capacitor (C27) has a charging current and a discharging current alternately applied to it, the duration of the alternate charging and discharging operations of the charge storage capacitor (C27) is evaluated and on this basis a control signal for the alternating control of the switching means (V2, V3) of the inverter is generated, the direct current for the power supply of the inverter is regulated by means of a step-up converter, in order to ensure power factor correction a control signal for the controllable switching means (V1) of the step-up converter being generated with the aid of the microcontroller (MC), by a second charge storage capacitor (C26) being recharged between different charge states, and the time periods for recharging the second charge storage capacitor (C26) being evaluated in order to generate the the control signal for the controllable switching means (V1) of the step-up converter.

25. The method as claimed in claim 24, characterized in that, with the aid of a first comparator (K2, K3), the charge state of the second charge storage capacitor (C26) is compared with a first voltage value and, with the aid of a second comparator (K4), the charge state of the second charge storage capacitor (C26) is compared with a second, lower voltage value, the charging operation of the second charge storage capacitor C26) being terminated and the discharging operation of the second charge storage capacitor (C26) being started when the first voltage value is reached, and the discharging operation of the second charge storage capacitor (C26) being terminated and the charging operation being started when the second, lower voltage value is reached.

26. The method as claimed in claim 25, characterized in that at least one of the first voltage value and the second voltage value is adjusted by means of a read/write memory (DR7).

27. The method as claimed in claim 24, characterized in that the charging current is generated by means of a current source (SQ1; SQ2), and the current intensity is adjusted by means of a read/write memory (DR1; DR6).

28. The method as claimed in claim 24, characterized in that the discharging current is generated by means of a current sink (SS1; SS2), and the current intensity is adjusted by means of a read/write memory (DR2; DR5).

29. The method as claimed in claim 24, characterized in that, with the aid of the microcontroller (MC), actual values of operating parameters of at least one of: (i) the inverter; (ii) the at least one electric lamp (LP1, LP2); and (iii) the DC supply circuit of the inverter are monitored and are evaluated, wherein the actual values of operating parameters are monitored and evaluated in order to: (a) control the charging or discharging operations of the first and second charge storage capacitors (C27; C26); (b) determine the reference value for the lamp electrode heating; and (c) determine the first voltage value and the second voltage value.