Patent Application
20210149018
In the quest for ever-safer and more convenient transportation options, many car manufacturers are developing self-driving cars which require an impressive number and variety of sensors. Among the contemplated sensing technologies are multi-input, multi-output radar systems to monitor the distances between the car and any vehicles or obstacles along the travel path. The radar system antennas are expected to operate from within a vehicle bumper or other housing that provides protection, but which may also cause the closest and strongest reflection of signal energy. Accumulations of dirt, mud, and/or ice on the housing can increase the reflection but also reduce the RADAR transmission power through the housing, thus degrading the ability of the radar system to detect obstacles. In any event, the reduced transmission through the RADAR housing may reduce the dynamic range of the detection signal, degrading signal-to-noise ratio (SNR) and thereby reducing accuracy of range and velocity measurements.
The problems identified above may be addressed at least in part by employing a radar-housing tone discriminator in frequency-modulated continuous wave (FMCW) radar systems. One illustrative embodiment of a radar system includes: a signal generator, a variable phase shifter element, and a mixer. The signal generator supplies a frequency modulated continuous wave (FMCW) signal to a transmit antenna protected by a housing, which causes a housing reflection having a frequency offset from the FMCW signal. The variable phase shifter element derives a reference signal from the FMCW signal by applying a time-dependent phase shift based on the frequency offset. The mixer obtains a receive signal including said housing reflection and multiplies the receive signal with the reference signal to produce a downconverted signal.
One illustrative embodiment of a radar signal downconversion method includes: supplying a frequency modulated continuous wave (FMCW) signal to a transmit antenna protected by a housing, which causes a housing reflection having a frequency offset from the FMCW signal; derives a reference signal from the FMCW signal by applying a time-dependent phase shift based on the frequency offset; and multiplying the reference signal with a receive signal including said housing reflection to produce a downconverted signal.
An alternative radar system embodiment includes: a signal generator, a mixer, and an analog-to-digital converter. The signal generator supplies a FMCW signal to a transmit antenna protected by a housing, which causes a housing reflection having a frequency offset from the FMCW signal. The mixer derives a downconverted signal from a receive signal including said housing reflection. The amplitude of the housing reflection can be determined from the downconverted signal.
Each of the foregoing embodiments can be employed individually or in conjunction, and may include one or more of the following features in any suitable combination: 1. a controller that adjusts the time-dependent phase shift to minimize phase noise in the downconverted signal. 2. a controller that adjusts the time-dependent phase shift to maximize a DC component of the downconverted signal. 3. An ADC that determines an amplitude of the housing reflection from the downconverted signal. 4. a safety engine that signals an error condition if the housing reflection exceeds a predetermined threshold. 5. the variable phase shifter element derives a reference signal from the FMCW signal, which the mixer uses to produce a downconverted signal. 6. a controller that applies a phase rotation to the reference signal to obtain an in-phase product signal with a minimum phase noise or maximum DC component, and a quadrature phase product signal with a maximum phase noise or minimum DC component. 7. a high-pass filter having a cutoff frequency below which low frequency components of the downconverted signal are attenuated. 8. the mixer multiplies the receive signal by a reference signal that shifts the offset frequency to or above the high pass filter cutoff frequency. 9. a processor that determines the amplitude of the housing reflection from phase noise in the downconverted signal near the cutoff frequency. 10. the mixer multiplies the receive signal by an in-phase reference signal or a quadrature-phase reference signal to produce the downconverted signal.
It should be understood that the following description and accompanying drawings are provided for explanatory purposes, not to limit the disclosure. To the contrary, they provide the foundation for one of ordinary skill in the art to understand all modifications, equivalents, and alternatives falling within the scope of the claims.
Using the interface, sensors, and actuators, ECU 202 may provide automated parking, assisted parking, lane following, lane-change assistance, obstacle and blind-spot detection, adaptive cruise-control, automated braking, autonomous driving, and other desirable features. In an automobile, the various sensor measurements are acquired by one or more electronic control units (ECU), and may be used by the ECU to determine the automobile's status. The ECU may further act on the status and incoming information to actuate various signaling and control transducers to adjust and maintain the automobile's operation.
To gather the necessary measurements, the ECU may employ, e.g., a constant frequency continuous wave (CW) or a frequency-modulated continuous wave (FMCW) radar system. Radar systems operate by emitting electromagnetic waves which travel outward from the transmit antenna before being reflected back to a receive antenna. The reflector can be any moderately reflective object in the path of the emitted electromagnetic waves. By measuring the travel time of the electromagnetic waves from the transmit antenna to the reflector and back to the receive antenna, the radar system can determine the distance to the reflector. For FMCW radar, the transmit signal frequency changes over time (chirp) and the target distance will be proportional to the frequency difference between the receive signal and the reference signal, which is the mixer out frequency. If multiple transmit or receive antennas are used, or if multiple measurements are made at different positions, the radar system can determine the direction to the reflector and hence track the location of the reflector relative to the vehicle. With more sophisticated processing, multiple reflectors can be tracked. At least some radar systems employ array processing to “scan” a directional beam of electromagnetic waves and construct an image of the vehicle's surroundings.
A control interface 305 enables the ECU or other host processor to configure the operation of the transceiver chip 300, including the test and calibration peripheral circuits 306 and the transmit signal generation circuitry 307. Circuitry 307 generates a carrier signal within a programmable frequency band, with a programmable chirp rate and range. Splitters and phase shifters enable the multiple transmitters TX-1 through TX-4 to operate concurrently if desired, and further provide a reference “local oscillator” signal to the receivers for use in the downconversion process. In the illustrated example, the transceiver chip 300 includes 4 transmitters (TX-1 through TX-4) each of which is fixedly coupled to a corresponding transmit antenna 308. In alternative embodiments, multiple transmit antennas are selectably coupled to each of the transmitters.
X
T(t)=cos(ωct+φn(t),
neglecting the frequency modulation. The (angular) carrier frequency is ωc, the phase noise is φn(t), and time is represented by t. The phase noise in the transmit signal may arise from various internal and environmental causes.
The transmit signal 402 passes through the radar housing (e.g. radome), which may be a bumper or other protective housing, 404 to encounter an obstacle 406, from which it returns to the receive antenna 302 as a reflection 408. The radar housing 404 also causes a reflection 409 to return to the receive antenna 302. The reflection is the closest and often the strongest reflection. The receive antenna signal can accordingly be represented as
X
R(t)=AB COS[(ωc+ωB)(t−tdB)+φn(t−tdB)]+AT COS[(ωc+ωT)(t−tdT)+φn(t−tdT)]
where AB and AT are the amplitudes of the radar housing reflection and target reflection, respectively, tdB and tdT are their round trip travel times, and ωB and ωT are the frequency offsets from the current carrier frequency resulting from the travel time delays.
X
LO(t)=cos(ωct+φn(t)+ϕ0)
where ϕ0 is the phase shift, which differs by
for the two reference signals.
The mixer 510 multiplies the receive antenna signal by the reference signal to obtain the downconverted (“intermediate frequency”) signal:
Y
IF(t)=LPF{XR(t)XLO(t)}
where LPF{ } is a low-pass filter operation that blocks the upconverted frequency component and serves as an anti-aliasing filter that blocks any tones above the ADC Nyquist frequency. A variable gain amplifier 522 operates using a gain setting from the controller 504 to provide automatic gain control for the mixer output. One or more filters 524, 526, may provide the LPF operation above as well as a high-pass filtering operation to block any undesired low frequency components prior to digitization by the ADC 303.
In the absence of any other targets, the result is just the radar housing tone with phase offsets due to travel time and phase noise
Y
IF(t)=AB COS[(ωc+ωB)tdB+ωBt+φn(t−tdB)−φn(t)+ϕ0]
As the housing is on the order of 4 cm away, tdB is expected to be well within the coherence period of the phase noise, such that φn(t−tdB)−φn(t)<<π. Because the housing position is known, the frequency offset ωB can be determined from the programmed sweep rate of the FMCW signal.
Before proceeding further with the analysis relating to the operation of phase shifter element 508, we pause here to note that the high pass filter 524 shown in
In a second contemplated embodiment of a technique for detecting radar housing reflectivity changes, rather than monitoring the radar housing tone frequency, the controller causes the signal generator 502 to generate a constant frequency CW signal, thereby eliminating any frequency offset in the receive signal. The downconverted signal from the mixer 510 is then essentially a DC signal with an amplitude which may be determined by, or at least dominated by, the reflection from the radar housing. The high pass filter is entirely or selectably omitted to enable the DC signal measurement by the ADC 303.
With regard to the first and second contemplated embodiments discussed above, the phase shifter element 508 is optional and may be omitted or bypassed for the reflectivity monitoring. Returning now to the analysis regarding the operation of the phase shifter element, we demonstrate potential advantages to its inclusion.
If we use the phase shifter element to provide a phase shift of:
then
Y
IF(t)≈ABtdB{dot over (φ)}n(t)≈ABφn(t).
On the other hand, if the phase shifter instead provides a phase shift of:
ϕ0=−ωBt−(ωc+ωB)tdB,
then
Y
IF(t)=AB COS[Φn(t−tdB)−φn(t)]≈AB
Stated in words, if the controller 504 modulates the phase shift provided by element 508 using the housing tone plus a constant phase component (which equals to the product between the round trip travel time to the radar housing tone plus the carrier tone), the housing tone is suppressed from the downconverted signal, leaving (in the absence of an obstacle) only a DC component approximately proportional to the amplitude of the radar housing reflection 409. If instead the controller 504 modulates the phase shift provided by element 508 using the radar housing tone, a constant quadrature component, and the constant phase component above (which equals to the product between the round trip travel time to the radar housing tone plus the carrier tone), the housing tone is suppressed from the downconverted signal, leaving only an approximation of the product between the radar housing reflection amplitude and the phase noise φn(t).
Extending this analysis to the situation where there is at least one target reflection in the antenna receive signal, the downconverted signal becomes:
Y
IF(t)=AB COS[(ωc+ωB)tdB+ωBt+φn(t−tdB)−φn(t)+ϕ0]+AT COS[(ωc+ωT)tdT+ωTt+φn(t−tdT)−φn(t)+ϕ0]
For the quadrature phase shift,
Y
IF(t)≈ABφn(t)+AT COS[(ωT−ωB)t+φn(t−tdT)−φn(t)],
(neglecting a constant phase term (ωc+ωT)(tdT−tdB)), and for the in-phase shift,
Y
IF(t)≈AB+AT COS[(ωT−ωB)t+φn(t−tdT)−φn(t)].
In other words, the downconverted signal with the constant quadrature phase component of the reference signal has the phase noise from the radar housing tone converted into an amplitude-modulated noise source, degrading the amplitude SNR, whereas the downconverted signal with the constant in-phase component of the reference signal includes only a DC component equal to the amplitude of the radar housing tone reflection, making it the preferred setup for analysis to detect obstacle reflections and determine associated distances and velocities. Conversely, adding the quadrature phase component to the reference signal is the preferred setup for characterizing the phase noise.
In practice, additional phase shift between the receive and reference signals may accumulate due to contributions from components along the transmit and receive paths, and may vary based upon age or environmental effects. The controller 504 may adapt the constant component of the phase shifter element 508 to minimize phase noise in the downconverted signal, or alternatively to maximize amplitude noise of the DC component of the downconverted signal. Alternatively, the DSP may capture both in-phase and quadrature-phase contributions and apply an adaptive phase rotation with the same optimization metric. Thus the use of a variable phase shifter element 508 to suppress the radar housing tone enables at least these three contemplated embodiments of a technique for minimizing the effect of phase noise. The residual phase noise in the target reflection is expected to be uncorrelated with the reference signal phase noise, but we can generalize the above embodiments to uncorrelated phase from target reflection noise under the assumptions that: φn(t−tdT)−n(t)<<π
In addition to providing a way to improve SNR and to characterize phase noise, the receiver of
As previously noted, some radar system embodiments may high-pass filter the downconverted signal to remove DC and attenuate other low frequency components typically associated with reflections from the housing and other nearby surfaces not intended to be measured by the radar system. If such filtering is employed where it is nevertheless desirable to monitor the radar housing reflectivity, the reference signal may be modified by the phase shifter element 508 to shift the radar housing tone to a frequency ω0 above the cutoff frequency of the high-pass filter by applying a linear time dependent phase shift of ω0t. Thus, in a fourth contemplated embodiment of a technique for detecting radar housing reflectivity changes, element 508 may provide a variable phase shift of
ϕ0=(ω0)t−(ωc+(ωB)tdB,
and the DSP may then be configured to measure signal energy at frequency ω0.
In a fifth contemplated embodiment of a technique for detecting radar housing reflectivity changes, the high-pass filter may be permitted to block the low frequency or DC component representing the peak of the radar housing reflection signal, with the recognition that the phase noise φn(t) extends over a significant frequency band and is expected to include components that would pass through the high-pass filter, perhaps with an acceptable degree of attenuation. Thus when the constant quadrature component of the reference signal is applied the downconverted signal would still include information about the amplitude of the radar housing reflection.
This receiver embodiment is suitable for implementing at least the first, second, and fifth contemplated embodiments of a technique for detecting radar housing reflectivity changes as previously described. Because this receiver embodiment does not modulate the phase shifter to drive the radar housing tone to DC, the downconverted signal may be represented as:
Y
IF(t)=AB COS[(ωc+COB)tdB+ωBt]+AT COS[(ωc+COT)tdT+ωTt],
where phase noise is neglected. To enable the DSP to monitor of the radar housing reflection amplitude AB, the high-pass filter may be omitted or its cutoff frequency set below that of the expected radar housing tone ωB. Note, however, that this embodiment lacks the radar housing tone phase-noise suppression provided by at least some of the previously described embodiments.
Numerous other modifications, equivalents, and alternatives, will become apparent to those of ordinary skill in the art once the above disclosure is fully appreciated. It is intended that the following claims be interpreted to embrace all such modifications, equivalents, and alternatives where applicable.
