This application claims priority to International PCT Application No. PCT/IB2011/052066 filed May 11, 2011, which claims priority to South African Application No. 2010/03368 filed May 12, 2010, the entire content of each of which are fully incorporated herein by reference.
This invention relates to modulation of signals. In particular, the invention relates to an OFDM modulation method, to a method of controlling the peak-to-average power ratio of an OFDM modulated signal, to a method of demodulating a method OFDM signal, to an OFDM modulator, to an OFDM demodulator and to an OFDM communication arrangement.
The inventors are aware of an OFDM signal schemes. However, a major disadvantage of an OFDM signal, is it's high peak-to-average power ratio (PAPR). Various methods have been suggested to reduce the PAPR, which includes clipping, coding, non-linear companding transforms, partial transmitted sequence, selective mapping, active constellation extension, tone reservation and constant envelope phase modulation. The inventors sought a method which requires low implementation complexity, which does not require any additional bandwidth expansion and which does not require side information to reconstruct the original message signal. Furthermore, the method should not lead to a severe bit error rate degradation as the number of carriers increase. The present invention aims to address these requirements and is descriptively referred to as offset modulation of an orthogonal frequency division multiplexing (OM-OFDM) signal.
According to one aspect of the invention, there is provided an OFDM modulation method, which includes
The method may include the prior step of independently scaling the real and imaginary components of the OFDM signal.
The scaling of the real and imaginary components of the OFDM signal may be given by:
where and respectively refer to the real and imaginary components of the OFDM message signal, ζ refers to a constant division term and where Φ1(t) and Φ2(t) represent the scaled real and imaginary OFDM components. These Φ1(t) and Φ2(t) are interchangeable.
The method may include the prior step of obtaining an inverse Fourier transform (IFT) of the OFDM message signal.
The IFT may be an N-point inverse Fourier transform given by:
where Ts is the symbol duration and Xk represents the complex signal written as ak+jbk.
The method may include the step of adding an offset term to the scaled real component of the OFDM signal.
The method may include the further step of subtracting the phase modulated OFDM scaled imaginary component from the phase modulated OFDM scaled real component.
The phase modulated addition of an offset term to the scaled real portion of the OFDM signal and the subtraction of the phase modulated OFDM scaled imaginary component can be referred to as offset modulation of an orthogonal frequency division multiplexing (OM-OFDM) signal and given by:
cos(2πfct+Φ1(t)+Ψos)−cos(2πfct+Φ2(t))
where fc is the carrier frequency, Φ1(t) and Φ2(t) represent the scaled real and imaginary OFDM components and Ψos represent the offset term.
The offset term Ψos may be chosen such that Ψos is sufficiently large and Φ1(t) and Φ2(t) are sufficiently small. Typically, Ψos may be approximately twenty two times larger than Φ1(t) and Φ2(t). The combination of Φos and ζ terms ensure that the receiver can successfully detect the original signal.
The method may include the further step of adding or subtracting a dominant frequency component to the modulated OFDM signal.
The dominant frequency component may be given by:
where γ is a dominant frequency component control factor, β is the adapted phase deviation of the real and imaginary OFDM signal and J0(β) is a Bessel function of the first kind of order 0 and argument β. The adapted phase deviation (β) may be determining by averaging the real and imaginary deviation of an OFDM signal, which can be represented by α1 and α2, respectively.
Thereafter these real and imaginary OFDM deviations are scaled and referred to as the adapted phase deviation of the real and imaginary signal OFDM signal (β), this process can be represented as follows, where E[.], is the expected value
According to another aspect of the invention, there is provided a method of controlling the PAPR of an OFDM signal in accordance with the method as described above, which further includes
The method may include, an OFDM modulator adjusting the dominant frequency component control factor
The method may further include, an OFDM demodulator detecting the dominant frequency component control factor by examining the PAPR of the incoming signal, from which the Ψos, ζ and γ terms can be extracted by using a look-up table.
According to another aspect of the invention, there is provided a method of demodulating an OFDM signal, which includes
The method may include the further step of demodulating the OFDM modulated signal, by removing the high frequency components in order to obtain the difference between the scaled real OFDM, offset term and the scaled imaginary OFDM components, which can be given by
−Φ2+Φ1+Ψos.
The method may include the further step of demodulating the incoming OFDM modulated signal, by multiplying it by a scaled phase modulated sinusoidal. Where the phase of the sinusoid is the scaled difference between the scaled imaginary OFDM components, the scaled real OFDM and offset term. This multiplication factor can be given by
The method may include the further step of demodulating the incoming OFDM signal, by removing the high frequency components in order to obtain the summation of the scaled real OFDM, offset term and the scaled imaginary OFDM components, which can be given by
Φ1+Ψos+Φ2.
The method may include the further step of demodulating the incoming OFDM modulated signal in order to obtain the real and imaginary OFDM components.
The method may include the further step of obtaining a fast Fourier transform (FFT) of the OFDM signal.
The method may include the further step of passing the signal through an equalizer.
According to another aspect of the invention, there is provided an OFDM modulator, which, when operated, executes an OFDM modulation method as described above.
An OFDM modulator, which includes a scaling unit for scaling the imaginary and real components of the transformed input signal;
According to another aspect of the invention, there is provided an OFDM demodulator which, when operated, executes a method of demodulating a modulated OFDM signal as described above.
According to another aspect of the invention, there is provided an OFDM demodulator, which includes
According to another aspect of the invention, there is provided a communication arrangement, which includes any one or both of a modulator as described above and a demodulator as described above.
The invention will now be described, by way of example only with reference to the following drawing(s):
In the drawing(s):
In
The modulator receives an input message signal 11 at a transmitter end, this signal is modulated and after passing this modulated signal via the communication channel 100, this signal is feed into the OM demodulator which provides a reconstructed output message signal 13 at a receiver end.
The OM-OFDM modulator 20 is shown in more detail in
which can be considered to be an OFDM signal, where Ts is the symbol duration and Xk represents the complex signal, which may also be written as ak+jbk. This signal may be modulated using the method described below.
At 22, by using a serial to parallel (S/P) converter, the incoming signal is separated into it's real and imaginary components.
The output of 22 produces real and imaginary components of the signal, which are then both scaled at 24 and 26 respectively, by dividing them with a constant scaling factor ζ represented by:
where and refer to the real and imaginary parts of the OFDM message signal, ζ refers to a constant division term, whereas Φ1(t) and Φ2(t) represent the equivalent scaled real and imaginary OFDM mapping. These Φ1(t) and Φ2(t) components are interchangeable.
At 28 a constant term ΨOS is added to the real component of the signal.
At 30 (30.1, 30.2, 30.3, 30.4), the adapted real and imaginary components are phase modulated. The difference between the respective phase modulated signals is taken at 32 (32.1, 32.2).
At 34 (34.1, 34.2) the baseband phase modulated signal is moved to a carrier frequency.
At 36 the resultant consinusoidal with the constant term ΨOS is given by:
cos(2πfCt+Φ1(t)+ΨOS)−cos(2πfCt+Φ2(t)).
or represented as:
where ΨOS refers to the offset term. The parameters (ΨOS, ζ) are chosen such that_ΨOS>>Φ2(t)−Φ1(t), when ΨOS is sufficiently large and Φ2(t), Φ1(t) are sufficiently small. In this instance the ΨOS term will dominate the expression.
At 38 the dominant frequency control factor given by,
is subtracted from the combined signal before the signal is transmitted at 40 across the communication channel 100.
In
The dominant frequency
is reinstated at 63.
At 64 the received signal is squared producing a series of baseband and high frequency components which can be expressed as
1−cos(4πfct+Φ1+Ψos+Φ2)−cos(−Φ2+Φ1+Ψos)+½ cos(2Φ2+4πfct)+½ cos(4πfct+2Φ1+2Ψos).
At 66 the high frequency components of the squared signal are removed by using a low pass filter (LPF), the resultant signal can be given by
1−cos(−Φ2+Φ1+Ψos).
At 68 a constant term is added to the baseband signal, which can be given by
cos(−Φ2+Φ1+Ψos).
At 70 the inverse cosinousoidal process is performed producing
−Φ2+Φ1+Ψos).
At 72 an additional constant multiplication term is introduced resulting in
At 74 the signal is phase modulated by a sinusoid resulting in
At 76 the signal is multiplied by a constant multiplication factor resulting in
At 78 the signal is transformed by a division process this results in
At 79 the received input signal is multiplied by a sinusoidal carrier in addition to the term at 78, this results in
At 80 the high frequency components are removed by using a low pass filter this results in a baseband signal represented by
At 82 a multiplication factor is introduced, this results in
At 84 an inverse co-sinusoidal operation is performed this results in
At 88 a multiplication factor is introduced this results in
Φ1+Ψos+Φ2.
The subsequent steps at 90 (90.1, 90.2), 92 (92.1, 92.2), 94 and 96 (96.1, 96.2) are used to isolate the real and imaginary OFDM components. The parallel to serial convertor (P/S) at 99 combine these real and imaginary OFDM components.
The most prominent advantage of OM-OFDM is that by controlling the dominant frequency, the PAPR of the signal can be controlled. Without prior knowledge of the dominant frequency at the OM-OFDM demodulator, the dominant frequency can be determined by means of a look-up table which maps the PAPR of the signal against the other parameters. Therefore the dominant frequency can be determined without additional signal information. As an example the lookup table for a 16 quadrature amplitude modulation scheme is as follows:
At 63 to 99 the signal is demodulated with a uniquely developed OM-OFDM demodulation scheme and at 96 (96.1, 96.2) the OFDM real and imaginary components are combined and presented at 99.
In
The inventors found that the OM-OFDM modulation method provides a spectrally efficient signal in which the PAPR can be adjusted without removing information from the signal and which does not lead to a severe bit error degradation. Furthermore, as the PAPR is used to determine the dominant frequency, no side information needs to be transmitted.
The inventors are of the opinion that the invention provides substantial advantages in the application of Digital Video Broadcasting (DVB), Worldwide Interoperability for Microwave Access (WiMax) and Long Term Evolution (LTE).
Number | Date | Country | Kind |
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2010/03368 | May 2010 | ZA | national |
Filing Document | Filing Date | Country | Kind | 371c Date |
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PCT/IB2011/052066 | 5/11/2011 | WO | 00 | 6/12/2013 |
Publishing Document | Publishing Date | Country | Kind |
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WO2011/141879 | 11/17/2011 | WO | A |
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Entry |
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International Search Report and Written Opinion from related application PCT/IS11/52066, mailed Jan. 17, 2012. |
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Number | Date | Country | |
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20150319022 A1 | Nov 2015 | US |