This application claims the priority under 35 U.S.C. § 119 of European patent application no. 15290065.0, filed Mar. 11, 2015 the contents of which are incorporated by reference herein.
This disclosure relates to a module for a radio receiver, and in particular, although not exclusively, to a digital front end module.
Modern radio receivers, such as those used for television applications, typically comprise both analogue and digital sections in order to efficiently process signals. A difficulty found in conventional radio receivers relates to providing the required channel selectivity and image rejection properties in an efficient manner.
According to a first aspect there is provided a module for a radio receiver, the module comprising:
The use of the correction path in parallel with the main signal path allows simpler filtering techniques to be applied prior to error detection and therefore simplifies implementation of the module. The simplification enabled by the use of the correction path can be afforded because filtering and error detection is applied to extracted signals rather than signals on the main path. The main path signals themselves may not be filtered by the correction path. The filtering techniques applied in the correction path may therefore not be constrained by the requirement of maintaining signal fidelity in a desired channel, as may be the case for signals on the main signal path. Simpler filtering may therefore be applied in the correction path than would be desirable for signals on the main signal path.
The in-phase and/or quadrature signals may comprise a desired channel. The second signal path may comprise a filter. The filter may be configured to filter the extracted in-phase and/or quadrature signals by passing a first portion of the desired channel. The filter may be configured to filter the extracted in-phase and/or quadrature signals by rejecting a second portion of the desired channel.
The filter may be a band pass filter. The band pass filter may be configured to perform band pass filtering on the extracted in-phase and/or quadrature signals prior to detecting an error in the extracting in-phase and/or quadrature signals. The band pass filter may be configured to pass only a sub-band of the desired frequency band. The band pass filter may be configured to pass a zero intermediate frequency, ZIF, signal or a near-zero intermediate frequency, NZIF, signal. The filtering performed by the second signal path may comprise band pass filtering extracted ZIF or NZIF in-phase and quadrature signals. The second signal path may comprise a second band pass filter provided in parallel with the first band pass filter. The second band pass filer may be configured to pass the other of the ZIF signal and the NZIF signal. The second signal path may further comprise a selector configured to enable either the first or second band pass filter for signal processing.
The module may comprise a channel selection unit at the output. The channel selection unit may be configured to select a channel and provide the channel for post processing.
The second signal path may be configured to extract the in-phase and/or quadrature signals at an extraction point of the main signal path. The second signal path may be configured to apply the correction to the in-phase and quadrature signals at a correction point of the main signal path. The extraction point may be downstream of the correction point in order to provide a feedback loop.
The error may comprise an amplitude mismatch. The module may comprise an amplitude mismatch detection unit configured to detect the amplitude mismatch. The module may comprise an amplitude mismatch correction unit configured to apply a correction to the in-phase and/or quadrature signals in the main signal path based on the detected amplitude mismatch. The error may comprise a phase mismatch. The module may comprise a phase mismatch detection unit configured to detect the amplitude mismatch. The module may comprise a phase mismatch correction unit configured to apply a correction to the in-phase and/or quadrature signals in the main signal path based on the phase mismatch. The error may comprise a direct current, DC, offset. The module may comprise a DC offset detection unit configured to detect the DC offset. The module may comprise a DC offset correction unit may be configured to apply a correction to the in-phase and/or quadrature signals in the main signal path based on the direct current offset.
According to a further aspect there is provided a radio receiver comprising:
The filter may be configured to pass signals having a centre frequency that is an eighth of a sampling frequency of the one or more analogue-to-digital converters.
According to a further aspect there is provided a method of operating a module for a radio receiver, the method comprising:
There may be provided a computer program, which when run on a computer, causes the computer to configure any apparatus, including a module, radio receiver, circuit, controller or device disclosed herein or perform any method disclosed herein. The computer program may be a software implementation, and the computer may be considered as any appropriate hardware, including a digital signal processor, a microcontroller, and an implementation in read only memory (ROM), erasable programmable read only memory (EPROM) or electronically erasable programmable read only memory (EEPROM), as non-limiting examples. The software may be an assembly program.
The computer program may be provided on a computer readable medium, which may be a physical computer readable medium such as a disc or a memory device, or may be embodied as a transient signal. Such a transient signal may be a network download, including an internet download.
One or more embodiments will now be described, by way of example only, and with reference to the accompanying figures in which:
The in-phase and quadrature analogue signals 106, 108 are provided to respective analogue-to-digital converters 110, 112 which digitize the analogue signals based on a sampling frequency Fs and provide a digital in-phase signal 114 and a digital quadrature signal 116 to a digital front end module 118.
The digital front end module (DFE) 118 performs various tasks. Two typical tasks for the DFE 118 are providing channel selection and image rejection. Channel selection can be achieved using multirate and multistage cascaded filters, implemented in a polyphase way and based on a cascaded integrator comb (CIC) decimator, for example. In order to reduce circuit complexity in NZIF examples, baseband conversion may be provided prior to channel selection. Image rejection by the DFE 118 is discussed further below with reference to
Returning to
The digital front end 300 receives digital in-phase and quadrature signals I(n), Q(n) from an analogue-to-digital converter 302 that provides the functionality of the analogue-to-digital converters discussed with reference to
The digital front end 300 comprises a number of subunits which provide the functionality of channel selection and image rejection. These subunits are provided in a linear chain, discussed in sequence below. Each subunit acts on both the in-phase and quadrature signals.
An integrated comb filter 304 down filters the in-phase and quadrature signals I(n), Q(n) by a factor R. A group 305 of units 306, 308, 310 provide detection and correction of the signals. The group 305 in this example comprises an I/Q DC offset unit 306, an I/Q phase mismatch unit 308 and an I/Q amplitude mismatch unit 310 in that order in series. The I/Q DC offset block 306 provides an estimate of, and compensation for, any DC offset in the signals. The I/Q phase mismatch unit 308 provide an estimate of, and compensation for, a phase mismatch in the signals. The I/Q amplitude mismatch unit 310 provides an estimate of, and compensation for, an amplitude mismatch in the signals.
The signals output by the group 305 of units are filtered by a low pass filter 312 and are processed by a derotator 314 in order to achieve base band conversion by applying a frequency translation exp(−j*2*pi*IFin*n*T) to the signal (where j^2=−1, n is a sample with a period T and IFin is an input intermediate frequency). The converted baseband signal is passed through a plurality of dowsampling filters 316, 318 and a channel selection filter 320 before undergoing digital automatic gain control (AGC) by an AGC unit 322. The signal output by the AGC unit 322 is then passed through a plurality of upscaling filters 324, 326. A ZIF signal from the upsampling filters 324, 326 is then translated back to an intermediate frequency by a rotator 328 by applying a frequency translation exp(j*IFout*n*T). Only a real part of the signal output by the derotator 328 is further processed. The real signal is passed through yet a further plurality of upscaling filters 330, 332. Finally, the signal output by the upscaling filters 330, 332 is converted to an analogue signal by digital-to-analogue converter 334. The digital-to-analogue converter 334 may be provided outside of the digital front end as part of a post processing step.
The ability of the digital front end 300 to compensate for mismatch in the desired channel is influenced by the interfering signals outside the band of interest of the desired channel. This can result in digital front end 300 applying poor compensation in systems in which the I/Q imbalance is frequency dependent. The performance of the digital front end 300 is discussed below with reference to
In
In order to simulate the frequency dependent I/Q imbalance phenomenon, an amplitude mismatch of 0.1 dB and a phase mismatch of 1 degree (0.1745 rd) is applied to the second interfering tone I2 and an amplitude mismatch of 0.12 dB and a phase mismatch of 1.1 degree (0.192 rd), is applied to the first interfering tone I1. That is, the first interfering tone I1 at 13.33 MHz has a 10% higher mismatch than the second interfering tone I2 at −2.333 MHz.
This digital front end of
A number of strategies may be used in order to manage the I/Q imbalance in systems in which the I/Q imbalance is frequency dependent.
A method in which management is performed in the analogue domain involves performing in-phase/quadrature (I/Q) calibration using a single tone reference signal at system start-up or when the desired channel changes frequency. A problem encountered with such a method is that the requirement for generating a tone and performing calibration increases settling time during use and results in greater circuit complexity.
Another method for management of I/Q imbalance for use in systems such as DVB-T involve generating reference pilots in the frequency domain in order to estimate/compensate for the I/Q imbalance. However, such methods require a frequency domain transformation and are dependent on the specific type of system in which it is implemented.
The digital front end 500 receives digital in-phase and quadrature signals I(n), Q(n) from an analogue-to-digital converter 502. The signals are down filtered by a factor R by an integrated comb filter 504 and a further low pass filter 512, which in this example acts on the signal before an in-phase and quadrature mismatch correction block 505a.
Each of the in-phone and quadrature signals from the mismatch correction block 550 follow two paths including a channel path 507a and an image path 507b.
The channel path 507a selects a wanted channel by band pass filtering unwanted frequencies around a channel signal frequency band. In the channel path 507a, the signals from the mismatch correction block 505a undergo zero intermediate frequency (ZIF) conversion by a channel path derotator 514a by applying a frequency translation exp(−j*2*π*IFin*n*T) to the channel path signal (where j is the square root of minus one, n is a sample with a period T and IFin is an input intermediate frequency). The converted signals are passed through a plurality of channel path decimator filters 516a, 518a and a channel path selection filter 520a before undergoing digital automatic gain control (AGC) by a channel path AGC unit 522a. The gain controlled signals are passed through a further plurality of channel path interpolator filters 524a, 526a before being translated back to output intermediate frequency IFout signals by a channel path rotator 528a which applies an exp(j*2*π*Fout*n*T) translation (where ZIF is a zero intermediate frequency signal on the channel path).
The image path 507b selects an image signal by band pass filtering unwanted frequencies around an image signal frequency band. The image signal frequency band is outside of the channel signal frequency band. In the image path 507b, the signals from the mismatch correction block 505a undergo zero intermediate frequency (ZIF) conversion by an image path rotator 514b by applying a frequency translation exp(j*2*π*IFin*n*T) to the channel path signal. The translated signals are passed through a plurality of image path decimator filters 516b, 518b and an image path channel selection filter 520b before undergoing amplification by an image path amplifier 522b. A gain of the image path amplifier 522b is controlled in response to a gain applied to the channel path 507a by the channel path AGC unit 522a. The gain controlled signals are passed through a further plurality of image path interpolator filters 524b, 526b before being translated back to output intermediate frequency IFout signals by image path derotator 528b which applies an exp(−j*ZIF*nT) translation (where ZIF is a zero intermediate frequency signal on the channel path).
The respective in-phase and quadrature signals output by the channel and image paths 507a, 507b are merged in a merging unit 530 in order to combine the channel path signals with the corresponding outputs of the image path 507b. The merging unit 530 comprises an in-phase summing unit 532 for the in-phase signals and a quadrature summing unit 534 for the quadrature signals.
In-phase and quadrature signals that are output by the merging unit 530 are provided to an I/Q mismatch detection unit 536. The I/Q mismatch detection unit 536 subsequently processes the combined channel and image path signals. The I/Q mismatch detection unit 536 determines whether there are any mismatch errors between the in-phase and quadrature signals. The I/Q mismatch detection unit 536 then provides error signals back to the in-phase and quadrature mismatch correction block 505a so that it can process signals it receives in order to reduce the error represented by the error signals. Errors in the in-phase and quadrature signals include amplitude mismatch, phase mismatch and direct current offset. In this way, the digital front end 500 compensates for I/Q mismatch errors in a corrected channel signal that includes the main signal path signal.
However, in order to provide such compensation the digital front end 500 requires a duplicate path (the image path 507b) and so increases the chip area required by the system which consequently increases the cost of the digital front end 500. The provision of a duplicate path also increases power consumption when the digital front end 500 is in use.
The further examples described below are directed to a different approach for addressing image rejection issues such as those encountered in radio frequency (RF) receivers systems. These examples may be implemented using recursive algorithms that are valid for zero intermediate frequency (ZIF) or near zero intermediate frequency (NZIF) modes. Such algorithms may be suitable for hybrid receivers (Analog/Digital) and can be implemented in systems that use a variety of standards, such as DVB-T, DVB-C, ATSC. As such, the examples can be considered to be independent of a standard that is used.
In this example, a channel selection unit 607 is provided at the output terminal 604 of the main signal path 606. The channel selection unit 607 provides signal processing related to selection of the desired channel. The channel selection unit 607 of the main signal path may be configured to perform channel selection using filtering and/or Fourier transformation of the received in-phase and quadrature signals. For example, the channel selection unit 607 may comprise one or more of the components of the channel path described with reference to
A second signal path is connected in parallel with the main signal path 606. The second signal path may also be referred to as a detection path 612 (estimation path) or correction path. The detection path 612 is configured to extract, from the main signal path 606, in-phase and quadrature signals at a signal extraction point 614 of the main signal path 606.
The extracted signals are digital time domain signals. Extracted in-phase and quadrature signals may be referred to as extracted signals.
The detection path 612 is configured to filter and perform signal processing on the extracted signals in order to detect mismatch errors in the extracted signals. Filtering the extracted signals may comprise applying a filter in which a portion of the desired channel is allowed to pass and another portion of a desired channel is rejected. That is, a subset of the desired channel frequency band may be filtered out in the correction path 612 before further processing. The filter may be configured to reject interference generated by, for example, analogue filters in an RF front end. The filtering may comprise, for example, band pass filtering. The error detected by the detection path 612 may relate to an amplitude mismatch, a phase mismatch or a direct current offset in the extracted signals, for example.
The detection path 612 is configured to apply a correction to in-phase and quadrature signals on the main signal path 606 at a correction point 616 of the main signal path 606 based on the error detected in the extracted signals. The detection path 612 of the module 600 may be used in the processing of ZIF or NZIF signals to compensate for a frequency dependent I/Q imbalance. A passband filter may be used in a NZIF mode or a low pass may be used in a ZIF mode. Different sets of processing loop gains may be put in place to speed up a convergence time of the module 600, depending on the intended application. An example implementation of a correction path is discussed below with regard to the
Returning to
The signal extraction point 614 is downstream of the correction point 616 on the main signal path 606 in order to provide a feedback loop. The correction applied at the correction point 616 may comprise modifying a gain of a filter in the main signal path 606 at the correction point 616 based on the processed extracted signals. The error may comprise one or more of an amplitude mismatch, a phase mismatch and a direct current offset. Depending on the type of error detected by the correction path 612, the correction path 612 may apply a correction to the main path signals at the correction point 616 based on the amplitude mismatch, phase mismatch or DC offset detected in the extracted signals.
The detection path 612 therefore enables an error in extracted signals to be determined separately from the main signal path 606 and a correction to be applied to the main path signals based upon the detected error. The correction path 612 may be configured to filter out frequency dependent mismatch created by the analogue filters (such as those in the radio-frequency front-end illustrated in
The module 600 addresses a number of the issues encountered in the I/Q imbalance compensation schemes described with reference to
The analogue-to-digital converter 701 is configured to digitize analogue in-phase and quadrature signals I(t), Q(t) based on a sampling frequency Fs and provides a digital in-phase signal I(n) and a digital quadrature signal to the module 700.
The module 700 is similar to the module described with reference to
The module 700 of
The correction path 712 has five sections including an NZIF filtering section 718, 720, a ZIF filtering section (not shown), a mode selector 722, error detection units 724, 726, 728 and error correction units 730, 732, 734, the arrangement and operation of which are described below. The error correction units 730, 732, 734 are provided at a correction point 716 on the main signal path 706.
The NZIF filtering section and the ZIF filtering section are each coupled between the extraction point 714 and respective NZIF and ZIF inputs of the mode selector 722. The components of the NZIF filtering section and the ZIF filtering section are configured to perform filtering of extracted in-phase and quadrature signals in order to remove artefacts, such as those caused by analogue filters in an RF front end.
The NZIF filtering section comprises a NZIF band pass filter 718 and a NZIF digital variable gain amplifier 720. The NZIF band pass filter 718 may be configured to pass a frequency band, or a portion of a band, associated with an NZIF channel. The ZIF filtering section (not shown) is similar to the NZIF filter and comprises a ZIF band pass filter and a ZIF digital variable gain amplifier. The ZIF band pass filter may be configured to pass a frequency band, or a portion of a band, associated with a ZIF channel. The digital variable gain amplifiers are each configured to increase the amplitude of the filtered I/Q components.
The band pass filters in the NZIF and ZIF filtering sections are configured to capture sufficient energy from the wanted channel in the extracted signals for efficient estimation of I/Q mismatch while removing adjacent interferers in order to avoid frequency dependent effects caused by analogue filters, such as those commonly used in radio frequency front ends. The module 700 reduces the effect of the frequency dependent mismatch created by the analogue filters around their cut-off frequencies by filtering out these components prior to I/Q imbalance detection. The band pass filters do not necessarily provide channel selection in a strict sense, which can be provided at the output on main signal path by a channel selection unit, for example.
The band pass filters in the NZIF and ZIF filtering sections may be relatively simple because the extracted signals are separate from the main signal path and so any artefacts introduced into the extracted signals by the band pass filters do not affect the output signals provided by the module 700. For example, the desired channel may be provided in a desired frequency band and the band pass filters may, as a side effect, substantially distort the extracted signals within the desired frequency band without deleteriously affecting operation of the main signal path 706 of the module 700. The NZIF digital variable gain amplifier 720 may therefore be optimised for determining an I/Q imbalance without being constrained to maintain channel signal fidelity.
The mode selector 722 is configured to enable the NZIF or ZIF band pass filter for signal processing by outputting the signals from either the ZIF filtering section or the NZIF filtering section in response to a selection signal received at a selection input terminal 723. The selection signal can be set by a controller in accordance with a mode of operation of the receiver.
The error detection units 724, 726, 728 include an amplitude mismatch detection unit 724, a phase mismatch detection unit 726 and a DC offset detection unit 728. The error correction units 730, 732, 734 include an amplitude mismatch correction unit 730, a phase mismatch correction unit 732 and a DC offset correction unit 734. The error detection units 724, 726, 728 have a first gain setting (Epsilon-0) and a second gain setting (Epsilon-1).
The first and second gain settings can be used to adjust the characteristics of the correction units in order to provide a desired mismatch precision for a given settling time.
The amplitude mismatch detection unit 724 is configured to detect, or estimate, an amplitude mismatch error between the in-phase and quadrature signals. This amplitude mismatch detection unit 724 may apply a recursive algorithm. In this example, the amplitude mismatch detection unit 724 is configured to:
The amplitude mismatch correction unit 730 is configured to subtract the amplitude mismatch correction signal, which relates to the mismatch error, from the in-phase signal 708 in the main signal path 706 at an amplitude mismatch correction point. The amplitude mismatch correction point is downstream of the amplitude mismatch extraction point in terms of signal propagation along the main signal path 706. In an alternative example, the amplitude mismatch correction unit 730 may operate on the quadrature signal. That is, the amplitude mismatch correction signal may be subtracted from the quadrature signal 710 in the main signal path 706.
The phase mismatch detection unit 726 is configured to detect a phase mismatch between the in-phase and quadrature signals and achieve a correlation between the inphase and quadrature signals. In an ideal case, the inphase and quadrature signals are orthogonal and so yield zero error. In this example, the phase mismatch detection unit 726 is configured to:
The phase mismatch correction unit 732 is configured to subtract the phase mismatch correction signal, which relates to the estimated phase mismatch, from the quadrature signal 710 in the main signal path 706 at a phase mismatch correction point. The phase mismatch correction point is downstream of the phase mismatch extraction point in terms of signal propagation along the main signal path 706. In an alternative example, the phase mismatch correction unit 732 may operate on the in-phase signal. That is, the phase mismatch correction signal may be subtracted from the in-phase signal 708 in the main signal path 706.
The direct current (DC) offset mismatch detection unit 728 is configured to detect a DC offset between the in-phase and quadrature signals. The DC offset detection unit 726 may be configured to estimate DC offset level using a recursive feedback loop algorithm. In this example, the offset mismatch detection unit 728 is configured to:
a) apply a one symbol period delay, using a third delay block 729, to a sum of the extracted in-phase signal and feedback from an output of the third delay block 729; and
The steps a) and b) above may be performed simultaneously.
In the DC offset correction unit 734, the in-phase DC offset correction signal is deducted from the in-phase signal 708 in the main signal path 706 and the quadrature DC offset correction signal is deducted from the quadrature signal 710 in the main signal path 706.
As discussed in regard to the module of
This improved performance of the module 700 of
In
In
Returning to
One of the four coefficients is cancelled out due to the selection of the relationship between the intermediate frequency and sampling frequency and so the complexity of the filter is reduced by almost 25%.
The bandwidth is 6 MHz in this example. The DC component 1002 is attenuated by more than 30 dB. A frequency component 1006 at 12.5 MHz are typical of those found around a cut-off frequency of an analogue filter of a radio frequency front end. The frequency component 1006 is attenuated by 57 dB by the FIR filter.
The filter and module may be provided in an advanced television standards committee (ATSC) tuner.
The method may be implemented using only hardware, only software or a combination of hardware and software. In general, the method may perform any of the steps described with reference to the module of
Number | Date | Country | Kind |
---|---|---|---|
15290065 | Mar 2015 | EP | regional |
Number | Name | Date | Kind |
---|---|---|---|
3739351 | Forbes | Jun 1973 | A |
3742461 | Forbes | Jun 1973 | A |
5617060 | Wilson | Apr 1997 | A |
5809090 | Buternowsky | Sep 1998 | A |
6317589 | Nash | Nov 2001 | B1 |
6744825 | Rimstad | Jun 2004 | B1 |
6990060 | Butash | Jan 2006 | B2 |
7130359 | Rahman | Oct 2006 | B2 |
7209526 | Kim | Apr 2007 | B2 |
7248625 | Chien | Jul 2007 | B2 |
7251298 | Hietala | Jul 2007 | B1 |
7421040 | Cowley | Sep 2008 | B2 |
7769105 | McIntire | Aug 2010 | B1 |
7962949 | Busson | Jun 2011 | B2 |
7983648 | Qian | Jul 2011 | B2 |
8121214 | Tal | Feb 2012 | B2 |
8135086 | Sychaleun | Mar 2012 | B1 |
8249129 | Fudge | Aug 2012 | B2 |
8269551 | Liao | Sep 2012 | B2 |
8326252 | Li | Dec 2012 | B2 |
8340167 | Feng | Dec 2012 | B2 |
8363695 | Mochizuki | Jan 2013 | B2 |
8379766 | Lococo | Feb 2013 | B2 |
8385471 | Lococo | Feb 2013 | B2 |
8401116 | Jamin | Mar 2013 | B2 |
8416899 | Laurent-Michel | Apr 2013 | B2 |
8503545 | Li | Aug 2013 | B2 |
8649464 | Eitel | Feb 2014 | B2 |
8798216 | Pullela | Aug 2014 | B2 |
8995589 | Qiu | Mar 2015 | B1 |
9031171 | Smail | May 2015 | B2 |
9036740 | Khoury | May 2015 | B2 |
9042500 | Wong | May 2015 | B1 |
9264156 | Wilhelmsson | Feb 2016 | B2 |
9325553 | Kaukovuori | Apr 2016 | B2 |
9571137 | Robert | Feb 2017 | B2 |
9621206 | Jamin | Apr 2017 | B2 |
9748964 | Kabir | Aug 2017 | B1 |
20010036152 | Butash | Nov 2001 | A1 |
20030206603 | Husted | Nov 2003 | A1 |
20040037375 | Cowley | Feb 2004 | A1 |
20040146121 | Brown | Jul 2004 | A1 |
20040183589 | Duncan et al. | Sep 2004 | A1 |
20040229589 | Behzad | Nov 2004 | A1 |
20050110550 | Shi et al. | May 2005 | A1 |
20050243219 | Yun | Nov 2005 | A1 |
20060068739 | Maeda | Mar 2006 | A1 |
20060083335 | Seendripu | Apr 2006 | A1 |
20060222117 | Rahman | Oct 2006 | A1 |
20060256216 | Takahiko | Nov 2006 | A1 |
20060256895 | Cho | Nov 2006 | A1 |
20070030931 | Arambepola | Feb 2007 | A1 |
20070047634 | Kang | Mar 2007 | A1 |
20070047669 | Mak | Mar 2007 | A1 |
20070071132 | May | Mar 2007 | A1 |
20070104291 | Yoon | May 2007 | A1 |
20070123188 | Mo | May 2007 | A1 |
20070211837 | Zipper | Sep 2007 | A1 |
20070223608 | Nakayama | Sep 2007 | A1 |
20070298733 | Cole | Dec 2007 | A1 |
20080055010 | Lerner | Mar 2008 | A1 |
20090190698 | Schwarzmueller | Jul 2009 | A1 |
20090310717 | Huang | Dec 2009 | A1 |
20100104045 | Santraine | Apr 2010 | A1 |
20100159837 | Dent | Jun 2010 | A1 |
20100159858 | Dent | Jun 2010 | A1 |
20110110473 | Keehr | May 2011 | A1 |
20110134334 | Stevenson | Jun 2011 | A1 |
20110134986 | Skull | Jun 2011 | A1 |
20110142172 | Bode | Jun 2011 | A1 |
20110150069 | Jamin | Jun 2011 | A1 |
20110206162 | Lococo | Aug 2011 | A1 |
20110222633 | Pullela | Sep 2011 | A1 |
20120230372 | Park | Sep 2012 | A1 |
20130039444 | Porret | Feb 2013 | A1 |
20130322570 | Harada | Dec 2013 | A1 |
20140029700 | Viswanathan | Jan 2014 | A1 |
20140056386 | Jomatsu | Feb 2014 | A1 |
20140133606 | Mochizuki | May 2014 | A1 |
20140169410 | Tanaka | Jun 2014 | A1 |
20140169504 | Smail | Jun 2014 | A1 |
20140177761 | Patel | Jun 2014 | A1 |
20140198689 | Loh | Jul 2014 | A1 |
20140241466 | Cajegas, III | Aug 2014 | A1 |
20150117577 | Valadon | Apr 2015 | A1 |
20150172086 | Khoshgard | Jun 2015 | A1 |
20150200628 | Rutten | Jul 2015 | A1 |
20150236887 | Kaukovuori | Aug 2015 | A1 |
20160028424 | Jamin | Jan 2016 | A1 |
Entry |
---|
Extended European Search Report for Patent Appln. No. 15290065.0 (dated Dec. 2, 2015). |
Nicolay, Thomas et al; “Baseband Signal Processing of Digital Video Broadcasting Direct-Conversion Zero-IF Tuners”; IEEE Transactions on Consumer Electronics, vol. 51, Issue 1; pp. 48-53 (2005). |
Schuchert, Andreas et al; “Frequency Domain Equalization of IQ Imbalance in OFDM Receivers”; IEEE International Conference on Consumer Electronics; pp. 28-29 (2001). |
Number | Date | Country | |
---|---|---|---|
20160269208 A1 | Sep 2016 | US |