Multicarrier personal access communication system

BACKGROUND OF THE DISCLOSURE

1. Field of the Invention

This invention relates generally to wireless telecommunications, and, more particularly, to methodologies and concomitant circuitry for high-speed wireless access services to fixed and mobile telecommunications users.

2. Description of the Background

The Personal Access Communication System (PACS) system provides high performance, low complexity radio technology for interoperable wireless access using licensed and unlicensed frequency spectra in the 2 GHz emerging technologies frequency band. A representative article which discusses both the history and technological innovations of the PACS system is the article entitled “PACS: Personal Access Communication System—A Tutorial”, authored by Noerpel et al. and published in the IEEE Personal Communications, June 1996, pages 32-43.

It is well-known in the industry that the Orthogonal Frequency Division Multiplexing (OFDM) technology is an effective means of mitigating Intersymbol Interference (ISI) on multipath fading channels when operated in environments where the Root-Mean-Square (RMS) delay spread is a significant impairment.

However, the art is devoid of teachings and suggestions for combining OFDM and PACS to extend the range of applications and capabilities of PACS, especially in an environment wherein the RMS delay spread is significant.

In accordance with the present invention, a so-called Multicarrier Personal Access Communication System (MPACS) system is a highly modular PACS-based system that combines the advantages of OFDM and PACS, as well as the well-known Time Division Multiple Access (TDMA) technology. MPACS is arranged to support higher-speed (higher than the 32 kbps of PACS) wireless access services to fixed and mobile users. For example, nominal user data rates of 32-to-356 kbps are attainable, and ever the higher speed of 768 kbps is possible for short ranges.

A primary design objective for MPACS, at the physical layer system, is that of retaining as many of the link-level system parameters of PACS as possible, in order to minimize incompatibilities between them. In this respect, the same Time Division Multiple Access (TDMA) frame format and approximately the same Radio Frequency (RF) channel structure is deployed in MPACS. For example, the main licensed version of the PACS system parameters that are of interest in the MPACS system are listed in Table 1 below.

The PACS baseband signal is based on a Square-Root Raised Cosine (SRC) transmit filter and has a single-sided 3 dB bandwidth of 96 kHz, and a roll-off factor of α=0.5, resulting in a total single-sided bandwidth of 144 kHz. Since the transmitted signal is double the single-sided bandwidth, the total bandwidth is 288 kHz (this bandwidth is frequently specified as 300 kHz as in Table 1). In MPACS a higher level Quadrature Amplitude Modulation (QAM), that is, above 4-level QAM (which is essentially the same as Differential Quadrature Phase Shift Key (DQPSK) of Table 1) is used to increase the data range beyond the values of Table 1. This use of higher level QAM has an impact on error rate performance and/or achievable range, and provision for each must be accommodated in the design of MPACS.

TABLE 1

PARAMETER

VALUE

Available Spectrum

10 MHz

RF Channel Bandwidth

300 kHz

Transmission Bit Rate

384 kbps

Symbol Transmission Rate

192 kbauds

User Throughput/Timeslot

32 kbps

Modulation Type

π/4 - DQPSK

Nyquist Filter Roll-Off Factor

α = 0.5

TDMA Frame Duration

2.5 ms

Number of Timeslots/Frame

8

Timeslot Duration

312.5 μs

Duplexing Method

Frequency Division Duplex/TDMA

SUMMARY OF THE INVENTION

The shortcomings and limitations of the prior art are obviated, in accordance with the present invention, by a methodology and concomitant circuitry wherein, generally, the advantageous properties of ODFM and PACS are combined, along with TDMA properties, to extend the range and capabilities of PACS.

Broadly, in accordance with one method aspect of the present invention, a method for propagating a sequence of input bits as a time-division multiple-access frame over a wireless channel includes: (a) converting the sequence of input bits into a corresponding set of input symbols wherein each of the input symbols represents a unique plurality of the input bits; (b) partitioning the time-division multiple-access frame into a plurality of slots each having a prescribed bandwidth and assigning one of the slots to the sequence of input bits; and (c) subdividing the prescribed bandwidth of the assigned slot into a plurality of orthogonal frequency division multiplexed subchannels each conveying a corresponding one of the input symbols from the set.

In accordance with another aspect of the invention, the method for propagating a sequence of input bits as a wireless channel signal includes: (a) converting the sequence of input bits into a corresponding set of input symbols wherein each of the input symbols represents a unique plurality of the input bits; (b) modulating each of the input symbols in the set to produce a corresponding set of complex symbols; (c) computing the inverse Discrete Fourier Transform of the set of complex symbols to produce a transformed set of symbols; (d) augmenting the transformed set of symbols with cyclic prefix symbols determined with reference to the transformed set to produce a set of output symbols; and (e) processing the set of output symbols to generate the wireless channel signal.

Broadly, in accordance with the system aspects of the present invention, these aspects include circuitry commensurate with the foregoing methodologies.

BRIEF DESCRIPTION OF THE DRAWINGS

The teachings of the present invention can be readily understood by considering the following detailed description in conjunction with the accompanying drawings, in which:

FIG. 1

illustrates OFDM channel bandwidth subdivision into narrowband, overlapping subchannels with a given separation between the subcarrier center frequencies;

FIG. 2

is an exemplary channel structure in accordance with the present invention for one RF channel and 4-QAM modulation on each OFDM subchannel;

FIG. 3

is an illustrative embodiment of a transmitter as a high-level block diagram in accordance with the present invention;

FIG. 4A

depicts the constellation points P

n

, n=1, 2, 3, and 4 for a minimal-energy 4-QAM constellation;

FIG. 4B

depicts the construction of a 2-bit Gray code for association with the constellation points of

FIG. 4A

;

FIG. 4C

depicts the association of the Gray code of

FIG. 4B

with the constellation points of

FIG. 4A

;

FIG. 5

depicts the constellation points and associated Gray code for an 8-QAM constellation;

FIG. 6

depicts the constellation points and associated Gray code for a 16-QAM constellation;

FIG. 7

depicts the constellation points and associated Gray code for a 32-QAM constellation;

FIG. 8A

depicts an exemplary constellation addressed by an input data stream to produce a complex symbol output for a fixed-rate modulator;

FIG. 8B

depicts an exemplary set of constellations addressed by an input data stream and a QAM-select signal to produce a complex symbol output for a variable-rate modulator;

FIG. 9

depicts an illustrative embodiment of the PAR and pilot symbol inserter of is

FIG. 3

;

FIG. 10

is an illustrative embodiment of a receiver as a high-level block diagram in accordance with the present invention;

FIG. 11

is illustrative of the symbol timing and carrier frequency offset estimator of

FIG. 10

;

FIG. 12

is illustrative of the channel estimator of

FIG. 10

;

FIG. 13

is illustrative of the frequency domain equalizer and special symbol remover of

FIG. 10

;

FIG. 14

is illustrative of the demodulator of

FIG. 10

using a table-look up algorithm;

FIG. 15

is a flow diagram depicting the operational steps carried out by the transmitter of

FIG. 3

;

FIG. 16

is a flow diagram depicting the operational steps carried out by the receiver of

FIG. 10

;

FIG. 17

is a flow diagram for generating the look-up table utilized for both modulation and demodulation;

FIG. 18

is a flow diagram depicting the operational steps carried out by the modulator of

FIG. 3

; and

FIG. 19

is a flow diagram depicting the operational steps carried out by the demodulator of FIG.

10

.

To facilitate understanding, identical reference numerals have been used, where possible, to designate identical elements that are common to the figures.

DETAILED DESCRIPTION

To fully appreciate the import of the signal processing system of the present invention, as well as to gain an appreciation for the underlying operational principles of the present invention, it is instructive to first present a high-level description of the fundamentals of the OFDM system advantageously deployed by MPACS. This overview also serves to introduce terminology so as to facilitate the more detailed description of the illustrative embodiment, which follows the overview of OFDM.

OFDM Principles

In an OFDM system, the transmission bandwidth is subdivided into N independent narrowband subchannels, with one and only one symbol transmitted on each subchannel. If the effective OFDM block duration is T

E

and the OFDM sampling rate is f

s

=1/T, then the effective OFDM block duration is T

E

=NT. In the case of an orthogonal subcarrier, the minimum bandwidth for each subchannel is B

s

=1/NT, which is also the minimum separation between consecutive overlapping subchannels. Hence the center frequency f

k

of the k

th

subchannel of the baseband OFDM signal is given by:

\begin{matrix} f_{k} = f_{0} + \frac{k}{NT}, where k = 0, 1, \dots, N - 1. & (1) \end{matrix}

A depiction of overlapping subchannels

101

, . . . ,

104

is shown in

FIG. 1

, wherein subchannel

101

is shown as overlapping subchannel

102

, whereas subchannel

102

overlaps both subchannels

101

and

103

. The subcarrier center frequencies are also depicted in

FIG. 1

, along with the minimum separation of 1/T between subcarrier center frequencies. Negative frequencies in the baseband signal can be accounted for by noting, for example, that if N is even, then f

0

=(−N/2+1)/NT. It should be noted that even though the minimum separation between the subchannels results in overlapping subchannels, orthogonality between the subchannels is still maintained.

Overlapping subchannel OFDM systems can be implemented using the Discrete Fourier Transform (DFT). In such implementations, however, each subchannel possesses a spectrum with non-negligible sidelobes. The sidelobes can be reduced by extending the data frame to include a so-called “cyclic prefix” composed of, for example, the last V samples from the output of an inverse DFT (IDFT), and prefixed to the IDFT.

Moreover, practical OFDM techniques always include a guard interval T

G

to remove Intersymbol Interference (ISI) caused by non-orthogonal subchannels due to deviations for the ideal model for orthogonal subchannels. When a guard interval is used, the resulting OFDM symbol duration becomes T

S

=T

E

+T

G

. If the guard interval is composed of V

1

cyclic prefix and V

2

cyclic postfix samples, then T

G

=VT , where V=V

1

+V

2

. As alluded to above, the guard interval is implemented in practice by prefixing (postfixing) the last V

1

(first V

2

) samples from the output of the IDFT to the sequence of IDFT output samples. If the precursor guard interval is greater than the maximum value of the delay of the multipath fading channel, then ISI due to Inter-Block Interference (IBI) is eliminated. If the channel is not ideal, ISI due to Inter-Channel Interference (ICI) (caused by interference among OFDM subchannels) cannot be totally eliminated. In practice, the guard interval is typically chosen to be in the range of 10% to 20% of the effective OFDM block length, although for MPACS the range may also be dependent upon the TDMA radio link, and may be adjusted accordingly.

Another parameter that is important to an OFDM system implementation is the achievable data rate. If there are N OFDM subchannels, and an M-ary modulation technique is deployed to assign the same number of data bits on each subchannel, then the maximum achievable rate is given by:

\begin{matrix} R_{b} = \frac{N \log_{2} M}{T_{E} + T_{G}} . & (2) \end{matrix}

As a result of the guard interval, the maximum achievable data rate is reduced by a factor of V/(N+V). As an example of this effect, suppose that N=4, M=16, T

E

=312 μs, and T

G

=88 μs, so that T

E

+T

G

=400 μs. Then R

b

=40 kbps instead of the rate of about 51.3 kbps assuming no guard interval.

On a frequency-selective fading channel, the subchannels suffer from different levels of attenuation, which might lead to different values of Signal-to-Interference Ratio (SIR) and Bit Error Rate (BER) on individual subchannels. Without transmitter and/or receiver optimization, the theoretical data rate of equation (2) may not be achievable because information will be lost on some subchannels. A simple method to increase BER performance (at the expense of throughput) is not using weak subchannels for transmission. The decision to leave out subchannels is, for example, based on a ranking of amplitudes in the subchannels.

OFDM Considerations

One of the principal advantages of OFDM is its utility for transmission at very near optimum performance on multipath fading channels without equalization. If OFDM is to operate at close to its theoretical performance limits, it must be arranged to overcome two main limitations, namely:

(a) peak envelope variations resulting in large Peak-to-Average power Ratio (PAR) or, equivalently, “crest factor”; and

(b) sensitivity to timing and frequency offset errors.

With respect to the first limitation, the composite OFDM baseband signal exhibits non-constant envelope variations. This property makes OFDM sensitive to nonlinear effects in high-power amplifiers. This phenomenon is frequently expressed as the PAR or crest factor. The peak envelope of the OFDM signal increases linearly with N, and the average envelope is proportional to the square root of N; hence, the theoretical PAR increases in proportion to the square root of N. There are a number of well-known techniques to overcome the effects of PAR.

With respect to the second limitation, carrier frequency offset can degrade the performance of OFDM much more than single-carrier systems because it introduces interference among the multiplicity of subcarriers in the OFDM signal that are closely spaced in frequency compared to the channel bandwidth. There are two deleterious effects caused by an offset in frequency between the transmitter and receiver local oscillator: (i) a reduction of signal amplitude in the output of each of the subcarriers; and (ii) introduction of ICI from other subcarriers that are no longer orthogonal to each other. In practical applications, carrier synchronization is achieved using pilot-tone-aided or pilot-symbol-aided techniques. The carrier frequency offset should be limited to about an order of magnitude or less than the subchannel bandwidth to insure adequate performance of the OFDM system. The impact of carrier frequency offset is less severe in OFDM systems that use differential detection techniques.

Timing offset in OFDM systems also rotates and attenuates transmitted symbols, and introduces ICI from other subchannels due to loss of orthogonality. If the guard interval T

G

is greater than the timing offset, then ICI due to loss of orthogonality from timing errors is mitigated. There are known techniques to reduce carrier frequency and timing offset errors.

By way of reiteration, some important advantages of OFDM to be exploited in accordance with the present invention are summarized as follows:

(a) allows the use of coding to trade bandwidth for peak envelope reduction;

(b) supports dynamic channel assignment to improve spectrum efficiency; and

(c) allows flexible use of aggregated timeslots/carriers for service needs.

On the other hand, two limitations of OFDM must be recognized and dealt with in accordance with the techniques of the present invention, these limitations being: (i) peak envelope variations require a PAR reduction scheme; and (ii) robust timing and/or frequency recovery circuits are required.

Illustrative Embodiment in Accordance with the Present Invention

General Considerations

The MPACS system, which illustrates the principles of the present invention, combines the advantages of OFDM and PACS to extend the range of applications and capabilities of, and provide performance improvements, over PACS. For expository purposes, the same TDMA frame format and RF channel structure as in PACS are used for the MPACS system. It is readily contemplated that these strictures may be relaxed, such as by eliminating the TDMA protocol, or by using OFDM in a multiple access realization.

As to the physical link layer, new aspects which have been incorporated into the MPACS system are highlighted (but this enumeration is not intended to be limiting):

(1) utilization of M-ary Quadrature Amplitude Modulation (M-QAM) with rectangular and square constellations, such as 4-QAM, 8-QAM, 16-QAM, and 32-QAM, as discussed in detail in the sequel; based upon the teachings of M-QAM, it is readily visualized how Quadrature Phase-Shift Keying (QPSK), Differential QPSK (DQPSK), and π/4 DQPSK may also be implemented

(2) one IDFT per Subscriber Unit (SU) transmitter, and one or more DFT's per SU receiver, depending upon the number of diversity branches. The same number of IDFT-DFT processors are required at each Radio Port (RP)

(3) Serial-to-Parallel (SPC) and Parallel-to-Serial Converter (PSC) pairs in the SU and RP transceivers

(4) PAR reduction circuitry

(5) guard interval insertion and removal circuitry

(6) symbol timing and carrier frequency offset estimation circuitry

(7) a look-up table with concomitant search algorithm to implement modulation by a modulator at the transmitter of the MPACS system, and demodulation in a demodulator at the receiver

(8) reduced computational complexity using the Fast Fourier Transform to implement the IDFT and the DFT

Channel Structure

An example of the frame structure of the MPACS system in accordance with the present invention is shown in

FIG. 2

for the exemplary case of 4-phase modulation (e.g., 4-QAM) on each OFDM subchannel. Eight timeslots

201

,

202

, . . . ,

208

, each 312.5 μs long, are associated with each 2.5 ms MPACS TDMA frame, on each RF channel having a bandwidth of nominally 288 kHz. Each timeslot supports one SU, which is the OFDM block length for single-timeslot operation.

In this configuration, OFDM is used to subdivide the RF channel bandwidth into N OFDM subchannels, and one symbol is transmitted on each subchannel. For example, suppose the requirement is to transmit 120 information bits between RP and SU in one timeslot. For four-phase modulation on each subchannel, then N=60 OFDM subchannels are required. If an exemplary single-carrier bandwidth is 288 kHz, then the OFDM subchannel bandwidth is B

s

=4.8 kHz, and the effective OFDM block duration T

E

=208.33 μs. Since the OFDM block length T

s

=312.5 μs, then the potential OFDM guard interval is T

G

=104.17 μs.

In practice, only a portion of this guard interval is dedicated to cyclic prefix samples used for multipath delay spread mitigation. This is because the maximum propagation delay of the multipath channel is, practically, well below this value. The remainder of the potential guard interval may, illustratively, be allocated to PAR symbols, channel coding, symbol timing recovery, and pilot recovery symbols (if utilized). Multi-timeslot operation is achieved by simply extending the number of OFDM subchannels in multiples of N. Table 2 below summarizes a range of radio link parameters, for single- and multiple-timeslot operations per SU, assuming four-phase modulation.

The MPACS system supports a 384 kbps transmission rate, and 32-256 kbps access speeds with four-phase modulation such as 4-QAM. For higher level QAM, the values of Table 2 remain the same; however, there is a corresponding increase in the data rate above that which is achievable to four-phase modulation. The calculations to derive Table 2 are those deployed in the previous paragraph.

TABLE 2

3-dB Bandwidth/

Subchannel

Time-

Sampling Rate

Number of

Bandwidth

Number of

slots

(kHz)

Carriers

(kHz)

Guard Samples

Per SU

Min.

Max.

Min.

Max.

Min.

Max.

Min.

Max.

1

192

288

60

90

3.20

4.80

0

30

2

192

288

120

180

1.60

2.40

0

60

3

192

288

180

270

1.067

1.60

0

90

4

192

288

240

360

0.800

1.20

0

120

5

192

288

300

450

0.640

0.960

0

150

6

192

288

360

540

0.533

0.800

0

180

7

192

288

420

630

0.457

0.685

0

210

8

192

288

480

720

0.400

0.600

0

240

Illustrtive Embodiment

1.) Transmitter

300

A high-level block diagram of transmitter

300

in accordance with the present invention is shown in

FIG. 3

wherein transmitter

300

is composed of the basically series arrangement of: (a) serial-to-parallel converter

310

, which receives a serial stream of input data bits over input path

310

and produces K (K and K are used interchangeably in the sequel) parallel data bit streams on paths

311

, . . . ,

312

; (b) modulator

320

, coupled to device

310

, for receiving K parallel data bit streams over paths

311

, . . . ,

312

as produced by converter

310

—modulator

320

is typically of type QPSK/DQPSK (Quadrature Phase Shift Keying/Differential.QPSK) or M-QAM (M-ary Quadrature Amplitude Modulation); (c) pilot and PAR symbol inserter

330

, coupled to modulator

320

, which also receives K parallel input streams over paths

321

, . . . ,

322

from modulator

320

, and which outputs N (N and N are used interchangeably in the sequel) parallel output streams onto paths

331

, . . . ,

332

; (d) N-point Inverse DFT (Discrete Fourier Transform) processor

340

, responsive to inserter

330

, for receiving N parallel data streams over paths

331

, . . . ,

332

as produced by inserter

330

; (e) PAR (Peak-to-Average) processor

350

, coupled to IDFT processor

340

, for receiving N parallel data streams over paths

341

, . . . ,

342

as produced by IDFT processor

340

; (f) cyclic prefix inserter

360

, coupled to PAR processor

150

, for receiving N parallel data streams over paths

351

, . . . ,

352

as supplied by PAR processor

350

; (g) parallel-to-serial converter

370

, responsive to both PAR processor

350

and inserter

360

, for receiving N+V parallel data streams—N data streams are received from PAR processor

350

over paths

351

, . . . ,

352

, and V data streams are produced by inserter

360

and supplied over paths

361

, . . ,

362

; (h) interference suppression shaping filter (e.g., a raised cosine filter)

380

, responsive to converter

370

, for receiving a serial stream of bits from converter

370

over path

371

; (i) RF processor

390

, responsive to filter

380

, for receiving serial data from filter

380

over path

381

—processor

390

effects: D/A conversion; up-conversion for converting the baseband signal on path 381 to a frequency suitable for over-the-air propagation utilizing a conventional local oscillator (not shown); and RF processing for efficiently propagating the up-converted baseband signal; and (j) over-the-air propagation device

395

(e.g., an antenna) for propagating the signal produced by processor

390

as delivered over path

391

.

By way of relating the OFDM principles to the transmitter

300

of

FIG. 3

, it is instructive to couch the OFDM principles in terms of the elements of FIG.

3

. Accordingly, the OFDM channel spectrum is subdivided into N independent narrowband channels, and a N-point DFT is used to approximate the subchannel carrier frequencies. From Table 2, N can range from 60 to 90 for single timeslot operation, and up to 480 to 720 for multi-timeslot operation, depending upon the bandwidth, and the number of timeslots assigned to each user. K of the N subchannels carry data symbols that are associated with one timeslot, with the remainder of the subchannels carrying PAR symbols plus pilot symbols, if required. No more than one symbol can be assigned to each subchannel. For the single timeslot per user case, 120 (or 60 log

2

M, if M>4) data bits are separated into 60 groups of log

2

M bits each of which is mapped into a complex symbol by modulator

320

and forwarded to IDFT processor

340

via inserter

330

. The IDFT processor inputs are frequency domain samples that are scaled by the complex input symbols. The K-point IDFT processing converts the N complex frequency domain symbols into N complex time domain samples. These time domain samples, which appear on paths

341

, . . . ,

342

, are further processed by PAR processor

350

before cyclic prefix inserter

360

adds an additional V samples to the input of parallel-to-serial converter

370

. The conventional signal processing provided by shaping filer

380

and RF processor

390

prepare the time samples for over-the-air propagation.

Typical values for the parameters are as follows: N=72, K=60, PAR symbols=8, pilot symbols=4, and number of cyclic prefix symbols V=8.

Details of the Transmitter Illustrative Embodiment

Whereas certain of the elements of transmitter

300

are typically realized conventionally, such as IDFT processor

340

, converter

370

, shaping filter

380

, and RF processor

390

, additional description with respect to certain of the other elements is in order to fully elucidate the illustrative embodiment of the present invention; further details pertaining to these latter elements are now presented.

1.1) Serial-to-Parallel Converter

310

Converter

310

is depicted as a simple serial-to-parallel converter. However, it may be desirable in certain embodiments, depending upon system requirements, to expand the capabilities of converter

310

to include error correcting coding, such as forward error correction (FEC) encoding, prior to the conversion of serial input stream

310

to K parallel output streams

311

, . . . ,

312

.

1.2) Modulator

320

The modulation produced by modulator

320

is based upon a look-up table which is generated during the system design phase, that is, it is generated off-line rather than in real-time. The look-up table modulation procedure is composed of two steps, namely, the construction of, for the illustrative embodiment, a QAM constellation table, and then the use of the table to produce electrical signals indicative of constellation symbols which are emitted by modulator

320

over parallel paths

321

, . . . ,

322

.

The symbols are complex, that is, each symbol has an in-phase component and a quadrature component, which for expository purposes can be considered to be the real and imaginary part of a complex symbol. Generally, the complex symbols for a range of M-ary constellations are generated and stored in the table; in this way, the table is versatile enough to handle variable-rate systems wherein each of a plurality of channels can be associated with a different M-ary constellation (e.g., channel

1

may be 4-QAM, channel

2

may be 8-QAM, and so forth). However, the focus of the illustrative embodiment is on fixed-rate systems wherein all channels are M-QAM with M being fixed for all the channels. To access the M-QAM constellation of interest, two sets of indices are employed to address the memory locations; both indices are discussed in more detail shortly. The first index relates to the corresponding Gray-coded bits, and the second index is a label from a set of reduced constellation points.

The layout of the table gives a “minimum energy” set of constellations. Most importantly, the time necessary to produce an output symbol indicative of the input data, which results from a table search, is reduced by a factor of about one-half over conventional table look-up implementations.

Any range of M-QAM constellations can be implemented, but for exemplary purposes, the technique to generate the constellation points for the sequence of M=4, 8, 16, and 32 is presented.

1.2.1) Generation of the M-QAM Constellation Table

The algorithm to generate the constellation table (with a minimal energy constraint) and addresses for constellation points in the table is described by the following steps:

(1) define the set of points in the first quadrant of the M-QAM constellation as S

M

, wherein the x and y coordinates of the points can take on only the values x=±1, ±3, ±5, . . . and y=±1, ±3, ±5, . . .

(2) start with the 4-QAM constellation and label the point in S

4

having x and y coordinates as P

0

, i.e., P

0

→(x,y).

(3) move in a counter-clockwise direction and label the other three points to complete the 4-QAM constellation as P

1

→(−x,y), P

2

→(−x,−y), and P

3

→(x,−y). The depiction of

FIG. 4A

illustrates the location of these constellation points.

(4) assign codes to the 4-QAM constellation based on a Gray code assignment; a table for generating such a Gray code assignment is shown in FIG.

4

B. The table of

FIG. 4B

is then mapped onto the constellation points of

FIG. 4A

to produce the constellation point-Gray code pairs of

FIG. 4C

, which summarizes the constellation points and their associated Gray codes. Thus P

0

has associated Gray code (01), P

1

has associated Gray code (00), P

2

is assigned (10), and P

3

is assigned (11). Moreover, for point P

0

, the x-value or in-phase value is 0 and the y-value or quadrature value is 1; and similar relations for the other constellation points are readily identifiable. The Gray-coded bit values are used to produce the complex symbols in the constellation.

It is noted that the “energy” is generally defined as E=x

2

+y

2

, so for the 4-QAM constellation, E=2 for the signal level values shown in FIG.

4

C. For instance, P

0

is located at (1,1), P

1

is located at (−1,1), and so forth, and each case x

2

+y

2

=2.

(5) labels for the 8-QAM and higher constellation points are determined in an iterative manner. Consider first the next point P

k

→(x,y) in S

M

not already in S

M−1

.

(a) find the lowest energy point x

2

+y

2

not in S

M−1

.

(b) if two or more points have equal energy, then

(i) if the sums (x+y) for the points are equal, choose the point with the smallest y value

(ii) if the sums are different, choose the point with the smallest max {x,y}

(c) choose the three points associated with P

k

as P

k+1

→(−x,y), P

k+2

→(−x,−y), and P

k+3

→(x,−Y)

(6) repeat step (5) until there are M/4 points in S

M

(7) if the M-QAM constellation is square, and 2

L

=M, then assign Gray codes to the constellation points based on an L/2 Gray code assignment to the decomposed in-phase and quadrature components

(8) if the M-QAM constellation is not square, then decompose it into the least number of contiguous square blocks, assign Gray codes to them based upon an int[L/2]-bit and {L−int[L/2]}-bit Gray code assignment. (Note that int[r] means integer division). If there are points in the constellation where Gray codes cannot be assigned, assign other codes that minimize the distance with neighboring points.

The constellation points and labels for 8-QAM are shown in

FIG. 5

based upon the above algorithm.

Similarly, the constellation points and labels for 16-QAM and 32-QAM are shown in

FIGS. 6 and 7

, respectively. (It is noted that

FIGS. 4C and 5

are to scale, but

FIGS. 6 and 7

are not to exact scale, but suffice for expository purposes.)

It is readily contemplated for higher-level constellations that once the points in the first quadrant are generated, the points in the remaining quadrants are easily obtained by mapping.

Also, it is noted that the constellation points shown in

FIGS. 4-7

are for un-normalized constellations. To obtain a M-QAM constellation with a peak value of A volts, each point in these constellations is scaled by A/({square root over (M−1))} for the square M-QAM constellation, and by A/3 and A/5 for 8-QAM and 32-QAM, respectively. It is further noted that each point in the constellation has two ways of being addressed, either by the Gray code label—which is different for each M-QAM constellation—, or by the point assignment number (e.g., P

k

)—which is unchanged for lower constellations. It is clear that constellation tables for 64-QAM or higher can be constructed in a straightforward manner based upon the prior teachings. Finally, it is clear that the constellation can be rotated without affecting the information conveyed by the constellation points.

1.2.2) Generating Complex Symbols from Gray-Code Input

To use the look-up constellation of

FIGS. 4C

,

5

,

6

, or

7

, each of the K-bit streams on each path

311

, . . ,

312

of

FIG. 3

serves as an input to the appropriate look-up table, and a complex symbol is emitted on each of the K output paths

321

, . . ,

322

in correspondence to the input stream. For instance, suppose that the constellation of

FIG. 7

serves as a complex symbol look-up table; this constellation is depicted as 32-QAM storage array

800

in FIG.

8

A. Path

801

is representative of each path

311

, . . . , or

312

of

FIG. 3. A

5-bit input stream arrives on path

801

, and is used to address storage array

800

to generate the complex symbol corresponding to the 5-bit stream. As an example, if the binary input is (10100), then the complex symbol emitted is (−3,1) which corresponds to constellation point P

5

.

It is noted that there are implementation alternatives to the table look-up technique depending upon the particular realization of modulator

320

. For instance, it may be desirable to replicate QAM storage array

800

so that each path

311

,

312

, . . . accesses a separate array

800

; such an implementation is useful for a parallel processing realization. On the other hand, if the overall rate of complex symbol generation is significantly higher than the symbol communication rate, the table look-up may be implemented in software so that there is physically only one storage array

800

(e.g., a data array). In this case, the complex symbols may be produced via a multiplicity of table queries.

The realization of modulator

320

depicted in

FIG. 8B

is used for a variable-rate M-QAM implementation. For this case, storage array

810

is composed of the following sub-arrays: (i) storage area

813

that stores the 4-QAM array of

FIG. 4C

; (ii) storage area

814

that stores the 8-QAM array of

FIG. 5

; (iii) storage area

815

that stores the 16-QAM array of

FIG. 6

; and (iv) storage area

816

that stores the 32-QAM array of FIG.

7

. Now there are two required inputs to storage array

810

, namely, the binary input stream interpreted as a Gray-code appearing on path

811

, and a QAM-select signal on path

812

. The number of bits appearing on path

811

determines which sub-array is to be accessed; for example, a 4-bit stream on path

811

requires the use of area

815

, which QAM select path

812

appropriately chooses. The complex symbol is emitted on path

817

. Again, depending upon the implementation, array

810

may be replicated or be a single array, such as a data array.

1.3) PAR and Pilot Symbol Inserter

330

Null symbols are inserted in the subcarriers that are reserved for PAR reduction, specifically, the middle subcarriers corresponding to the high frequency components that will be attenuated by the transmit filter. PAR processor

350

will populate these subcarriers with their actual values.

Pilot symbols are used for both “Channel and SINR Estimation” and “Symbol Timing and Carrier Frequency Offset Estimation”, which are discussed below when a receiver illustrative embodiment is presented. The pilots for the latter are inserted once every F blocks, where F could be on the order 8 or 16, to be synergistic with the PACS multi-frame (8 frames, 20 ms). For “Channel and SINR Estimation”, the pilots can be any non-zero symbol; for exemplary purposes, the pilots are chosen to be the constellation point representing the number 1. For “Symbol Timing and Carrier Frequency Offset Estimation”, the pilots are simply null symbols, i.e. 0. The block diagram of

FIG. 9

, which is a more detailed representation of inserter

330

, shows an example with P=8 PAR reduction subcarriers and W=4 pilots with K=60; thus N=72.

In

FIG. 9

, for exemplary purposes, it is assumed that K=60, and parallel paths

321

, . . . ,

322

convey these K sets of input bits. The null PAR symbols are inserted symmetrically into these K parallel paths by inserter

335

, that is, of the 68 parallel paths

336

: the first 30 are derived from the K sets of input bits; the next eight are from inserter

336

; and the remaining 30 derived from the rest of the K sets of input bits. Then pilot symbol inserter

337

inserts W pilot symbols symmetrically into the

68

sets of bits to derive the N parallel paths

331

, . . .

332

shown repeated in FIG.

9

. As depicted in

FIG. 9

, the N paths are derived as follows: the first path is associated with inserter

337

; the next

17

paths are derived from path

326

; the next path is associated with inserter

337

; the next

17

paths are derived from path

326

; and so forth.

1.4) Processing by PAR Processor

350

Let ψ (d_{i}) = \max_{k} \frac{{&LeftBracketingBar; d_{i} (k) &RightBracketingBar;}^{2}}{\frac{1}{N} \sum_{i = 0}^{N - 1} {&LeftBracketingBar; d_{i} (l) &RightBracketingBar;}^{2}} = \max_{k} \frac{{&LeftBracketingBar; d_{i} (k) &RightBracketingBar;}^{2}}{σ^{2}}, where

d

i

is a vector representing the N-symbol parallel output of the “N-Point Inverse DFT” processor

340

, for the i

th

OFDM block passing through the transmitter.

Definitions:

Define the following parameters:

(i) C is the target maximum PAR

(ii) P is the number of non-information-bearing subcarriers

(iii) J is the size-P set of indices of subchannels that are non-information-bearing

(iv) μ controls the speed of convergence.

Parameter Settings:

Example parameter settings (with suggested range in parentheses):

(i) C=7 dB (5 dB-9 dB)

(ii) α=3 (2-6)

(iii) η

0

=40 (10-40)

(iv) μ=4 (2-8)

Algorithm:

1. Obtain d

i

from the output of “N-Point Inverse DFT” processor

340

2. Set counter η=0. If ψ(d

i

)<C, then algorithm is DONE. Otherwise, proceed to step 3.

3. Pre-compute the following factors for 1≦l≦P: Y

l

=e

j2πJ(l)/N

, where J(l) is the l

th

member of J.

4. Find the largest α peaks and construct b

i

from these, using

c_{i} (k) = {\begin{matrix} d_{i} (k) & if sample k is not one of the a peaks \\ C \arg (d_{i} (k)) & if sample k is one of the a peaks \end{matrix}

and

b

i

(

k

)=

d

i

(

k

)−

c

i

(

k

)

5. Let the corresponding indices of the α largest peaks be k

1

, k

2

, . . . , k

a

. Compute the following update kernels for 1≦l≦P:

β

l

=Y

l

−k

1

bk

1

+Y

l

−k

2

b

k

2

+ . . . +Y

l

−k

ab

k

a

6. Update the vector d

i

by the following for 0≦k≦N−1:

d_{i} (k) = {[d_{i} (k)]}_{previous iteration} - \frac{2 μ}{N^{2}} \sum_{l = 1}^{P} Y_{l}^{k} B_{l}

7. Set counter η=η+1. If ψ(d

i

)<C or η=η

0

, then algorithm is DONE. Otherwise, return to step 4.

DONE: stop processing current block of data; use d

i

as the output of PAR processor

350

on paths

351

, . . . ,

352

.

1.5) Cyclic Prefix Inserter

360

As alluded to in the OFDM Principles Section above, deployment of the DFT results in each subchannel possessing a spectrum of non-negligible sidelobes. The sidelobes are mitigated by extending the data frame from N to N+V symbols by a cyclic prefix generated, for instance, by the appending the last V samples from the output of processor

340

to the symbols produced by PAR processor

350

. For example, for the parameters used as exemplary, V=8 is a suitable choice. Thus, if N=72, and these seventy-two points appearing on paths

351

, . . . ,

352

are labeled

1

,

2

, . . . , V+1, V+1, V+2, . . . , N, then paths

361

, . . . ,

362

convey the symbols V+1, V+2, . . . , N. The input to converter

370

thus has eighty-nine symbols (identified by the labels

1

, . . . , N+V) ordered as V+1, V+2, . . . , N, 1, 2, . . . V, V+1, V+2, . . . , N.

2.) Receiver

400

Before presenting the arrangement of elements composing an illustrative embodiment of receiver

400

, it is instructive to first present an overview discussion of the operational principles of receiver

400

, and the reason why certain processing capabilities are required in receiver

400

.

The transmission channel or wireless path between transmitter

300

of FIG.

3

and receiver 400 of

FIG. 10

distorts the transmit signal, impairs the orthogonality of the subchannels, and introduces interchannel/intersymbol interference. An L-branch (e.g., L=2) diversity combining technique is exploited to improve the error-rate performance of the MPACS receiver

400

. There are numerous well-known diversity combining techniques that offer a tradeoff between performance and complexity. For instance, maximal ratio combining yields the largest possible improvement that any diversity systems can produce over fading channels. Antenna diversity combining may also be implemented to further improve performance.

After signal filtering at the front-end of receiver

400

, and removal of the cyclic prefix as discussed in more detail shortly, each of the outputs of L diversity branches is converted to parallel format by a serial-to-parallel converter and then processed by an N-point DFT. The frequency domain PAR symbols and pilot symbols (if any) are removed before DFT outputs from each of the L branches are combined, or one of the L branches is selected. The sequence of K symbols are then demodulated, decoded if necessary, and then converted from a parallel incoming to a serial outgoing bit stream. The pilot symbols are used for carrier recovery and coherent detection. A simple look-up table technique is used for demodulation; the table is essentially a replicated version of the table of modulator

320

in transmitter 300. In addition, a hybrid symbol timing and carrier frequency offset estimation technique is used in receiver

400

.

A high-level block diagram of receiver

400

in accordance with the present invention is shown in FIG.

10

. Receiver

400

is configured for the general case of detecting the over-the-air transmission from transmitter

300

via plurality of processing paths for “diversity reception”, that is, each receiver

400

may be arranged with a plurality of receiving paths

401

, . . . ,

402

(e.g., receiving antennas) to detect the propagating RF signal from transmitter

300

. At the front-end of receiver

400

, a plurality of essentially identical parallel paths process the incoming signal detected by each corresponding antenna. The results of the processing by the parallel paths are then combined in down-stream circuitry to produce the “best” received signal, as will be discussed in more detail shortly.

A representative one of the front-end processing paths, as associated with antenna

401

, includes: (a) RF processor

405

, responsive to antenna

401

, for generally effecting the inverse of the functions performed by RF processor

390

of transmitter

300

(of particular relevance is the existence of a receiver local oscillator that nominally operates at the frequency of the transmitter local oscillator; however, to ensure proper reception of the symbols emitted by transmitter

300

, techniques to control the receiver local oscillator are necessitated in the receiver, as set forth in detail shortly); (b) receiving filter

410

, responsive to RF processor

405

, for filtering the signal received from processor

405

over path

407

; (c) serial-to-parallel converter

420

, responsive to the serial stream of bits on path

412

from filter

410

, for converting such serial stream to N+V parallel paths conveying N+V symbols, respectively, which are the received counterparts to N+V symbols at the input to converter

370

of transmitter

300

; (d) symbol timing and Carrier Frequency Offset (CFO) estimator

415

, responsive to the signal on path

412

, for controlling symbol timing and CFO estimation in processor

405

as well as in converter

420

, as discussed in more detail shortly; (e) cyclic prefix remover

425

, responsive to converter

420

, for removing the prefix symbols introduced by inserter

360

of transmitter

300

to thereby produce N output symbols from the N+V incoming symbols; (f) N-point DFT transform processor

430

, responsive to remover

425

, for computing the Discrete Fourier Transform on the N incoming symbols; (g) frequency domain equalizer (FEQ) and special symbol remover

440

, responsive to processor

430

, the function of which is discussed in more detail below; and (h) channel and SINR estimator

435

, also responsive to processor

430

, the function of which is discussed in more detail below.

A second diversity path that has elements and operates in essentially the same manner as the above-described path includes the following elements identified by their associated reference numerals along with the reference numerals of the corresponding elements from the first path, in parenthesis: RF processor

406

(RF processor

405

); receiver filter

411

(filter

410

); symbol timer and CFO estimator

416

(estimator

415

); converter

421

(converter

420

); cyclic prefix remover

426

(remover

425

); N-point DFT processor

431

(processor

430

); estimator

436

(estimator

435

); and FEQ and special symbol remover

441

(remover

440

).

At this point in the processing by receiver

400

, each of the L diversity branches outputs K symbols, as depicted by the K symbols on paths

442

, . . . ,

443

for the first diversity path, and paths

445

, . . . ,

446

for the branch L or the L

th

diversity path. The L parallel sets of K symbols serve as an input to diversity selector

450

; in addition, selector

450

has as inputs the outputs of channel estimators

435

, . . . ,

436

on leads

437

. . . ,

438

, respectively. Diversity selector

450

selects one of the branches as providing the “best” signal for detection and provides the K symbols for the selected branch on paths

451

, . . . ,

452

. The “best” signal may utilize, for instance, signal-to-noise ratio computations. The chosen symbol set of K symbols then serves as input to demodulator

460

, which performs the obverse functions of modulator

320

of FIG.

3

. Demodulation is effected by a table look-up procedure wherein the real and imaginary coordinates of each complex symbol is located in the table and the code associated with each such complex symbol is outputted from demodulator 460 as a parallel stream of bits on K paths

461

, . . . ,

462

. The final step in the processing is to convert the K parallel streams into a serial output stream; this is accomplished by parallel-to-serial converter

470

, and the output bit stream on output path 403 is the detected counterpart to input bit stream

310

of transmitter

300

.

Details of the Receiver Illustrative Embodiment

Whereas certain of the elements of receiver

400

are typically realized conventionally, such as RF processors

405

and

406

, filters

410

and

411

, converters

420

and

421

, cyclic prefix removers

425

and

426

, DFT processors

430

and

431

, and converter

470

, additional description with respect to certain of the other elements is in order to fully elucidate the illustrative embodiment of the present invention; further details pertaining to these latter elements are now presented.

2.1) Symbol Timing and CFO Estimation by Estimator

415

(see

FIG. 11

)

Symbol timing recovery and fine CFO estimation rely on the correlation properties of the cyclic prefix. Coarse CFO estimation makes use of known pilot symbols.

Definitions:

1. 1−χ is a “forgetting factor”, weighting contributions to the symbol timing and fine CFO estimates from previous processing blocks of

FIG. 11

2. ρ is a correlation coefficient

3. Ξ is the number of OFDM blocks (including the present one) to be included in the symbol timing and fine CFO recovery.

4. {circumflex over (ε)}

coarse,i

and {circumflex over (Δ)}

i

are integer-valued offset estimates, of coarse CFO and of symbol timing, respectively. {circumflex over (ε)}

fine,i

is a fractional (CFO) offset estimate. {circumflex over (ε)}

coarse,i

and {circumflex over (ε)}

fine,i

are normalized to the subcarrier bandwidth.

5. B

s

is the (known) subcarrier bandwidth.

6. δ

f,i

is the estimated CFO, where δ

f,i

=B

S

({circumflex over (ε)}

coarse,i

+{circumflex over (ε)}

fine,i

)

7. W is the number of pilot symbols, and V is the number of (regular, either data or PAR) subchannels between consecutive pilots.

8. s(k) are the discrete-time samples using the recovered timing offset and fine CFO offset, obtained from r(kT) as explained in the Algorithm description below.

Parameter Settings:

Example parameter settings (with suggested range in parentheses):

1. χ=0.7(0.5-0.9)

2.

ρ = \frac{SNR}{SNR + 1}

where SNR is the average signal-to-noise ration of the subchannels in that branch, obtained from channel estimator

435

, . . . ,

436

, depending upon the particular diversity branch

3. Ξ=5 (1-10)

4. W=4, V=17

Algorithm:

PART ONE: Timing Offset and Fine CFO Estimation and Recovery

1. Compute

Λ_{i, 1} (Δ, ε) = \sum_{k = Δ}^{Δ + v - 1} r (kT) r^{*} ((k + N) T),

as computed by processing elements

1105

,

1110

, and

1115

of

FIG. 11

; also, compute

Λ_{i, 2} (Δ, ε) = \sum_{k = Δ}^{Δ + v - 1} {&LeftBracketingBar; r (kT) &RightBracketingBar;}^{2} + {&LeftBracketingBar; r ((k + N) T) &RightBracketingBar;}^{2},

as computed by processing elements

1110

,

1145

,

1150

and

1152

of FIG.

11

. The ‘i’ in the subscript is the block index.

2. The timing estimate (for the ith block) is

{\overset{⋒}{Δ}}_{i} = \arg \max_{Δ} (&LeftBracketingBar; \sum_{ξ = 0}^{Ξ} Λ_{i - ξ, 1} (Δ, ε) {(1 - χ)}^{ξ} &RightBracketingBar; - \frac{ρ}{2} \sum_{ξ = 0}^{Ξ} Λ_{i - ξ, 2} (Δ, ε) {(1 - χ)}^{ξ}),

as further obtained by the processing effected by elements

1155

,

1156

,

1160

,

1120

,

1125

,

1130

,

1140

,

1141

, and

1165

3. Using the timing estimate {circumflex over (Δ)}

i

, the fine CFO estimate

{\hat{ε}}_{fine, i} = - \frac{1}{2 π} ∠ (\sum_{ξ = 0}^{Ξ} Λ_{i - ξ, 1} {\overset{⋒}{(Δ}}_{i}, ε) {(1 - χ)}^{ξ}),

as further obtained by the processing effected by elements

1120

,

1125

,

1130

, and

1135

.

PART TWO: Coarse CFO Estimation and Recovery (Elements

1170, 1175, 1180, 1166, 11851).

1. s(k)=r((k+{circumflex over (Δ)}

i

)T)e

−j2π{circumflex over (ε)}

fine,i

(k+{circumflex over (Δ)}

l

)/N

—delayed by timing offset estimate, then rotated −2π{circumflex over (ε)}

fine,i

k/N

2. For a reasonable range of expected coarse CFO (e.g. +/−5 subchannel spacings, ε=−5,−4, . . . , 4,5), compute the metric

F (ε) = \sum_{n = 0}^{W - 1} {&LeftBracketingBar; \sum_{k_{2} = 0}^{W - 1} [(\sum_{k_{1} = 0}^{V - 1} s (k_{1} W + k_{2}) ⅇ^{- j2 π ε n_{1} / V}) ⅇ^{- j2 π ε k_{2} / N}] ⅇ^{- j2 π {nk}_{2} / W} &RightBracketingBar;}^{2} .

(The coarse CFO estimator can estimate coarse CFO only up to ±V/2. Since V=17 in the example, the maximum range is ±8. However, the design specification for receiver

400

requires that a local oscillator is sufficiently precise that the magnitude of the CFO is upper-bounded by 4, allowing the search to be limited to a range of ±5.)

3. Pick ε

coarse,i

=arg

ε

min F(ε)

2.2) Channel and SINR Estimation by Estimator

435

(see

FIG. 12

)

Definitions:

1. {overscore (H)}

n

is the channel estimate for subcarrier n based only on the current block, and {overscore (H)} is the length-N vector representing the channel estimate for all N subcarriers, based only on the current block. Ĥ is the corresponding vector representing the channel estimate for all N subcarriers, based on a weighted average of current and preceding blocks.

2. 1−ε is a forgetting factor controlling the influence of the previous blocks on the current estimates

Parameter Settings:

1. β=0.7 (0.5-0.9)

Algorithm:

1. Pilot symbols are extracted by element

1205

from N paths

1201

, . . . ,

1202

. Since the pilot symbols have been scaled and rotated by the channel, a complex-number division via element

1210

by the known original pilot symbols is used to estimate the channel for the pilot subchannels.

2. “Frequency Interpolator”

1215

shall use standard interpolation techniques to interpolate the estimates to a set of estimates for all subchannels.

3. Estimates from previous blocks, via blocks

1220

and

1225

, are factored in with a forgetting factor, to provide the final estimate.

SINR (Signal-to-Interference Noise Ratio) Estimation

Definitions:

1. {overscore (E)} is the length-N vector representing the SINR estimate for all N-subcarriers, based on the current received pilots and {overscore (H)}. {overscore (E)} is the corresponding vector based on a weighted average of current and preceding blocks.

2. 1−β is a forgetting factor controlling the influence of the previous blocks on the current estimates

Algorithm:

1. Starting with the channel estimates, “Compute W Error Signals” processor

123

computes the W error signals as follows: E

n

=R

n

−Ĥ

n

A

n

, where R

n

is the symbol received on the nth subcarrier (n

th

output of the DFI), and A

n

is the corresponding known pilot symbol as provided by element

1255

.

2. “Error Interpolator” element

1235

uses standard interpolation techniques to interpolate the error estimate for all subchannels with inputs from element

1230

.

3. “Estimate SINR” element

1250

computes SI{circumflex over (N)}R

n

=|H

n

|

2

/|E

n

|

2

in conjunction with delay block

1245

and summer

1240

It is noted that the foregoing technique for SINR is merely exemplary; other techniques are also possible, and is an implementation detail not within the purview of the present invention.

2.3) FEO (Frequency-Domain Equalizer) and Special Symbol Remover

440

(see

FIGS. 12 and 13

)

Definitions:

1. H

n

is the channel response for the nth subchannel (estimated by “Channel and SINR Estimation” estimator

435

of

FIG. 12

, as appearing on lead

1203

—which is equivalent to lead

431

for Branch

1

of

FIG. 10

)

2. SINR

n

is the SINR of the n

th

subchannel (estimated by “Channel and SINR Estimation” estimator

435

of

FIG. 12

, as appearing on lead

1204

—which is equivalent to lead

437

of Branch

1

of

FIG. 10

)

3. G

n

is the one-tap equalizer for the nth subchannel, e.g. the one-tap MMSE equalizer defined by

G_{n} = \frac{H_{n}^{*}}{{&LeftBracketingBar; H_{n} &RightBracketingBar;}^{2} + 1 / {SINR}_{n}}

4. {circumflex over (D)}

n,l

refers to the equalized output of this functional block, where l is used in the subscript to explicitly refer to the branch number (between

1

and L) in preparation for “Diversity Selection” by selector

450

Algorithm:

1. Remove PAR and pilot symbols by processing effected by element

1304

and elements

1305

, respectively

2. Equalize the remaining subchannels using G

n

in FEQ element

1310

.

2.4) Diversity Selector

450

Branch diversity can be an effective means of reducing the degrading effects of interference, such as inter-channel interference. There are a variety of diversity techniques that offer a trade-off between performance and complexity. A discussion of representative techniques is presented in an article entitled “Comparison of Diversity Combining Techniques for Rayleigh-Fading Channels” by T. Eng et al., as published in the IEEE Transactions of Communications, Vol. 44, No. 9, pp. 1117-1129, September, 1996. One exemplary technique for the illustrative embodiment is as follows:

Definitions:

1. {circumflex over (D)}

n

refers to the output of selector

450

.

2. {circumflex over (D)}

n,l

refers to the input to selector

450

for the n

th

subchannel from the L

th

branch

Algorithm:

1. Compare SINR

1

for all the L branches and pick the branch with the largest, e.g. branch l′.

2. Set {circumflex over (D)}

1

={circumflex over (D)}

1,l′

3. Repeat for subchannels

2

through K

2.5) Demodulation by Demodulator

460

The same look-up table generated off-line and stored in modulator

320

of

FIG. 3

is replicated and stored in demodulator

460

. Thus, in one case, the appropriate constellation depicted in

FIG. 4C

,

5

,

6

, or

7

, depending upon the M-QAM selected, is stored in demodulator

460

. In the alternative for a variable rate case, all constellations of interest would be stored in demodulator

460

and the particular constellation addressed would be accomplished via a QAM-select signal. Each of these cases was described with respect to

FIGS. 8A and 8B

for modulation, and a similar description applies now to the case of demodulation.

For demodulation, consider the constellation points being labeled as P

4k+n

for k=0, 1, . . . , M/4, and n=0, 1, 2, 3. As an example of this notation, if k=0, then the constellation points under consideration are those in the first quadrant of

FIGS. 4C

,

5

,

6

or

7

, where the x and y axis define Cartesian axes. The constellation points P

4k+n

are used in the demodulator search algorithm to find the complex symbol in the constellation that is closest the to demodulated symbol that appears on paths

451

, . . . ,

452

; each such symbol is referred to as the “detected” complex symbol. Moreover, each detected complex symbol is treated independently of any other detected complex symbol to effect demodulation, but the search algorithm to convert each detected complex symbol to a corresponding stream of output bits is the same, as now described:

(a) first, rotate the detected complex symbol, denoted as c

k

→(x

k

, y

k

), into the first quadrant and search M/4 locations for the constellation point that yields the “minimum distance” computation defined as follows:

min{(

x−|x

k

|)

2

+(

y−|y

k

|)

2

},

where P

4k

→(x,y) for k=0, 1, 2, . . . , M/4. Suppose the k-value yielding the minimum is denoted m.

(b) next, search over the four quadrants via the four locations P

4m+n

for n=0, 1, 2, 3 and locate the constellation point that gives the minimum distance between c

k

→(x

k

, y

k

) and the complex symbols at the four locations.

(c) finally, map the P

4m+n

location to the (Gray) coded bits corresponding to the complex symbol closest to the detected complex symbol.

As an example of this algorithm, suppose the detected complex symbol arriving on path

451

is (−0.5, 0.75) and 32-QAM is the given modulation. From

FIG. 7

, using step (a) of the demodulation algorithm and the coordinates of the detected symbol, it is evident that the point P

0

is closest to the detected symbol based upon the minimum distance computation. Now, according to step (b), besides P

0

, only the points P

1

, P

2

, P

3

need to be investigated to determine which point yields the minimum distance. Again, it is clear that point P

1

yields the minimum distance. Finally, according to step (c), the Gray code associated with the located complex symbol is (

10110

), which becomes the stream of output bits emitted on path

461

for the detected complex symbol arriving on path

451

.

It is instructive to compare the demodulation algorithm with the straightforward or “brute force” manner of locating the constellation point nearest to the detected complex symbol. With the straightforward approach, thirty-two minimum distance type computations would be needed for 32-QAM. On the other hand, using the demodulation algorithm described above, only eight computations are needed in step (a), and only four additional computations are required to complete step (b), for a total of twelve computations. This reduced number of computations (

12

versus

32

) speeds up the demodulation process. As the level of M-QAM increases, the efficiency becomes even more apparent.

The block diagram of

FIG. 14

illustrates an arrangement of elements to effect the demodulation process. The detected complex symbol arrives on path

1401

, with is representative of any of the paths

451

, . . . ,

452

. Comparator and search logic circuit

1410

carries out steps (a)-(c) above of the demodulation algorithm. Lead

1411

from circuit

1410

is an address lead for accessing constellation points within look-up table

1420

, which is 32-QAM for exemplary purposes. Based upon the address information on lead

1411

, an output symbol corresponding to the address is emitted on path

1403

and serves as an input to circuit

1410

. The minimum distance computations are carried out on each symbol provided by path

1403

. As a result of the computations completed by circuit

1410

, the last address provided on lead

1411

is the location of the constellation point nearest the detected symbol arriving on path

1401

; this last address causes device

1420

to emit the stream of output bits corresponding to the Gray code associated with the constellation point nearest the detected symbol on path

1402

, which is representative of any path

461

, . . . ,

462

.

2.6) Parallel-to-Serial Converter

470

Converter 470 is depicted as a simple parallel-to-serial converter. However, if converter

310

of

FIG. 3

employs encoding techniques, such as forward error correction (FEC) encoding, it is necessary to augment converter

470

with FEC decoding capability, commensurate with the encoding of converter

310

, prior to the conversion of parallel streams on K paths

461

, . . . ,

462

to serial output stream

403

.

Flow Diagrams

Transmitter

300

The flow diagram of

FIG. 15

depicts the operational steps carried out by transmitter

300

in its most generic realization. In particular, the processing effected by transmitter

300

is carried out by processing blocks

1510

-

1580

as follows:

1510

: convert input bits into a set of unique symbols wherein each symbol represents a unique plurality of bits;

1520

: modulate each of the symbols in the set to produce a corresponding set of complex symbols;

1530

: augment the set of complex symbols with pilot symbols and energy-adjusting symbols (e.g., PAR symbols), as needed, to produce an augmented set of complex symbols;

1540

: compute the inverse Discrete Fourier Transform of the augmented set to produce a transformed set of symbols;

1550

: modify the transformed set of symbols as determined by the energy in the transformed set;

1560

: augment the transformed set with cyclic prefix symbols determined from the transformed set to produce a set of output symbols;

1570

: convert the set of output symbols to a serial stream of output bits and filter the output bits to suppress intersymbol interference; and

1580

: propagate the output bits as an RF signal over the wireless channel.

Receiver

400

The flow diagram of

FIG. 16

depicts the operational steps carried out by receiver

400

in its most generic realization. In particular, the processing effected by receiver

400

is carried out by processing blocks

1610

-

1680

as follows, wherein it is assumed there are no other diversity branches:

1610

: process a radio frequency signal conveying complex symbols to produce a received signal;

1620

: recover carrier frequency synchronization information and complex symbol timing information from the received signal;

1630

: shift the received signal to a baseband signal using the carrier synchronization information;

1640

: sample the baseband signal using the recovered timing information to produce sampled complex symbols;

1650

: remove cyclic prefix symbols from the sampled complex symbols to produce a reduced set of complex symbols;

1660

: compute the Discrete Fourier Transform of the reduced set of complex symbols to produce a set of transformed symbols;

1670

: frequency equalization and remove special symbols from the transformed symbols to produce a set of detected complex symbols; and

1680

: demodulate the detected complex symbols to generate an output stream of bits.

Flow Diagram to Generate the Look-Table Utilized for Modulation and Demodulation

The flow diagram of

FIG. 17

depicts the operational steps carried out off-line to produce the look-up table for modulation and demodulation. The steps have been set forth in detail in a foregoing section, but are reiterated here in correspondence to the processing blocks of FIG.

17

.

1710

: define the set of points in the first quadrant of the M-QAM constellation as S

M

, wherein the x and y coordinates of the points can take on only the values x=±1, ±3, ±5, . . . and y=±1, ±3, ±5, . . .

1720

: start with the 4-QAM constellation and label the point in S

4

having x and y coordinates as P

0

, i.e., P

0

→(x,y).

1730

: move in a counter-clockwise direction and label the other three points to complete the 4-QAM constellation as P

1

→(−x,y), P

2

→(−x,−y), and P

3

→(X,−y). The depiction of

FIG. 4A

illustrates the location of these constellation points.

1740

: assign codes to the 4-QAM constellation based on a Gray code assignment such that P

0

has associated Gray code (

01

), P

1

has associated Gray code (

00

), P

2

is assigned (

10

), and P

3

is assigned (

11

).

1750

: labels for the 8-QAM and higher constellation points are determined in an iterative manner. Consider first the next point P

k

→(x,y) in S

M

not already in S

M−1

.

(a) find the lowest energy point x

2

+y

2

not in S

M−1

.

(b) if two or more points have equal energy, then

(i) if the sums (x+y) for the points are equal, choose the point with the smallest y value

(ii) if the sums are different, choose the point with the smallest max {x,y}

(c) choose the three points associated with P

k

as P

k+1

→(−x,y), P

k+2

→(−x,−y), and P

k+3

→(x,−y)

1760

: repeat processing step

1750

until there are M/4 points in S

M

1770

: assign codes to the constellation points, using Gray codes when feasible, according to the following rules:

(a) if the M-QAM constellation is square, and 2

L

=M, then assign Gray codes to the constellation points based on an L/2 Gray code assignment to the decomposed in-phase and quadrature components

(b) if the M-QAM constellation is not square, then decompose it into the least number of contiguous square blocks, assign Gray codes to them based upon an int[L/2]-bit and {L−int[L/2]}-bit Gray code assignment. (Note that int[r] means integer division). If there are points in the constellation where Gray codes cannot be assigned, assign other codes that minimize the distance with neighboring points.

Process of Modulation Carried Out by Modulator

320

The flow diagram of

FIG. 18

depicts the operational steps effected for modulation in transmitter

300

. The steps have been set forth in detail in a foregoing section, but are reiterated here in correspondence to the processing blocks of FIG.

18

.

1810

: the input for processing is composed of a set of input bits;

1820

: access a minimal energy constellation of points expressed as pairs of in-phase and quadrature components representative of a given modulation technique wherein the number of points equals the number of combinations of input bits and each point is assigned a unique one of the combinations (the constellation of points is determined in a manner commensurate with the flow diagram of FIG.

17

);

1830

: select the in-phase and quadrature components of one of the points corresponding to the input bits as a complex output symbol; and

1840

: the selected complex symbol is the output symbol representative of the set of input bits.

Process of Demodulation Carried Out by Demodulator

460

The flow diagram of

FIG. 19

depicts the operational steps effected for demodulation in receive

400

. The steps have been set forth in detail in a foregoing section, but are reiterated here in correspondence to the processing blocks of FIG.

19

.

1910

: the input to the process is a detected complex symbol;

1920

: rotate the detected complex symbol, denoted as c

k

→(x

k

, y

k

), into the first quadrant and search M/4 locations for the constellation point that yields the “minimum distance” computation defined as follows:

min{(

x−|x

k

|)

2

+(

y−|y

k

|)

2

},

where P

4k

→(x,y) for k=0, 1, 2, . . . , M/4. Suppose the k-value yielding the minimum is denoted m.

1930

: search over the four quadrants via the four locations P

4m+n

for n=0, 1, 2, 3 and locate the constellation point that gives the minimum distance between c

k

→(x

k

,y

k

) and the complex symbols at the four locations.

1940

: map the P

4m+n

location to the (Gray) coded bits corresponding to the complex symbol closest to the detected complex symbol.

1950

: the output of the process is a stream of bits corresponding to the code assigned to P

4m+n

in the look-up table.

Although various embodiments which incorporate the teachings of the present invention have been shown and described in detail herein, those skilled in the art can readily devise many other varied embodiments that still incorporate these teachings.

Number	Name	Date	Kind
5329552	de Couasnon et al.	Jul 1994	A
5428643	Razzell	Jun 1995	A
5548582	Brajal et al.	Aug 1996	A
6175550	van Nee	Jan 2001	B1
6385190	Malkamaki et al.	May 2002	B1
6449246	Barton et al.	Sep 2002	B1
6487252	Kleider et al.	Nov 2002	B1

Multicarrier personal access communication system

Information

Patent Number

Date Filed

Date Issued

Inventors

Original Assignees

Examiners

Agents

CPC

US Classifications

Field of Search

US

International Classifications

Abstract

Description

Claims

CROSS-REFERENCE TO RELATED APPLICATION

US Referenced Citations (7)

Non-Patent Literature Citations (2)

Provisional Applications (1)