Multiplexer with low parasitic capacitance effects

Abstract

A differential multiplexer includes a plurality of multiplexing circuits. Each multiplexing circuit inputs a corresponding differential input signal including a positive input signal and a negative input signal, and outputs positive and negative output signals. Each multiplexing circuit includes first, second, third and fourth transistors. The first and second transistors input the positive input signal. The third and fourth transistors input the negative input signal. Outputs of the first and third transistors are connected to the positive output signal. Outputs of the second and fourth transistors are connected to the negative output signal. The positive and negative output signals are controlled using gate voltages on the first and fourth transistors. The second and third transistors are turned off when the differential multiplexer is in use. The transistors are cross-coupled to make leakage between the positive and negative input signals common mode in the positive and negative output signals.

Description

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to multiplexers, and, more particularly, to multiplexers with low cross-talk between signals.

2. Related Art

A subranging analog to digital converter (ADC) architecture is suitable for implementing high-performance ADC's (i.e. high speed, low power, low area, high resolution). FIG. 1 shows a generic two-step subranging architecture, comprising a reference ladder 104, a coarse ADC 102, a switching matrix 103, a fine ADC 105, coarse comparators 107, fine comparators 108 and an encoder 106. In most cases, a track-and-hold 101 is used in front of the ADC. In this architecture, an input voltage is first quantized by the coarse ADC 102. The coarse ADC 102 compares the input voltage against all the reference voltages, or against a subset of the reference voltages that is uniformly distributed across the whole range of reference voltages. Based on a coarse quantization, the switching matrix 103 connects the fine ADC 105 to a subset of the reference voltages (called a “subrange”) that is centered around the input signal voltage.

Modem flash, folding and subranging analog to digital converters (ADC's) often use averaging techniques for reducing offset and noise of amplifiers used in the ADC. One aspect of averaging is the topology that is used to accomplish averaging, i.e., which amplifier outputs in which arrays of amplifiers are averaged together.

In general, flash, folding and subranging ADC's use cascades of distributed amplifiers to amplify the residue signals before they are applied to the comparators. These residue signals are obtained by subtracting different DC reference voltages from an input signal V_in. The DC reference voltages are generated by the resistive ladder (reference ladder) 104 biased at a certain DC current.

High-resolution ADC's often use auto-zero techniques, also called offset compensation techniques, to suppress amplifier offset voltages. In general, autozeroing requires two clock phases (φ₁ and φ₂). During the auto-zero phase, the amplifier offset is stored on one or more capacitors, and during the amplify phase, the amplifier is used for the actual signal amplification.

Two different auto-zero techniques can be distinguished, which are illustrated in FIGS. 2 and 3. The technique shown in FIG. 2 connects an amplifier 201 in a unity feedback mode during the auto-zero clock phase φ₁. As a result, a large part of the amplifier 201 input offset voltage is stored on input capacitors C1a, C1b. The remaining offset is stored on output capacitors C2a, C2b if available.

The second technique, shown in FIG. 3, shorts the amplifier 201 inputs during the auto-zero phase φ₁ and connects them to a DC bias voltage V_res. Here, the amplifier 201 output offset voltage is stored on the output capacitors C2a, C2b. Many ADC architectures use a cascade of several (auto-zero) amplifiers to amplify the input signal prior to applying to the comparators 107, 108. In general, flash, folding and subranging ADC's use arrays of cascaded amplifiers, and averaging and interpolation techniques are used to improve performance.

Unfortunately, the performance of cascaded arrays of amplifiers degrades significantly at high clock and input signal frequencies. The cause of this degradation is illustrated in FIG. 4 when the reset technique shown in FIG. 3 is used, and where R_SW is shown as a circuit element, and the current flow I_C is explicitly shown.

When the amplifier 201 is in the auto-zero phase φ₁, the input capacitors C1a, C1b are charged to the voltage V_sample that is provided by the track-and-hold amplifier 101. As a result, a current I_C will flow through the input capacitors C1a, C1b and an input switch (not shown). Due to the finite on-resistance R_SW of the input switch (see FIG. 4), an input voltage is generated, which will settle exponentially towards zero. This input voltage is amplified by the amplifier 201 and results in an output voltage that also slowly settles towards zero (assuming the amplifier 201 has zero offset).

Essentially, the auto-zero amplifier 201 is in a “reset” mode one-half the time, and in an “amplify” mode the other one-half the time. When in reset mode, the capacitors C1a, C1b are charged to the track-and-hold 101 voltage, and the current I_C flows through the capacitors C1a, C1b and the reset switches, so as to charge the capacitors C1a, C1b.

When the ADC has to run at high sampling rates, there is not enough time for the amplifier 201 output voltage to settle completely to zero during the reset phase. As a result, an error voltage is sampled at the output capacitors C2a, C2b that is dependent on the voltage V_sample. This translates into non-linearity of the ADC, and often causes inter-symbol interference (ISI).

The problem of ISI occurs in most, if not all, ADC architectures and various approaches exist for attacking the problem. The most straightforward approach is to decrease the settling time constants. However, the resulting increase in power consumption is a major disadvantage.

Another approach is to increase the time allowed for settling, by using interleaved ADC architectures. However, this increases required layout area. Furthermore, mismatches between the interleaved channels cause spurious tones. The ISI errors can also be decreased by resetting all cascaded amplifiers during the same clock phase. Unfortunately, this is not optimal for high speed operation either.

FIG. 9 illustrates a simple conventional multiplexer. As shown in FIG. 9, two inputs, V1 and V2 are fed into two transistors, or switches, 902A, 902B, respectively. Depending on which of the switches 902A, 902B is on or off, the output voltage Vout is switched between V1 and V2. Note that each of the transistors 902A, 902B has parasitic capacitance Cp. For clarity, only one of the transistors is shown with the parasitic capacitance, but it will be appreciated that all such transistors have some parasitic capacitance. The parasitic capacitance Cp causes signal feedthrough (also known as “leakage,” or “crosstalk”) of V2 to V_OUT when the transistor 902B is supposed to be off. Similarly, when transistor 902A is off, its parasitic capacitance causes signal feedthrough to Vout. This effect is undesirable in multiplexers.

FIG. 10 illustrates a conventional approach to decreasing signal feedthrough. This approach relies on the addition of two switches (transistors) to ground, 906A and 906B, and additional switches 904A, 904B, connected as shown in FIG. 10. The switches 906A, 906B help reduce the signal feedthrough. However, this approach has two problems. It requires twice as many transistors for the same on-resistance Ron. Also, there is four times as much total gate capacitance in the overall circuit, which is important if the switches are clocked.

SUMMARY OF THE INVENTION

The present invention is directed to a multiplexer with low parasitic capacitance effects that substantially obviates one or more of the problems and disadvantages of the related art.

In one aspect of the invention there is provided a differential multiplexer including a plurality of multiplexing circuits. Each multiplexing circuit inputs a corresponding differential input signal including a positive input signal and a negative input signal, and outputs positive and negative output signals. Each multiplexing circuit includes first, second, third and fourth transistors. The first and second transistors input the positive input signal. The third and fourth transistors input the negative input signal. Outputs of the first and third transistors are connected to the positive output signal. Outputs of the second and fourth transistors are connected to the negative output signal. The positive and negative output signals are controlled using gate voltages on the first and fourth transistors. The second and third transistors are turned off when the differential multiplexer is in use.

In another aspect of the invention, there is provided a differential multiplexer including a plurality of multiplexing circuits. Each multiplexing circuit inputs a corresponding differential input signal including a positive input signal and a negative input signal, and outputs positive and negative output signals. Each multiplexing circuit includes a plurality of transistors cross-coupled to make leakage between the positive and negative input signals common mode in the positive and negative output signals.

Additional features and advantages of the invention will be set forth in the description which follows, and in part will be apparent from the description, or may be learned by practice of the invention. The advantages of the invention will be realized and attained by the structure particularly pointed out in the written description and claims hereof as well as the appended drawings.

It is to be understood that both the foregoing general description and the following detailed description are exemplary and explanatory and are intended to provide further explanation of the invention as claimed.

BRIEF DESCRIPTION OF THE FIGS.

The accompanying drawings, which are included to provide a further understanding of the invention and are incorporated in and constitute a part of this specification, illustrate embodiments of the invention and together with the description serve to explain the principles of the invention. In the drawings:

FIG. 1 illustrates a conventional averaging topology.

FIGS. 2 and 3 illustrate conventional amplifier topologies with reset switches.

FIG. 4 illustrates a conventional amplifier topology and the source of the inter-symbol interference problem.

FIG. 5 illustrates a source of inter-symbol interference in greater detail.

FIG. 6 illustrates one embodiment of the present invention.

FIG. 7 illustrates another embodiment of the present invention.

FIG. 8 illustrates a reduction in inter-symbol interference using the present invention.

FIG. 9 illustrates a conventional multiplexer.

FIG. 10 illustrates a conventional approach to decreasing signal feedthrough.

FIG. 11 illustrates how the circuit of FIG. 6 can be used as a differential multiplexer.

FIG. 12 illustrates how the circuit FIG. 11 can be adapted to a 4:1 multiplexer.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Reference will now be made in detail to the preferred embodiments of the present invention, examples of which are illustrated in the accompanying drawings.

Recently, a technique to address the nonlinearity was published by Miyazaki et al., “A 16 mW 30 M Sample/s pipelined A/D converter using a pseudo-differential architecture,” ISSCC Digest of Tech. Papers, pp. 174–175 (2002), see particularly FIG. 10.5.2 therein. The technique applies only to amplifiers that use the auto-zero technique of FIG. 2.

In Miyazaki, four extra switches and two extra capacitors are required. The resulting circuit topology has a common-mode transfer function of “1” and a differential-mode transfer function of “0” during the reset clock phase.

However, an important disadvantage of the circuit shown in Miyazaki is that it requires twice the amount of capacitance. This has a serious impact on the ADC layout area. Furthermore, the capacitive loading of the track-and-hold 101 doubles, which significantly slows down the charging of the capacitors C1a, C1b (roughly by a factor of two).

FIG. 5 shows the rationale for the present invention. In FIG. 5, the track-and-hold amplifier 101 outputs a step function to the sampling capacitors C1a, C1b. Due to the finite resistance RSW, the pulse becomes a spike (i.e., it is effectively high-pass filtered) by the time it gets to the amplifier 201, which is the first amplifier in a cascade. The next set of capacitors C2a, C2b sees a “smeared-out” pulse, which, by the time it is amplified by the next amplifier in a cascade (amplifier 202), and charges the next stage capacitors C3a and C3b, becomes further “smeared-out”. The spike being transferred throughout the cascaded amplifiers causes inter-symbol interference.

The problem of ISI can be solved in a very elegant way by complementing the reset switches shown in FIG. 3 with some additional switches before the fine amplifiers of the fine ADC 105. The resulting circuit is shown in FIG. 6. The extra switches are contained in the dashed box 510 (a transfer matrix or transfer circuit). FIG. 7 shows a modification of the new circuit that works in a similar way.

The transfer circuit shown in the dashed box 510 has a transfer function of “1” for common-mode signals at all times, so that the common mode transfer function is H_CM(φ₁)=1, H_CM(φ₂)=1. However, the transfer function varies for differential signals depending on the clock phase (φ₁ or φ₂). More specifically, the transfer function for differential signals is H_DM(φ₁)=0, and H_DM(φ₂)=1. Hence, a differential voltage created across nodes 1 and 2 (due to the charging of the input capacitors C1a, C1b) is not transferred to input nodes 3 and 4 of the amplifier 201 during φ₁. Therefore, the output voltage of the amplifier 201 is not affected by V_sample in any way, reducing the occurrence of ISI. The input capacitors C1a, C1b subtract track-and-hold amplifier 101 voltage from a reference ladder 104 voltage.

The technique presented herein can find application in various types of ADC architectures that use auto-zero techniques for combating amplifier offsets.

FIG. 6 shows one embodiment of the present invention. φ₁ and φ₂ represent two phases of a clock, preferably non-overlapping phases. As shown in FIG. 6, the sampling voltage V_sample is differentially connected to two sampling capacitors C1a, and C1b, which are in turn connected to three switch transistors Ma, Mb and Mc. Gates of the switch transistors Ma, Mb, Mc are connected to φ₁, a drain of the transistor Ma is connected to V_res, and a source of the transistor Mc is connected to the reset voltage V_res. Between the amplifier 201 and the switch transistors Ma, Mb, Mc, the transfer matrix 510 comprises four transistors M1, M2, M3 and M4. Gates of the transistors M2 and M3 are connected to φ₁. Gates of the transistors M1 and M4 are connected to V_dd, the supply voltage. Sources of the transistors M1 and M2 are tied together and to the node 1, which is also connected to the sampling capacitor C1a. Sources of the transistors M3 and M4 are tied together and also connected to a node 2, which is also connected to the sampling capacitor C1b. Drains of the transistors M3 and M1 are tied together and to node 3, which is the “+” input of the amplifier 201. Drains of the transistors M2 and M4 are tied together and to node 4, which is also connected to the “−” input of the amplifier 201.

Thus, the circuit within the dashed box 510 may be referred to as a transfer matrix that has a property such that its differential mode transfer function H(φ₁)=0, H(φ₂)=1. This is different from a conventional approach, where the transfer function may be thought of as being H=1 for both φ₁ and φ₂.

It will be appreciated that while the overall transfer function of the transfer matrix 510 is H_DM(φ₁)=0, H_DM(φ₂)=1, H_CM(φ₁)=1, H_CM(φ₂)=1, this is primarily due to the switches M1–M4, which essentially pass the differential voltage of nodes 1 and 2 through to nodes 3 and 4 respectively, on φ₂. However, the gain factor need not be exactly 1, but may be some other value. The important thing is that it be substantially 0 on φ₂.

FIG. 7 represents another embodiment of the present invention. The elements of FIG. 7 correspond to the same-numbered elements of FIG. 6, however, the position of the transfer matrix 510 is before the three transistors Ma, Mb and Mc, rather than after. This results in lower noise operation, compared to the embodiment shown in FIG. 6. The embodiment shown in FIG. 6, however, generally allows for higher frequency operation, compared to the embodiment of FIG. 7.

Note that either PMOS or NMOS transistors may be used as switches in the present invention. Note further that given the use of the FET transistors as switches (rather than the amplifiers), the drain and the source function equivalently.

FIG. 8 illustrates the improvement in the signal due to the transfer matrix 510. Note that the transistors Ma, Mb, Mc and the transistors of the transfer matrix M1–M4, are PMOS transistors, with the negative supply Vss used instead of the positive supply V_dd. As may be seen from FIG. 8, the amount of spike seen by the amplifier 201 after a step function outputted from the track-and-hold 101 is dramatically decreased due to the transfer function of the transfer matrix 510. φ_1e in FIG. 8 refers to an “early” phase φ₁ of the two-phase clock. The small spike seen in FIG. 8 is due to a mis-match of the transistors M1–M4, and disappears entirely if the transistors are made bigger. In the event there is no spike (i.e., the transistors M1–M4 are perfectly matched), an approximately 50% improvement in speed is expected.

Note further that in the event of using a plurality of cascaded amplifier stages for a pipeline architecture (designated A, B, C, D), if the A and B stage switches are driven by the phase φ₁, and the C and D stages are driven by φ₂, the transfer matrix 510 is only needed for the A stage and the C stage. On the other hand, if the switches of the stages A, B, C and D are driven by alternating clock phases (i.e., φ₁, φ₂, φ₁, φ₂), each stage will need its own transfer matrix 510.

Although the above discussion is primarily in terms of analog to digital converters, and the application of the circuit 510 shown in FIG. 6 is directed to analog to digital converters, it will also be appreciated that the circuit 510 can be used as a multiplexer, in applications other than analog to digital converters. This is because the signal feedthrough in a circuit such as 510 is substantially less than in conventional multiplexers.

FIG. 11 illustrates how the circuit discussed previously, here labeled 510A and 510B, can be used as a differential multiplexer. As shown in FIG. 11, the differential multiplexer includes two circuits 510A, 510B, each of which is identical and includes four transistors, M1A, M2A, M3A, and M4A in circuit 510A, and corresponding transistors in the circuit 510B. The circuit 510A inputs the positive differential signal V1 (V_1,POS, V_1,NEG) and the circuit 510B inputs the differential signal V2 (V_2,POS, V_2,NEG). The output of the two circuits 510A, 510B is thus the differential output (V_O,POS, V_O,NEG). Thus, for each circuit 510, for each group of four transistors, the middle two transistors (e.g., M2A, M3A) are always off; their purpose is that feedthrough from V2pos now goes to both V_O,POS and V_O,NEG. In other words, the middle switches, M2A, M3A, M2B, M3B are used for isolation. The purpose of the middle switches, M2A, M3A, M2B, M3B, is to provide effectively a “mirror parasitic capacitance” for the outer switches M1A, M4A, M1B, M4B, etc.

Therefore, the feedthrough, or leakage is common mode, and does not appear in the differential output voltage (V_O,POS, V_O,NEG). Thus, for the circuit 510A, both the common mode and the differential mode transfer function are one. For the circuit 510B, both transfers functions are zero.

This circuit has the advantage that there is no need to have switches in series, therefore the on-resistance Ron is not higher than in the conventional circuit shown in FIG. 9. Also, if the switches are clocked, the gate capacitance is lower than in the conventional circuit of FIG. 10.

FIG. 12 illustrates how the multiplexer circuit 510 described above can be adapted to not just a 2:1 multiplexer, but to, for example, a 4:1 multiplexer. This figure illustrates how the multiplexing concept shown in FIG. 11 can be generalized to any N:1 multiplexer. Four multiplexing circuits 510A–510D are arranged as shown, with the differential inputs V1–V4 fed into the four circuits 510A–510D. A single differential output (V_O,POS, V_O,NEG) is generated, with minimal feedthrough from any of the non-selected inputs to the output.

CONCLUSION

It will be understood by those skilled in the art that various changes in form and details may be made therein without departing from the spirit and scope of the invention as defined in the appended claims. Thus, the breadth and scope of the present invention should not be limited by any of the above-described exemplary embodiments, but should be defined only in accordance with the following claims and their equivalents.

Claims

1. A differential multiplexer comprising: a plurality of multiplexing circuits, each multiplexing circuit inputting a corresponding differential input signal including a positive input signal and a negative input signal, and outputting positive and negative output signals, and each multiplexing circuit comprising: first, second, third and fourth transistors,wherein the first and second transistors input the positive input signal,wherein the third and fourth transistors input the negative input signal,wherein outputs of the first and third transistors are connected to the positive output signal,wherein outputs of the second and fourth transistors are connected to the negative output signal; andwherein the positive output signals of the multiplexing circuits are coupled together at a first output node and the negative output signals are coupled together at a second output node.

2. The differential multiplexer of claim 1, wherein the positive and negative output signals are controlled using gate voltages on the first and fourth transistors.

3. The differential multiplexer of claim 1, wherein the second and third transistors are turned off when the differential multiplexer is in use.

4. A differential multiplexer comprising: a plurality of multiplexing circuits;each multiplexing circuit inputting a corresponding differential input signal including a positive input signal and a negative input signal, and outputting positive and negative output signals;each multiplexing circuit comprising: a plurality of transistors cross-coupled to make leakage between the positive and negative input signals common mode in the positive and negative output signals; andwherein the positive output signals of the multiplexing circuits are coupled together at a first output node and the negative output signals are coupled together at a second output node.

5. The differential multiplexer of claim 4, wherein each multiplexing circuit comprises: first, second, third and fourth transistors,wherein the first and second transistors input the positive input signal,wherein the third and fourth transistors input the negative input signal,wherein outputs of the first and third transistors are connected to the positive output signal,wherein outputs of the second and fourth transistors are connected to the negative output signal.

6. The differential multiplexer of claim 5, wherein the positive and negative output signals are controlled using gate voltages on the first and fourth transistors.

7. The differential multiplexer of claim 5, wherein the second and third transistors are turned off when the differential multiplexer is in use.