The present embodiments relate generally to communications systems, and specifically to preamble signal design for physical-layer frames in an orthogonal frequency-division multiplexing (OFDM) communication system.
In an orthogonal frequency-division multiplexing (OFDM) communication system, a transmitter encodes digital information and modulates it onto an analog carrier signal. Subsequently, a receiver demodulates and decodes the information. In such a system, the receiver should be well synchronized to the transmitter to minimize any performance degradation due to synchronization errors (e.g., time, frequency, and/or phase errors). This sensitivity to synchronization accuracy is especially pronounced in a high signal-to-noise ratio (SNR) environment including, for example, a wired communication system (e.g., a coaxial (“coax”) cable system).
Transceiver synchronization is sensitive to various signal impairments that affect the quality of the transmitted and received signals. Signal impairments may result from non-idealities in the front-ends of the transceivers or in the processing circuits therein. For example, mismatched active and passive elements (e.g., quadrature mixers, filters, digital-to-analog converters, and/or analog-to-digital converters) in the I and Q (in-phase and quadrature) signal paths introduce I/O mismatch impairments in the transmitted and received signals. I/O mismatch, which also may be referred to as I/O offset, is present in both the transmitter and receiver. In another example, carrier frequency offset (CFO) in the receiver, resulting from the difference in carrier frequency at the transmitter and the receiver (e.g., a difference in frequency of local oscillators that provide the carrier frequency in the transmitter and receiver), may impair the received signals. Channel effects (e.g., signal convolution with the channel) may also impair signals.
As services to be delivered over the communication system become more complex and multimedia rich, more data is sent. The complexity, speed, and sensitivity of the communication system are constantly pushed to the limit. Accordingly, there is a need for improved techniques to achieve transceiver timing synchronization and to estimate and compensate for signal impairments such as I/O offsets, CFO, channel effects, and/or other impairments to communication signals.
The present embodiments are illustrated by way of example and are not intended to be limited by the figures of the accompanying drawings. Like numbers reference like elements throughout the drawings and specification.
Techniques are disclosed for synchronizing a receiver with a transmitter, and more specifically, for compensating in the receiver for signal impairments introduced in the transmission between the transmitter and the receiver. In some embodiments, the techniques include detecting a preamble signal in a physical layer frame, estimating the signal impairments based on the preamble signal, and compensating for the estimated impairments. The embodiments provided herein enable accurate and low complexity acquisition and synchronization.
In some embodiments, a method of signal generation includes selecting a subset of contiguous OFDM symbols from a set of contiguous OFDM symbols, selecting a subset of contiguous subcarriers from a set of subcarriers, and generating a preamble that occupies the subset of contiguous subcarriers in the subset of contiguous OFDM symbols. The preamble includes portions in respective OFDM symbols of the subset of contiguous OFDM symbols. In the time domain each portion of the preamble corresponds to a repeating sequence of samples when subcarriers outside of the subset of contiguous subcarriers are filtered out. Generating the preamble includes flipping the sign of one or more occurrences of the repeating sequence of samples for a final portion of the preamble in one or more final OFDM symbols of the subset of contiguous OFDM symbols.
In some embodiments, a method of signal generation includes selecting a subset of contiguous OFDM symbols from a set of contiguous OFDM symbols, selecting a subset of contiguous subcarriers from a set of subcarriers, and generating a preamble that occupies the subset of contiguous subcarriers in the subset of contiguous OFDM symbols. The preamble includes portions in respective OFDM symbols of the subset of contiguous OFDM symbols. To generate the preamble, modulation symbols are placed on regularly spaced subcarriers in the subset of contiguous subcarriers within each portion of the preamble, in the frequency domain. The modulation symbols on the regularly spaced subcarriers for a respective portion of the preamble are phase-shifted with respect to the modulation symbols on the regularly spaced subcarriers for a previous portion of the preamble.
In some embodiments, a communications device includes a transmitter to transmit frames on multiple subcarriers. The frames each include multiple contiguous OFDM symbols. A respective frame includes a preamble that occupies a contiguous subset of the multiple subcarriers and includes portions in respective OFDM symbols of a contiguous subset of the multiple contiguous OFDM symbols. In the time domain each portion of the preamble corresponds to a repeating sequence of samples when subcarriers outside of the subset of contiguous subcarriers are filtered out. The sign of one or more occurrences of the repeating sequence of samples is flipped for a final portion of the preamble in one or more final OFDM symbols of the subset of contiguous OFDM symbols.
In some embodiments, a receiver includes a filter to extract samples corresponding to a signal carried on a contiguous group of subcarriers that form a subset of a set of available subcarriers. The receiver also includes a preamble detector to detect a preamble in the extracted samples. The preamble includes a repeating sequence of samples. The receiver further includes a preamble boundary searcher to identify an end of the preamble as indicated by one or more occurrences of the repeating sequence of samples having flipped signs with respect to previous occurrences of the repeating sequence of samples.
In the following description, numerous specific details are set forth such as examples of specific components, circuits, and processes to provide a thorough understanding of the present disclosure. Also, in the following description and for purposes of explanation, specific nomenclature is set forth to provide a thorough understanding of the present embodiments. However, it will be apparent to one skilled in the art that these specific details may not be required to practice the present embodiments. In other instances, well-known circuits and devices are shown in block diagram form to avoid obscuring the present disclosure. The term “coupled” as used herein means connected directly to or connected through one or more intervening components or circuits. Any of the signals provided over various buses described herein may be time-multiplexed with other signals and provided over one or more common buses. Additionally, the interconnection between circuit elements or software blocks may be shown as buses or as single signal lines. Each of the buses may alternatively be a single signal line, and each of the single signal lines may alternatively be buses, and a single line or bus might represent any one or more of a myriad of physical or logical mechanisms for communication between components. The present embodiments are not to be construed as limited to specific examples described herein but rather to include within their scopes all embodiments defined by the appended claims.
The synchronization can be generally categorized into time synchronization and frequency synchronization. For purposes of discussion herein, time synchronization may include PHY frame synchronization and OFDM symbol synchronization. Frequency synchronization may include carrier frequency synchronization and sampling frequency synchronization. Synchronization accuracy is of particular importance when the transceivers operate in a high signal-to-noise ratio (SNR) environment including, for example, a wired (e.g., coax cable) communication system.
As is further discussed in detail in relation to
In accordance with one or more embodiments, each of the PHY frames 210 (e.g., frames 210(1), 210(2), and 210(3)) includes a plurality of OFDM symbols 220. The preamble signal 230 in each of the PHY frames 210 is a narrowband signal that functions as one of the available reference signals. As shown in
Further, the preamble signal 230 is depicted in
According to some embodiments, the preamble signal portion 232 is configured and constructed in a way that, after band-pass or low-pass filtering (more details of which are described regarding
In one or more embodiments, the last NCP samples of the last sequence 250(M) may be copied and appended to the front of the first sequence 250(1) to complete the OFDM symbol. The copied portion is referred to as cyclic prefix (CP) 252, and NCP is the CP length. The CP 252 may provide support to mitigate inter-symbol interference (ISI) caused by frequency-selective fading, or to perform symbol synchronization and some limited CFO estimation. It is noted that, in the embodiment of
As shown in
The first time-domain preamble portion 332 results from transformation of the first frequency-domain preamble portion 330 from the frequency domain to the time domain and CP insertion. The first time-domain preamble portion 332 is an example of the preamble signal portion 232,
In some embodiments, the selected subcarriers are symmetrical about a direct current (DC) subcarrier 340. It is noted that the selected subcarriers for the preamble signal need not be symmetrical about the DC subcarrier 340. For embodiments in which the selected subcarriers are not symmetrical about the DC subcarrier 340, a band-pass filter and a down-conversion circuit may be used instead of a low-pass filter in the preamble searcher circuit 700 (
Because of filter imperfection (e.g., in the search path filter 720,
In one or more embodiments, because the (M−1) number of subcarriers out of every M subcarriers are nullified, the transmitter 102 may boost the power of modulation symbols (e.g., QAM symbols) placed on the selected subcarriers in the subset 336, to compensate for the energy loss caused by nullification. For example, if M=2 represents a 3 dB energy loss in the transmission of the preamble signal, the transmitter may compensate for this loss by increasing the power on the selected subcarriers by 3 dB.
According to some embodiments, a cyclic time-shift is applied to maintain the periodic structure of the preamble signal across multiple OFDM symbols. The cyclic time-shift may be applied to the second frequency-domain preamble portion 430 by multiplying the modulation (e.g., QAM) symbols contained within the first frequency-domain preamble portion 330 with a phase ramp and using the resulting modulation symbols in the second frequency-domain preamble portion 430. The modulation symbols on the selected subcarriers in the second frequency-domain preamble portion 430 thus are phase-shifted with respect to the modulation symbols on the selected subcarriers in the first frequency-domain preamble portion 330. This multiplication may be represented in the following equation:
p
n+1(f)=pn(f)×exp(j2πfNCP/NSC), (Eq. 1)
where p stands for a modulation symbol (e.g., QAM symbol) on a selected subcarrier in the subset 336, n represents the index of the OFDM symbol 220, f is the subcarrier index, and the item (j2πfNCP/NSC) represents the phase shift.
As illustrated in
The last frequency-domain preamble portion 530 may be transformed in a similar manner as the second frequency-domain preamble portion 430 (
However, in some embodiments, a 180-degree phase rotation (which results in a sign flip in the time domain) may be performed on one or more sequences (e.g., one or more of the sequences 250(1) through 250(M),
Transmissions in communication systems such as the system 100 (
As previously described, the preamble signals of one or more embodiments select one out of every M subcarriers for preamble signal transmission, and nullify the (M−1) subcarriers that are not selected. However, for those embodiments that employ both pilot symbols 640 and the preamble signal 630, depending on the situation, it may not be necessary to nullify those pilot symbols 640 on subcarriers within the preamble signal 630 (i.e., those pilot symbols 640 that fall within the preamble signal 630). For example, if a respective pilot symbol 640 is present in the preamble signal 630, but the respective pilot symbol is not on the one or more subcarriers that are selected for preamble signal transmission, then the transmitter leaves the respective pilot symbol 640 in place. Some preamble signal quality degradation may be observed because of the pilot symbols 640 within the preamble signal 630. The receiver 106 is configured to tolerate the degradation of preamble signal quality that results from the overlapping of pilot symbols 640 with the preamble signal 630.
If a respective pilot symbol 640 is present in the preamble signal 630, and the respective pilot symbol 640 overlaps with the one or more subcarriers that are selected for preamble signal transmission (i.e., is on one of the selected subcarriers), then the transmitter 102 may be configured to either (1) overwrite the preamble signal 630 with the pilot symbol 640, or (2) overwrite the pilot symbol 640 with the preamble signal 630. In the former case, the modulation symbol in the preamble signal 630 that overlaps with the pilot symbol 640 is replaced with the pilot symbol 640. In the latter case, the pilot symbol 640 is replaced with the modulation symbol with which it overlaps in the preamble signal 630. Each approach may insert signal distortion that affects the overall functionality of the overwritten signals, and different approaches may be selected in different embodiments.
L
1
[k]=|Φ
rr
[k]|−(Prr[k])/2, (Eq. 2)
where Φrr[k] is the correlation value, and Prr[k] represents the energy detected.
The correlation function Φrr[k] may be expressed as:
Φrr[k]=Σi=0L-1r*[k+i]·r[L+k+i], (Eq. 3)
where k represents the sample index, r stands for received samples, and L is the duration of a single period (e.g., the period of sequences 250(1) through 250(M),
And, the energy function Prr[k] may be expressed as:
P
rr
[k]=Σ
i=0
L-1
|r[i]|
2
+|r[L+i]|
2. (Eq. 4)
According to some embodiments, the preamble signals may be detected when the likelihood metric as provided in Eq. 2 shows a transition from a relatively low value to a relatively high value. In some alternative embodiments, the likelihood metric may be inversed, and the preamble signals may be detected when the likelihood metric shows a transition from a relatively high value to a relatively low value.
After the preamble signals are detected, carrier frequency offset (CFO) estimation and frame boundary detection may be performed based on the preamble signals. In one or more embodiments, the CFO estimation and boundary detection may be based on observing the phase of the correlation function Φrr[k] (Eq. 3). For purposes of CFO estimation and frame boundary detection, L is preferably selected to be the duration of a single period. Nonetheless, depending on the embodiments, correlations over multiple periods may be accumulated to increase CFO estimation accuracy. The CFO estimation may be based on the actual correlation phase. The amount of the CFO detected is directly proportional to the phase deviation between two sequential preamble signals. As previously mentioned, frame boundary detection may be achieved by detecting a 180-degree phase rotation (i.e., a sign flip). In one or more embodiments, the CFO estimation is performed before the frame boundary detection, and therefore no sign-flip is assumed during the CFO estimation. Likewise, in some embodiments, the frame boundary detection is performed after the CFO estimation, and therefore no CFO is assumed during the boundary detection.
In some embodiments, other types of signal distortion may be estimated and compensated based on the preamble signals in addition to the above-mentioned CFO. These other types of signal distortion may include, for example, multi-path effects, I/O mismatches, and so forth.
With simultaneous reference to
The preamble searcher circuit 700 includes a search path filter 720, a sliding correlator 730, a preamble detector 740, and a preamble boundary searcher 750. The receiver 106 (
At the beginning of the acquisition phase (e.g., when the system is first turned on), the preamble searcher circuit 700 searches for the preamble signal among the samples received in the sample buffer 710 (802). To perform the search, the samples are first filtered by the search path filter 720, which at this time is coupled to the sample buffer by a switch 712. The search path filter 720 may be a low-pass filter, a band-pass filter, or any suitable type of filter that is configured to extract the preamble signals (e.g., to extract the K subcarriers in the subset 336 from the NSC subcarriers 338,
Because the preamble has not yet been found (804—No) at this point in the acquisition phase, there is not yet any CFO estimation or compensation. The samples extracted by the search path filter 720 are therefore sent directly to the sliding correlator 730, without being adjusted by a CFO compensation module 764 coupled between the search path filter 720 and the sliding correlator 730. The sliding correlator 730 calculates the correlation function Φrr[k] of Eq. 3. Two samples are provided to the sliding correlator 730 in a given cycle: a sample r[k] and a sample r[k−L] that has been delayed by L cycles by a delay stage 725. (While
The samples r[k] and r[k−L] as well as the output Φrr[k] of the sliding correlator 730 are delivered to the preamble detector 740 through a switch 738, which selectively couples the sliding correlator 730 to the preamble detector 740. The preamble detector 740 includes circuitry to perform energy calculations on the received samples by calculating the energy function Prr[k] (Eq. 4). To calculate Prr[k], the preamble detector determines the squared magnitude (“Abs( )2”) of the samples r[k] and r[k−L], adds the squared magnitudes of these two samples using a combiner, and generates a moving sum using an integrator. The moving sum output by the integrator is Prr[k], in accordance with Eq. 4.
The preamble detector 740 determines a likelihood metric based on the energy function Prr[k] and the correlation function Φrr. [k], and determines if any preamble signal is received based on the likelihood metric. In the example of
L
2
[k]=(Prr[k])−2|Φrr[k]| (Eq. 5).
To determine the likelihood metric L2[k], the preamble detector 740 determines a value equal to twice the magnitude (2*Abs( )) of the correlation function Φrr[k], and uses a combiner to subtract this value from the energy function Prr[k]. The output of the combiner is the likelihood metric L2[k].
In the example depicted in
After the preamble is found (804—Yes), the preamble searcher circuit 700 directs the samples (which are samples of the preamble signal) from the search path filter 720 to CFO estimation and compensation circuitry 760 in the receiver 106. The CFO estimation and compensation circuitry 760 may estimate (806) and compensate for CFO based on the preamble signals in the manner described above. A switch 768 closes in response to assertion of the “Enable CFO Estimation” signal, thereby coupling the output of the search path filter 720 to the input of a CFO estimation module 762, which estimates (806) the CFO. In addition to the CFO estimation module 762, the CFO estimation and compensation circuitry 760 includes a CFO compensation module 764 coupled between the search path filter 720 and the sliding correlator 730, and a CFO compensation module 766 that is selectively coupled to the sample buffer 710 by the switch 712. The CFO compensation modules 764 and 766 compensate for the estimated CFO (ωTs and ωTs,ds) as determined by the CFO estimation module 762. In many embodiments, CFO estimation and compensation circuitry 760 may include other types of modules to compensate other types of signal distortions including, for example, multi-path effects, I/O mismatches, and so forth.
After the preamble is found, the preamble searcher circuit 700 may also start to search for the frame boundary by searching for the sign-flip (808). At this stage, the correlation results (e.g., the values of the correlation function Φrr[k], Eq. 3) from the correlator 730 are redirected to the preamble boundary searcher 750, which looks for the 180-degree phase change (i.e., the sign flip) among the samples it receives. For example, the CFO estimation module 762 asserts an “Enable Sign Flip Search” signal upon generating a CFO estimate. In response to assertion of the “Enable Sign Flip Search” signal, the switch 738 couples the output of the sliding correlator 730 to the input of the preamble boundary searcher 750. The preamble boundary searcher 750 determines the real portion (“Re( )”) of this input and then searches (808) for the sign flip. (In the embodiment illustrated in
If the preamble boundary searcher 750 finds the sign-flip (810—Yes), it determines that the boundary of the preamble signal (or an OFDM symbol at or near the end of the preamble signal) is found, and that the acquisition phase is to be transferred into the tracking phase. (If the preamble boundary searcher 750 does not find the sign-flip (810—No), it continues to search (808) for the sign flip.) The CFO estimate as determined during the acquisition phase is applied to compensate signal distortions during the tracking phase. For example, the preamble boundary searcher 750 asserts an “Enable CP-based Search” signal in response to finding the sign-flip. In response, the switch 712 couples the sample buffer 710 to the CFO compensation module 766. The CFO compensation module 766 performs CFO compensation for the samples from the sample buffer 710, based on the estimated CFO (ωTs) provided by the CFO estimation module 762, and provides the compensated samples to a CP-based searcher 770. Also, the preamble boundary searcher 750 provides to the CP-based searcher 770 a value τds indicating the boundary in the down-sampled domain of the last OFDM symbol carrying the preamble. The CP-based searcher 770 uses the value τds to perform a CP-based search for OFDM symbol boundaries at times τ. Thus, during the tracking phase, the preamble searcher circuit 700 continues to perform finer-level time synchronization based, for example, on CP processing (812).
Attention is now directed to methods of generating signals that include preambles.
In the method 900, a subset of contiguous OFDM symbols (e.g., the second, third, and fourth OFDM symbols 220 in the PHY frame 210(1), 210(2), or 210(3),
A preamble (e.g., a preamble signal 230,
In some embodiments, to generate the preamble, regularly spaced subcarriers (e.g., with the 1/M spacing shown in
The method 900 may further include placing one or more pilot symbols (e.g., pilot symbols 640,
In the method 950, a subset of contiguous OFDM symbols is selected (902) from a set of contiguous OFDM symbols and a subset of contiguous subcarriers is selected (904) from a set of subcarriers, as described for the method 900 (
A preamble (e.g., a preamble signal 230,
In some embodiments, to generate the preamble, modulation symbols (e.g., QAM symbols) are placed (954) on regularly spaced subcarriers (e.g., with the 1/M spacing shown in
The method 950 may further include placing one or more pilot symbols (e.g., pilot symbols 640,
While the methods 900 and 950 include a number of operations that appear to occur in a specific order, it should be apparent that the methods 900 and 950 can include more or fewer operations, which can be executed serially or in parallel. An order of two or more operations may be changed, performance of two or more operations may overlap, and two or more operations may be combined into a single operation. Furthermore, the methods 900 and 950 may be combined into a single method, such that the preamble is generated in a combination of the operation 906 (e.g., including the operations 908, 910, and 912) and the operation 952 (e.g., including the operations 954, 956, 958, and 960).
In the foregoing specification, the present embodiments have been described with reference to specific exemplary embodiments thereof. It will, however, be evident that various modifications and changes may be made thereto without departing from the broader scope of the disclosure. The specification and drawings are, accordingly, to be regarded in an illustrative sense rather than a restrictive sense.
This application claims priority to U.S. Provisional Patent Application No. 61/738,367, titled “Narrow Band Preamble for Orthogonal Frequency Division Multiplexing System,” filed Dec. 17, 2012, which is hereby incorporated by reference in its entirety.
Number | Date | Country | |
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61738367 | Dec 2012 | US |