The exemplary and non-limiting embodiments of this invention relate generally to wireless communication systems, methods, devices and computer programs and, more specifically, relate to processing a receive-diversity signal.
This section is intended to provide a background or context to the invention that is recited in the claims. The description herein may include concepts that could be pursued, but are not necessarily ones that have been previously conceived or pursued. Therefore, unless otherwise indicated herein, what is described in this section is not prior art to the description and claims in this application and is not admitted to be prior art by inclusion in this section.
In at least the high speed downlink packet access (HSDPA) system, the receiver which employs the equalizer and receiver diversity is termed a type 3 receiver according to the defining specifications (e.g., 3GPP TS25.101, “User equipment (UE) radio transmission and reception (FDD) (release 8)”, V8.2.0 2008-03).
Extensive studies have been done on the HSDPA equalizer technique with receiver diversity. Typically, the linear minimum mean squared error (LMMSE) based equalizer, which jointly utilizes the received signals from two receiver branches, is considered for such a type 3 receiver. The complexity of the MMSE equalizer may be high (mainly such high complexity could be due to the required matrix inversion). For example, a paper by Jianzhong Zhang, Tejas Bhatt and Giridhar Mandyam, entitled “Efficient linear equalization for high data rate downlink CDMA signalling” (proceeding of IEEE 37th Asilomar conference 2003, vol. 1, pp 141-145) describes a fast Fourier transform (FFT) based MMSE approximation equalization method where the matrix inversion is avoided. The FFT based MMSE equalizer can be straightforwardly extended to the receive diversity case given the assumption that the noise variances at two receive branches are equal. However, due to the fact that two receive antennas located in the UE are independently separated, in addition to different propagation channels, the geometry values, or signal to noise ratios, associated with two receive branches can also be very different.
What is needed in the art is a receiver and method for receiving diversity signals that has reasonable computational complexity and that exploits the difference in noise variation at the different receive diversity branches. For example, it would be advantageous to have a receiver that does not require correlation matrix inversion. As another example, it would be advantageous to have a receiver that does not assume, when processing a received signal on two or more receive branches, that noise variation across the diversity receive branches is always the same regardless of actual noise conditions.
In accordance with a first exemplary aspect of the invention there is a method comprising: determining a first noise variance for a signal received on a first diversity branch; determining by the apparatus a second noise variance for a signal received on a second diversity branch; scaling the signal received on the second diversity branch as a function of a ratio of the first noise variance and the second noise variance; and estimating a received signal by combining the signal on the first diversity branch with the scaled signal on the second diversity branch.
In accordance with a second exemplary aspect of the invention there is a memory storing a program of computer readable instructions that when executed by at least one processor result in actions. In this second aspect the actions comprise: determining a first noise variance for a signal received on a first diversity branch of a receiver; determining a second noise variance for a signal received on a second diversity branch of the receiver; scaling the signal received on the second diversity branch as a function of a ratio of the first noise variance and the second noise variance; and estimating a received signal by combining the signal on the first diversity branch with the scaled signal on the second diversity branch.
In accordance with a third exemplary aspect of the invention there is an apparatus comprising a memory storing a program of computer readable instructions; and at least one processor. The at least one processor is configured, with the memory, to: determine a first noise variance for a signal received on a first diversity branch of the apparatus; determine a second noise variance for a signal received on a second diversity branch of the apparatus; scale the signal received on the second diversity branch as a function of a ratio of the first noise variance and the second noise variance; and estimate a received signal by combining the signal on the first diversity branch with the scaled signal on the second diversity branch.
In accordance with a fourth exemplary aspect of the invention there is an apparatus comprising first determining means; second determining means; scaling means; and estimating means. The first determining means is for determining noise variance for a signal on a first diversity branch of a receiver. The second determining means is for determining noise variance for a signal on a second diversity branch of the receiver. The scaling means is for scaling the signal on the second diversity branch as a function of a ratio of the first noise variance and the second noise variance. And the estimating means is for estimating a received signal by combining the signal on the first diversity branch with the scaled signal on the second diversity branch.
Following is described a process (e.g., method, algorithm) for noise variance estimation, which are the necessary information for the design of the LMMSE equalizer. Such an equalizer may be employed in a particular embodiment within a HSDPA receiver with two receive branches. While such a use is the context of the description below, the HSDPA receiver is not a limitation to these teachings; they may be employed in any diversity receiver such as may be operating in a UTRAN (universal mobile telecommunications terrestrial radio access network) system, or an E-UTRAN (evolved UTRAN) system, or a WLAN (wireless local area network) system, or a WCDMA (wireless code division multiple access) system, a cognitive radio system, and the like. Additionally, it will be apparent these teaching may be employed in any receiver operating in a multiple-input and multiple-output system.
Notations used below are conventional, as follows. Upper- and lowercase boldface letters denote matrices and vectors, respectively. (.)T denotes the transpose; (.)H denotes the Hermitian transpose; and (.)* defines the complex conjugate operation. Matrix In stands for the identity matrix of order n. E(a·b*) denotes the correlation coefficient between two random variables. A carat over a matrix or vector) ({circumflex over (·)}) indicates it is an estimate.
As noted above, the type 3 HSDPA receiver typically applies the LMMSE method for the equalizer. Let F and L define the equalizer filter length and the channel impulse response length in samples, respectively. The received signal model sampled at two times the chip rate from two receive antennas can be described as
Terms from equation [1] are defined as follows.
denotes the chip vector consisting of the n th and
onwards transmitted chips.
As shown at equation [3], the LMMSE estimates needs the information about the channel impulse responses (CIRs) and noise variances with respect to two receive antennas. Conventional approaches to processing diversity signals on different branches assumes that the noise variance on both branches is equal (in a HSDPA type 3 receiver, this is reflected as assuming the common pilot channel (CPICH) power is one).
As a result, the estimated CIR is a scaled version of the true CIR, i.e., ĥi=√{square root over (γP)}hi+nCIR, therefore the LMMSE solution shown at equation [3] using the CIR estimates becomes
As shown at equation [4], in addition to the CIR estimate ĥ, the LMMSE solution for the type 3 receiver also needs the noise variance estimates and pilot power to transmitted signal power ratio Ec/Ior (γP) estimate.
Most of the current type 3 receiver algorithms typically assume that both receivers have the same noise variance, for example, σn
As shown at equation [5], it is not necessary to explicitly estimate the pilot Ec/Ior and noise variance for the LMMSE solution. Instead, a single diagonal loading factor κ needs to be determined. Ideally, κ can be obtained by calculating the product of γP and ση2. In practice, some suboptimal solutions are usually utilized to provide κ. For instance, the signal to noise ratio (SNR) for the CPICH is measured at the output of the equalizer or the de-spreader, which is then used to adjust the diagonal loading factor to maximize the SNR of the signal being measured.
To further reduce the complexity of matrix inversion (2F×2F) required by equation [5], by virtue of the matrix-inversion lemma (see for example G. H. Golub and C. V. Loan, “Matrix Computation”, 3rd ed. Johns Hopkins University Press, 1996, page 50), equation [5] can be formulated as:
The equalizer taps for two receive branches can be further obtained from equation [6] as
Then the estimate of the composite chip s[n−D] in equation [3] above can be expressed as follows:
ŝ[n−D]=w
1,LMMSE
H
·r
1
[n]+w
2,LMMSE
H
·r
2
[n] [8]
In addition to the dimension reduction of the matrix inversion in equation [7], for example, from a size (2F×2F) matrix to a size
the matrix,
demonstrates a Toeplitz structure, so that the FFT based algorithm in the paper cited in background above by Jianzhong Zhang, Tejas Bhatt and Giridhar Mandyam can be directly applied in the receiver diversity case as well.
The preceding development of the solution shown in equations [5] through [8] is based on the assumption that two receive branches have the same noise variance. However, due to a slow moving channel and independent radio frequency (RF) receiver branches, it has been observed in practical wireless systems that the noise variances at two baseband receivers may not be equal in general, and sometimes they can have a few dB differences so that the solution shown in equation [7] may not exhibit a good performance.
Next, exemplary embodiments of an LMMSE solution which accommodates for different noise variances are described. The exemplary embodiments presented herein estimate the noise variances for two receive branches. As will be shown, the LMMSE solution of those exemplary and non-limiting embodiments have similar complexity as equation [7] by virtue of the estimated noise variances. More specifically, exemplary embodiments presented herein estimate the noise variance ση
Exemplary embodiments of this invention utilize the autocorrelation and cross-correlation coefficients of the received signals from two receive branches, and the channel impulse response (CIR) estimates ĥ from the CPICH channel to estimate the noise variances ση
While the examples detailed below are in the context of two receive diversity branches (e.g., two receive antennas), this is not a limit to these teachings as noted above. A receiver according to these teachings may have two or three or more diversity branches, and the pair-wise diversity processing may be done on any pair or on multiple pairs of those receive diversity branches. For example, a receiver with three diversity branches may process according to the below examples on branch 1 and 2, and also on branch 2 and 3, and then process the two results as another pair of diversity branches for combining and eventual output as the estimate of the received signal.
The three correlation coefficients for the two branches are calculated as follows
Re-arranging terms of equation [9] shows that the pilot power to the total transmitted signal power ratio (Ec/Ior or γP) and noise variances {circumflex over (σ)}η2 can be estimated as:
However, in practice the receiver has only the estimates of Ci,j,i, j=1,2, for example, Ĉi,j. Further, due to the constraints of the pilot power to the total transmitted signal power ratio Ec/Ior and noise variances, the pilot power ratio Ec/Ior and noise variances can be estimated as follows:
where k+ stands for the minimum of the noise variance in the system. This minimum may be set as a design factor in a practical receiver, and in the simulations detailed below it is set to 0.1.
With the knowledge of the pilot power ratio Ec/Ior and noise variance estimates, the LMMSE solution is derived similar to equation [7]. Without loss of generality, we proceed further by assuming {circumflex over (σ)}η
The computations then are:
It is clear that
The LMMSE estimate of s[n−D] from the observation vector {tilde over (r)}[n] then becomes:
Using the terms of equation [12], equation [7] is then changed to
where κ={circumflex over (γ)}P{circumflex over (σ)}η
Then ŝ[n−D] in eq. [8] becomes:
ŝ[n−D]={tilde over (w)}
1,LMMSE
H
·r
1
[n]+
2,LMMSE
H
·{tilde over (r)}
2
[n]. [14]
It should be pointed out that the example detailed above can be extended to the case where more than 2 receive antennas are used in the user equipment (UE). If a higher computation complexity is allowed in the UE, for instance, the UE is able to estimate the covariance matrix of the receive signals from two receiver antennas, that is, the autocorrelation and cross-correlation functions of the received signals are available in the UE. The pilot Ec/Ior and noise variances can be estimated according to known methods or methods yet to be developed.
Along the first diversity branch 110 an estimate of the channel impulse response (CIR) ĥ1 of the channel over which a first pilot signal was received (which is the same channel over which the baseband first diversity signal r1(n) was received) is estimated at a first channel estimation block 116. This first-branch CIR estimate ĥ1 is then output to both a noise variance and pilot power ratio block 130, and to a tap solver block 140.
In one implementation specific to the HSDPA system, the pilot signals are received on the CPICH which is transmitted from the transmitter's antenna. In current specifications for HSDPA, the CPICH is transmitted at the same time as other data channels that are processed on the diversity branches, but the CPICH uses a different spreading code than the data channels. The CPICH is therefore considered a superimposed training sequence, and so is considered herein as the ‘same’ channel as the data. At the receiver, both branches use the CPICH to correlate the respective received signal to obtain the CIR estimate. It is noted that this correlation based CIR estimation method is also typical in downlink for the WCDMA system where the superimposed training signal is used. Other systems which use time multiplexed training signal, for example GSM (global system for mobile communications), CDMA2000 (code division multiple access 2000) and TD-SCDMA (time division-synchronous code division multiple access), would typically employ a more complex method such as for example a least squares algorithm to find the CIR estimate.
Along the second diversity branch 120 an estimate of the channel impulse response (CIR) ĥ2 of the channel over which a second pilot signal was received (which is the same channel over which the baseband second diversity signal r2(n) was received) is estimated at a second channel estimation block 126. The second CIR ĥ2 is output to both the noise variance and pilot power ratio block 130 and towards the tap solver block 140. In an exemplary embodiment there is only one channel estimation block doing both estimations.
Also, from the baseband first diversity signal r1(n) and second diversity signal r2(n) there is computed at a correlation estimator 150 correlation coefficients Ci,j, which is an estimate from the baseband signals on the ith (first) and jth (second) diversity branches. Equation [9] above is one way in which the correlation estimator may compute the correlation coefficients C12, C11 and C22. These are also input to the noise variance and pilot power ratio block 130.
The noise variance and pilot power ratio block then has three separate inputs: the three correlation coefficients from the correlation estimation block; the first channel impulse response ĥ1; and the second channel impulse response (CIR) ĥ2. From these are computed the three values shown at equation [10-a]: the estimated pilot channel power to total transmitted signal power ratio {circumflex over (γ)}P; the noise variance {circumflex over (σ)}η
From these three results the noise variance and pilot power ratio estimator block 130 outputs two values: the power adjusted noise (diagonal loading factor) κ={circumflex over (γ)}p{circumflex over (σ)}η
for the second diversity branch 120. The noise variance scaling factor
is used to scale the second channel impulse response ĥ2 at a first multiplier 129a; and to scale the baseband second diversity signal r2(n) at a second multiplier 129b.
Then the outputs of the tap solver block 140, which may be computed according to equation [13] above, are used to set the filter coefficients of a first finite impulse response (FIR) filter 118 along the first diversity branch 110, and to set the filter coefficients of a second finite impulse response (FIR) filter 128 along the first diversity branch 110. The filtered baseband signals diversity signals r1(n) and r2(n) are finally combined at a combiner 160 so as to finally output the estimate for the D th delayed transmitted composite chip ŝ[n−D] according to equation [14] above.
The noise variance and pilot power ratio estimation block 130 provides the proper scaling factor
for the CIR estimate and for the received signal corresponding to the second receive branch, as well as the diagonal loading factor κ={circumflex over (γ)}p{circumflex over (σ)}η
The technique outlined with respect to
Now is detailed an exemplary environment for the exemplary embodiment of
The UE 10 includes a controller, such as a computer or a data processor (DP) 10A, a computer-readable memory medium embodied as a memory (MEM) 10B that stores a program of computer instructions (PROG) 10C, and a suitable radio frequency (RF) transceiver 10D for bidirectional wireless communications with the access node 12 via one or more antennas (112, 122 in
Similarly, the higher node 14 includes a controller, such as a computer or a data processor (DP) 14A, a computer-readable memory medium embodied as a memory (MEM) 14B that stores a program of computer instructions (PROG) 14C, and a modem (not shown) for communication with the access node 12 over the data/control path 13. The higher node 14 may also interface the access node to other networks such as for example a publicly switched telephone network or the Internet.
While the exemplary embodiment of
According to an exemplary embodiment of the invention the UE 10 may be assumed to also include a noise variance and pilot SNR estimation block 10E, and the access node 12 may include a noise variance and pilot SNR estimation block 12E. Each of these is functionally similar to the similar block 130 detailed with respect to
In general, the various embodiments of the UE 10 can include, but are not limited to, cellular telephones, personal digital assistants (PDAs) having wireless communication capabilities, portable computers having wireless communication capabilities, image capture devices such as digital cameras having wireless communication capabilities, gaming devices having wireless communication capabilities, music storage and playback appliances having wireless communication capabilities, Internet appliances permitting wireless Internet access and browsing, as well as portable units or terminals that incorporate combinations of such functions.
The computer readable MEMs 10B and 12B may be of any type suitable to the local technical environment and may be implemented using any suitable data storage technology, such as semiconductor based memory devices, flash memory, magnetic memory devices and systems, optical memory devices and systems, fixed memory and removable memory. The DPs 10A and 12A may be of any type suitable to the local technical environment, and may include one or more of general purpose computers, special purpose computers, microprocessors, digital signal processors (DSPs) and processors based on a multicore processor architecture, as non-limiting examples.
Within the sectional view of
Signals to and from the camera 28 pass through an image/video processor 44 which encodes and decodes the various image frames. A separate audio processor 46 may also be present controlling signals to and from the speakers 34 and the microphone 24. The graphical display interface 20 is refreshed from a frame memory 48 as controlled by a user interface chip 50 which may process signals to and from the display interface 20 and/or additionally process user inputs from the keypad 22 and elsewhere.
Certain embodiments of the UE 10 may also include one or more secondary radios such as a wireless local area network radio WLAN 37 and a Bluetooth® radio (BT) 39, which may incorporate an antenna on-chip or be coupled to an off-chip antenna. Throughout the apparatus are various memories such as random access memory RAM 43, read only memory ROM 45, and in some embodiments removable memory such as the illustrated memory card 47 on which the various programs 10C are stored. All of these components within the UE 10 are normally powered by a portable power supply such as a battery 49.
The aforesaid processors 38, 40, 42, 44, 46, 50, if embodied as separate entities in a UE 10 or in access node 12, may operate in a slave relationship to the main processor 10A, 12A, which may then be in a master relationship to them. In an exemplary embodiment the noise variance and pilot power ratio block 130 (also shown as 10E for the UE and 12E for the access node) is embodied within the baseband chip 42, though it is noted that other embodiments need not be disposed there but may be disposed across various chips and memories as shown or disposed within another processor that combines some of the functions described above for
Note that the various chips (e.g., 38, 40, 42, etc.) that were described above may be combined into a fewer number than described and, in a most compact case, may all be embodied physically within a single chip.
Specific embodiments of the access node 12 may reflect in part the various chips and memories shown at
The specific exemplary embodiment of
Based on the foregoing it should be apparent that the exemplary embodiments of this invention provide a method, apparatus and a memory storing a computer program(s) that when executed by a processor result in actions which are detailed at
In accordance with more specific implementations, a first autocorrelation coefficient is computed from the signal on the first diversity branch and a second autocorrelation coefficient is computed from the signal on the second diversity branch and a cross correlation coefficient is computed from both of those signals (block 602). These three correlation coefficients are then used to determine the two noise variances of the above paragraph.
In accordance with another specific implementation which may further be combined with the correlation coefficient aspect, a first channel impulse response for the channel over which the first signal was received is estimated from a pilot signal, and a second channel impulse response for the channel over which the second signal was received is estimated from a pilot signal (block 604). The noise variances are determined using these channel impulse response estimates.
In accordance with another specific implementation which may further be combined with the channel impulse response aspect (and also with the correlation coefficient aspect), there is also determined a ratio of pilot channel power to total signal power (block 606), where the pilot channel power is for the channel over which the pilot signals were received and total signal power is for the signal being estimated. In an embodiment, that power ratio is used to set coefficients for a first filter for filtering the signal on the first diversity branch and to set coefficients for a second filter for filtering the scaled signal on the second diversity branch (block 608) before they are combined into the estimated signal.
In a specific exemplary embodiment of the invention as apparatus, such an apparatus includes first determining means for determining noise variance for a signal on a first diversity branch (e.g. the noise variance estimation function at block 130 of
The various blocks shown in
In general, the various exemplary embodiments may be implemented in hardware or special purpose circuits, software, logic or any combination thereof. For example, some aspects may be implemented in hardware, while other aspects may be implemented in firmware or software which may be executed by a controller, microprocessor or other computing device, although the invention is not limited thereto. While various aspects of the exemplary embodiments of this invention may be illustrated and described as block diagrams, flow charts, or using some other pictorial representation, it is well understood that these blocks, apparatus, systems, techniques or methods described herein may be implemented in, as nonlimiting examples, hardware, software, firmware, special purpose circuits or logic, general purpose hardware or controller or other computing devices, or some combination thereof.
It should thus be appreciated that at least some aspects of the exemplary embodiments of the inventions may be practiced in various components such as integrated circuit chips and modules, and that the exemplary embodiments of this invention may be realized in an apparatus that is embodied as an integrated circuit. The integrated circuit, or circuits, may comprise circuitry (as well as possibly firmware) for embodying at least one or more of a data processor or data processors, a digital signal processor or processors, baseband circuitry and radio frequency circuitry that are configurable so as to operate in accordance with the exemplary embodiments of this invention.
Various modifications and adaptations to the foregoing exemplary embodiments of this invention may become apparent to those skilled in the relevant arts in view of the foregoing description, when read in conjunction with the accompanying drawings. However, any and all modifications will still fall within the scope of the non-limiting and exemplary embodiments of this invention.
For example, while the exemplary embodiments have been described above in the context of the HSDPA system, it should be appreciated that the exemplary embodiments of this invention are not limited for use with only this one particular type of wireless communication system, and that they may be used to advantage in other wireless communication systems such as for example UTRAN (universal mobile telecommunications system terrestrial radio access network), E-UTRAN (evolved UTRAN or long term evolution LTE of UTRAN), WCDMA (wideband code division multiple access), WLAN (wireless local area network), GSM (global system for mobile communications), and others.
It should be noted that the terms “connected,” “coupled,” or any variant thereof, mean any connection or coupling, either direct or indirect, between two or more elements, and may encompass the presence of one or more intermediate elements between two elements that are “connected” or “coupled” together. The coupling or connection between the elements can be physical, logical, or a combination thereof. As employed herein two elements may be considered to be “connected” or “coupled” together by the use of one or more wires, cables and/or printed electrical connections, as well as by the use of electromagnetic energy, such as electromagnetic energy having wavelengths in the radio frequency region, the microwave region and the optical (both visible and invisible) region, as several non-limiting and non-exhaustive examples.
Further, the formulas and expressions that use these various parameters may differ from those expressly disclosed herein. Further, the various names assigned to different channels (e.g., CPICH) are not intended to be limiting in any respect, as these various channels may be identified by any suitable names.
Furthermore, some of the features of the various non-limiting and exemplary embodiments of this invention may be used to advantage without the corresponding use of other features. As such, the foregoing description should be considered as merely illustrative of the principles, teachings and exemplary embodiments of this invention, and not in limitation thereof.
Filing Document | Filing Date | Country | Kind | 371c Date |
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PCT/FI2009/051014 | 12/18/2009 | WO | 00 | 6/15/2011 |
Number | Date | Country | |
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61203406 | Dec 2008 | US |