Partial cascode phase locked loop architecture

Abstract

Various embodiments for a partial cascode phase locked loop architecture are described. In one embodiment, an apparatus may include a phase locked loop circuit having a plurality of partial cascode circuits. The plurality of partial cascode circuits may be arranged to reduce phase noise from a ground power supply voltage and a power supply voltage. Other embodiments are described and claimed.

Description

BACKGROUND

Phase-locked loop (PLL) circuits are often used to reduce noise and improve timing throughout a circuit. Timing throughout a circuit becomes particularly critical for applications requiring high-speed processing of information, such as in communications applications and video processing applications. When noise is introduced by various system components, the timing may deviate from the system clock.

Variations in power supplies may increase noise and have a significant impact on overall system performance. Several shortcomings in the conventional PLL circuit lead to a low Power Supply Rejection Ratio (PSRR) in analog cells. Lower PSRR leads to higher phase noise in the PLL, which is not desirable for processing applications. Accordingly, there is a need for a PLL circuit that provides improved PSRR.

SUMMARY

One embodiment may include an apparatus comprising a phase locked loop circuit. The phase locked loop circuit may comprise a plurality of partial cascode circuits. The plurality of partial cascode circuits may include at least a first partial cascode circuit and a second partial cascode circuit. The first partial cascode circuit may be driven by a first bias voltage and may be connected to a ground supply voltage. The second partial cascode circuit may be driven by a second bias voltage and may be connected to a power supply voltage. The first partial cascode circuit may reduce phase noise from the ground power supply voltage. The second partial cascode circuit may reduce phase noise from the power supply voltage.

One embodiment may include a system comprising a self-biasing multiplier; and a voltage controlled oscillator to receive a first bias voltage and a second bias voltage from the self-biasing multiplier. At least one of the self-biasing multiplier and the voltage controlled oscillator may comprise a plurality of partial cascode circuits including at least a first partial cascode circuit and a second partial cascode circuit. The first partial cascode circuit may be driven by a first bias voltage and may be connected to a ground supply voltage. The second partial cascode circuit may be driven by a second bias voltage and may be connected to a power supply voltage. The first partial cascode circuit may reduce phase noise from the ground power supply voltage. The second partial cascode circuit may reduce phase noise the power supply voltage.

One embodiment may include a method to control operation frequency of a variable controlled oscillator comprising a plurality of delay cells. The method may comprise converting input voltage from a low-pass filter into low-pass filter transconductance, determining a time constant for each of the plurality of delay cells based on the low-pass filter transconductance and free run transconductance, determining a total time delay based on a plurality of the time constants, and controlling the operational frequency based on the total time delay.

Other embodiments are described and claimed.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates one embodiment of a partial cascode differential inverter voltage controlled oscillator (VCO).

FIG. 2 illustrates one embodiment of a VCO delay cell.

FIG. 3 illustrates one embodiment of a partial cascode circuit.

FIG. 4 illustrates one embodiment of an equivalent circuit of the partial cascode circuit of FIG. 3.

FIG. 5 illustrates one embodiment of a partial cascode circuit.

FIG. 6 illustrates one embodiment of an equivalent circuit of the partial cascode circuit of FIG. 5.

FIG. 7 illustrates one embodiment of an equivalent circuit of the VCO delay cell of FIG. 2.

FIG. 8 illustrates one embodiment of a time delay graph for the equivalent circuit of FIG. 8.

FIG. 9 illustrates one embodiment of a partial cascode self-biasing multiplier.

FIG. 10 illustrates one embodiment of a PLL circuit.

FIG. 11 illustrates one embodiment of a linear model of the PLL circuit of FIG. 10.

FIG. 12 illustrates one embodiment of a timing diagram.

FIG. 13 illustrates one embodiment of a loop filter.

DETAILED DESCRIPTION

Numerous specific details have been set forth herein to provide a thorough understanding of the embodiments. It will be understood by those skilled in the art, however, that the embodiments may be practiced without these specific details. In other instances, well-known operations, components and circuits have not been described in detail so as not to obscure the embodiments. It can be appreciated that the specific structural and functional details disclosed herein may be representative and do not necessarily limit the scope of the embodiments.

It is also worthy to note that any reference to “one embodiment” or “an embodiment” means that a particular feature, structure, or characteristic described in connection with the embodiment is included in at least one embodiment. The appearances of the phrase “in one embodiment” in various places in the specification are not necessarily all referring to the same embodiment.

Various embodiments may be directed to a PLL circuit architecture comprising a partial cascode differential inverter VCO and/or a partial cascode self-biasing multiplier (PCSBM). In various implementations, the partial cascode differential inverter VCO and the PCSBM may be arranged to provide lower PLL noise, significant PSRR improvement, higher tolerance for lower power supply voltage without requiring a large overhead voltage penalty, increased output impedance to help matching current biasing for the VCO, and/or enhanced calibration functionality for the VCO to compensate process variation and store the result in local memory for process compensation.

FIG. 1 illustrates one embodiment of a partial cascode differential inverter VCO 100. In various embodiments, the differential inverter VCO 100 may be arranged to generate a desired output frequency F_o in a PLL circuit architecture. As shown, the partial cascode differential inverter VCO 100 may comprise a plurality of VCO delay cells, such as VCO delay cells 102-1-4. In various implementations, bias voltages (Vbp, Vbn) may determine the amount of delay for each of the VCO delay cells 102-1-4 within the partial cascode VCO 100. The bias voltage Vbp and the bias voltage Vbn may be received, for example, from a PCSBM coupled to the partial cascode differential inverter VCO. It can be appreciated that although a limited number of VCO delay cells are shown by way of example, a greater number or fewer number of VCO delay cells may be employed for a given implementation.

In various embodiments, the VCO delay cells may be arranged such that voltage outputs of a particular VCO delay cell provide voltage inputs to a subsequent VCO delay cell. As shown in FIG. 1, for example, voltage outputs of a first VCO delay cell 102-1 provide voltage inputs to a second VCO delay cell 102-2. Voltage outputs of the second VCO delay cell 102-2 provide voltage inputs to a third VCO delay cell 102-3. Voltage outputs of the third VCO delay cell 102-3 provide the voltage inputs to a fourth VCO delay cell 102-4. Voltage outputs from the fourth VCO delay cell 102-4 are feedback as voltage inputs to the first VCO delay cell 102-1 and provided to a differential amplifier 104. The embodiments are not limited, however, to the example depicted by FIG. 1.

In various implementations, the input voltage of a low-pass filter (V_LPF) is used to control the frequency of oscillation of the partial cascode differential inverter VCO 100. Since V_LPF is used as input to self-bias for controlling current to the partial cascode differential inverter VCO 100, all differential VCO delay cells 104-1-4 with loaded capacitance (C_load) are charged and discharged by differential current sink/source. In various embodiments, the input voltage is converted to a current in a partial cascode self-biasing circuit which is multiplied and mirrored to each fully partial cascode VCO. Each fully partial cascode differential inverter in the VCO operates in current mode, providing a much wider operating frequency range with better PSRR and high common-mode noise immunity because of the partial cascode topology on both ends without the larger overhead voltage penalty. As a result, improvement in phase noise is achieved.

FIG. 2 illustrates one embodiment of a VCO delay cell 200. In various embodiments, the VCO delay cell 200 may comprise one of the delay cells 102-1-4 implemented by the partial cascode differential inverter VCO 100 shown in FIG. 1. The embodiments are not limited in this context.

In various embodiments, the VCO delay cell 200 may comprise a plurality of transistors, such as transistors (M1-M10) 202-1-10, for example. Each transistor may comprise a field effect transistor (FET) such as a junction FET (JFET), a metal-oxide semiconductor FET (MOSFET), or a metal semiconductor FET (MESFET), a bipolar junction transistor (BJT), or any other type of suitable transistor. The transistors may comprise n-type or p-type semiconductor material and may be fabricated using various silicon-based processes such as MOS, complementary MOS (CMOS), bipolar, bipolar CMOS (BiCMOS), and so forth. In one embodiment, the VCO delay cell 200 may comprise n channel transistors (M1-M4) 202-1-4 and p channel transistors (M5-M10) 202-5-10. The VCO delay cell 200 also may comprise a plurality of loaded capacitances, such as first loaded capacitance (C_L1) 204-1 and second loaded capacitance (C_L2) 204-2.

In various embodiments, the VCO delay cell 200 may comprise a plurality of partial cascode circuits, such as partial cascode circuits 206-1-3, for example. As shown in FIG. 2, a first partial cascode circuit 206-1 comprises n channel transistor (M1) 202-1 and n channel transistor (M2) 202-2. A second partial cascode circuit 206-2 comprises p channel transistor (M5) 202-5 and p channel transistor (M7) 202-7, and a third partial cascode circuit 206-3 comprises p channel transistor (M6) 202-6 and p channel transistor (M8) 202-8. In this embodiment, transistor (M1) 202-1 and transistor (M2) 202-2 of the first partial cascode circuit 206-1 are driven by bias voltage Vbn. Transistor (M5) 202-5 and transistor (M7) 202-7 of the second partial cascode circuit 206-2 are driven by bias voltage Vbp. Transistor (M6) 202-6 and transistor (M8) 202-8 of the third partial cascode circuit 206-3 also are driven by bias voltage Vbp.

In various embodiments, the first partial cascode circuit 206-1 may be connected to a ground supply voltage (Vss). The second partial cascode circuit 206-2 and the third partial cascode circuit 206-3 may be connected to a power supply voltage (Vdd). In such embodiments, the partial cascode circuits 206-1-3 may implement partial cascode topology to both ends (n and p) of the VCO delay cell 200 to provide a wider operating frequency range with increased PSRR and high common-mode noise immunity without a large overhead voltage penalty. For example, the first partial cascode circuit 206-1 may reduce phase noise and provide improved PSRR with respect to the ground supply voltage (Vss). The second partial cascode circuit 206-2 and the third partial cascode circuit 206-3 may provide reduce phase noise and provide improved PSRR with respect to the power supply voltage (Vdd). In various implementations, the ground supply voltage (Vss) may be a few mV, and the power supply voltage (Vdd) may be 1.8V, for example. The embodiments are not limited, however, to the example depicted by FIG. 2.

FIG. 3 illustrates one embodiment of a partial cascode circuit 300. In various embodiments, the partial cascode circuit 300 may comprise or be implemented as the first partial cascode circuit 206-1 of the VCO delay cell 200 shown in FIG. 2. The embodiments are not limited in this context.

In one embodiment, the partial cascode circuit 300 comprises n channel transistor (M1) connected in series to n channel transistor (M2). As shown, the source of transistor (M1) is connected to the drain of transistor (M2). The gate of transistor (M1) is driven by a bias voltage Vbn1, and the gate of transistor (M2) is driven by a bias voltage Vbn2. In various embodiments, the gate of transistor (M1) and the gate of transistor (M2) may be connected together and driven by a common bias voltage Vbn at a single node.

FIG. 4 illustrates one embodiment of an equivalent circuit 400 of the partial cascode circuit 300 shown in FIG. 3. In various embodiments, the following equations may characterize the operation of the equivalent circuit 400. $V_{\mathrm{gs}1}=-V_2$$V_2=I_O*r_{2\mathrm{ds}2}$$I_O=g_{m1}\left(-I_O*r_{\mathrm{ds}2}\right)+\frac{V_O}{r_{\mathrm{ds}1}}-I_O\frac{r_{\mathrm{ds}2}}{r_{\mathrm{ds}1}}$$I_O\left(1+g_{m1}{r}_{\mathrm{ds}2}+\frac{r_{\mathrm{ds}2}}{r_{\mathrm{ds}1}}\right)=\frac{V_O}{r_{\mathrm{ds}1}}$$R_O=\frac{V_O}{I_O}$$R_O=\frac{\left(1+g_{m1}{r}_{\mathrm{ds}2}+\frac{r_{\mathrm{ds}2}}{r_{\mathrm{ds}1}}\right)}{r_{\mathrm{ds}1}}\approx g_{m1}{r}_{\mathrm{ds}2}{r}_{\mathrm{ds}1}$

With respect to the foregoing equations, V_o is the output voltage, I_o is the output current, R_o is the output impedance, g_m1 is the small-signal transconductance of transistor (M1), r_ds1 is the drain-to-source channel resistance of transistor (M1), and r_ds2 is the drain-to-source channel resistance of transistor (M2). Accordingly, it can be demonstrated that the output impedance R_o of the partial cascode circuit 300 may be increased by approximately the common gate voltage gain of transistor (M1) multiplied by r_ds2 without requiring a large overhead voltage penalty. Thus, the partial cascode circuit 400 may be used to reduce noise and improve PSRR, for example.

FIG. 5 illustrates one embodiment of a partial cascode circuit 500. In various embodiments, the partial cascode circuit 500 may comprise or be implemented as the second partial cascode circuit 206-2 of the VCO delay cell 200 shown in FIG. 2. The embodiments are not limited in this context.

In one embodiment, the partial cascode circuit 500 comprises p channel transistor (M7) connected in series to p channel transistor (M5). As shown, the source of transistor (M7) is connected to the drain of transistor (M5). The gate of transistor (M7) is driven by a bias voltage Vbp1, and the gate of transistor (M5) is driven by a bias voltage Vbp2. In various embodiments, the gate of transistor (M7) and the gate of transistor (M5) may be connected together and driven by a common bias voltage Vbp at a single node.

FIG. 6 illustrates one embodiment of an equivalent circuit 600 of the partial cascode circuit 500 shown in FIG. 5. In various embodiments, the following equations may characterize the operation of the equivalent circuit 600. $V_{\mathrm{gs}1}=-V_2$$V_2=I_O*r_{\mathrm{ds}5}$$I_O=g_{m1}\left(-I_O*r_{\mathrm{ds}5}\right)+\frac{V_O}{r_{\mathrm{ds}7}}-I_O\frac{r_{\mathrm{ds}5}}{r_{\mathrm{ds}7}}$$I_O\left(1+g_{m5}{r}_{\mathrm{ds}5}+\frac{r_{\mathrm{ds}5}}{r_{\mathrm{ds}7}}\right)=\frac{V_O}{r_{\mathrm{ds}7}}$$R_O=\frac{V_O}{I_O}$$R_O=\frac{\left(1+g_{m5}{r}_{\mathrm{ds}5}+\frac{r_{\mathrm{ds}5}}{r_{\mathrm{ds}7}}\right)}{r_{\mathrm{ds}7}}\approx g_{m5}{r}_{\mathrm{ds}5}{r}_{\mathrm{ds}7}$

With respect to the foregoing equations, V_o is the output voltage, I_o is the output current, R_o is the output impedance, g_m5 is the small-signal transconductance of transistor (M5), r_ds5 is the drain-to-source channel resistance of transistor (M5), and r_ds7 is the drain-to-source channel resistance of transistor (M7). Accordingly, it can be demonstrated that the output impedance R_o of the partial cascode circuit 500 may be increased by approximately the common gate voltage gain of transistor (M5) multiplied by r_ds7 without requiring a large overhead voltage penalty. Thus, the partial cascode circuit 500 may be used to reduce noise and improve PSRR, for example.

Referring again to FIG. 2, the small signal output impedance of both ends of the VCO delay cell 200 may be increased by implementing the partial cascode topology. In various implementations, the small signal impedance of the n end of the VCO delay cell 200 may be increased by the common gate voltage gain of transistor (M1), and the small signal impedance of the p end of the VCO delay cell 200 may be increase by the common gate voltage gain of transistor (M5). As a result, PSRR may be improved on both ends without requiring a large overhead voltage which can lead to a smaller common input mode range. In addition, improvement in phase noise immunity may be achieved from both the ground supply and the power supply.

As shown in FIG. 2, transistor (M3) 202-3 may receive a voltage input Vin_p, and transistor (M4) 202-4 may receive a voltage input Vin_n. In various embodiments, transistor (M5) 202-5 and transistor (M7) 202-7 may act as a current source for transistor (M3) 202-3. When transistor (M3) 202-3 is not conducting, the supplied current does not pass through transistor (M3) 202-3. Transistor (M6) 202-6 and transistor (M8) 202-8 may act as a current source for transistor (M4) 202-4. When transistor (M4) 202-4 is not conducting, the supplied current does not pass through the transistor (M4) 202-4.

In various embodiments, transistor (M3) 202-3 and transistor (M4) 202-4 may act as switches and determine the actual delay for the VCO delay cell 200. For example, the delay provided by the VCO delay cell 200 may be the duration between turning on transistor (M3) 202-3 and turning off transistor (M4) 202-4, and when the voltages Vin_p and Vin_n are equal. At this point, transistors in the next VCO delay cell may be activated, and output voltages Vout_p and Vout_n of VCO delay cell 200 may be provided as input voltages Vin_p and Vin_n to the next delay cell.

In various implementations, first loaded capacitance (C_L1) 204-1 and second loaded capacitance (C_L2) 204-2 charge and discharge to affect the voltages Vin_p and Vin_n, which rise and fall. For instance, when transistor (M3) 202-3 is on and transistor (M4) 202-4 is off, the charges on the first loaded capacitance 204-1 and the second loaded capacitance 204-2 will be affected. In various embodiments, when transistor (M3) 202-3 is on and transistor (M4) 202-4 is off, first loaded capacitance (C_L1) 204-1 charges and second loaded capacitance (C_L2) 204-2 discharges. The charging of first loaded capacitance (C_L1) 204-1 may result in Vout_p changing from low (V_L) to high (V_H) at saturation. The discharging of second loaded capacitance (C_L2) 204-2 may result in Vout_n changing from high (V_H) to low (V_L). As shown in FIG. 2, transistor (M9) 202-9 and transistor (M10) 202-10 may be arranged to provide a smaller overhead voltage such that transistor (M3) 202-3 and transistor (M4) 202-4 are prevented from moving out off the saturation mode while Vout_n is crossing Vout_p.

FIG. 7 illustrates one embodiment of an equivalent circuit 700 of VCO delay cell 200 shown in FIG. 2. As shown, the equivalent circuit 700 comprises a loaded capacitance (C_L), which is charged by a current source (I) when a switch is open and discharges to provide a current source (2I) when the switch is closed. FIG. 8 illustrates one embodiment of a time delay graph 800 for the equivalent circuit 700. The embodiments are not limited in this context.

In various embodiments, the following equations may characterize the operation of equivalent circuit 700.

Since the slop of the voltage across C_L is $V_d={\mathrm{dV}}_{\mathrm{CL}1}-{\mathrm{dV}}_{\mathrm{CL}2}$${\mathrm{dV}}_{\mathrm{CL}1}=\frac{I_1}{C_{L1}}\mathrm{dt}$${\mathrm{dV}}_{\mathrm{CL}2}=\frac{I_2}{C_{L2}}\mathrm{dt}$$V_d=\left(V_H-{\mathrm{dV}}_{\mathrm{CL}1}\right)-\left(V_L-{\mathrm{dV}}_{\mathrm{CL}2}\right)$$V_d=\left(V_H-\frac{I_1}{C_{L1}}\mathrm{dt}\right)-\left(V_L+\frac{I_2}{C_{L2}}\mathrm{dt}\right)$$V_d-V_H+V_L=-\frac{I_1}{C_{L1}}\mathrm{dt}-\frac{I_2}{C_{L2}}\mathrm{dt}$$I_1=I_2=I$$C_{L1}=C_{L2}=C\left(V_H-V_L\right)-V_d=\frac{2I}{C}\mathrm{dt}$$\mathrm{dt}=\frac{\left(\left(V_H-V_L\right)-V_d\right)C}{2}\frac{C}{I}$$\frac{dt}{d\mathrm{Vlpf}}\frac{d\mathrm{Vlpf}}{dI}=\frac{\left(\left(V_H-V_L\right)-V_d\right)}{2}\frac{C}{\frac{dI}{d\mathrm{Vlpf}}}$$\frac{dt}{d\mathrm{Vlpf}}\frac{d\mathrm{Vlpf}}{dI}=\frac{\left(\left(V_H-V_L\right)-V_d\right)C}{2\beta \left(V_{\mathrm{gs}}-V_{\mathrm{th}}\right)}$

In various embodiments, VCO operation may be based on the transconductance of a self-biased multiplier (SBM). For example, where $I=\frac{\beta }{2}{\left(V_{\mathrm{gs}}-V_{\mathrm{th}}\right)}^2$ and V_gs=V_lp in SBM, a time constant (t) may determined as follows: $t=\frac{\left(\left(V_H-V_L\right)-V_d\right)C}{g_{\mathrm{m\_lpf}}+g_{\mathrm{m\_Free}\mathrm{\_runf}}}$

As demonstrated in the foregoing equation, the time constant (t) is a function of low-pass filter (LPF) transconductance (g_m—lpf) and free run transconductance (g_{m —Free—runf}). In various embodiments, g_{m—Free—runf} is a fixed frequency and is not a function of LPF voltage, while g_m—lpfis dynamic.

In various implementations, the VCO operational frequency may be based on a total time delay (T) comprising time constants for a plurality of VCO delay cells. For example, in a VCO comprising three VCO delay cells, the total time delay (T) may be determined as follows: $\frac{T}{2}=t_1+t_2+t_3$$\frac{T}{2}=\frac{d{t}_1}{d\mathrm{Vlpf}}\frac{d\mathrm{Vlpf}}{dI}+\frac{d{t}_2}{d\mathrm{Vlpf}}\frac{d\mathrm{Vlpf}}{dI}+\frac{d{t}_3}{d\mathrm{Vlpf}}\frac{d\mathrm{Vplf}}{dI}$$T=\frac{N\left(\left(V_H-V_L\right)-V_d\right)C}{g_{\mathrm{m\_lpf}}+g_{\mathrm{m\_Free}\mathrm{\_runf}}}$

In various embodiments, g_{m—Free—runf} would be multiplied by a multiplier circuit in order to have tuning condition on the VCO for N number of delay cells.

Based on the foregoing, in various embodiments, the VCO frequency of operation (F_VCO) and the gain for the VCO transfer function (K_VCO) may be expressed as follows: $F_{\mathrm{VCO}}=\frac{1}{T}=\frac{\left(g_{\mathrm{m\_lpf}}+g_{\mathrm{m\_Free}\mathrm{\_runf}}\right)}{N\left(\left(V_H-V_L\right)-V_d\right)C}$$K_{\mathrm{VCO}}=\frac{d{F}_{\mathrm{VCO}}}{d{V}_{\mathrm{lpf}}}=\frac{{\beta }_{\mathrm{m\_lpf}}}{N\left(\left(V_H-V_L\right)-V_d\right)C}$ The embodiments, however, are not limited in this context.

FIG. 9 illustrates one embodiment of a PCSBM 900. In various embodiments, the PCSBM 900 may be arranged to provide bias voltage Vbp and bias voltage Vbn to the partial cascode differential inverter VCO 100 of FIG. 1. For example, the PCSBM 900 may convert the input voltage from a low-pass filer (LPF) to current which is multiplied and mirrored to each fully partial cascode VCO. The embodiments are not limited in this context.

As shown in FIG. 9, the PCBM 900 may comprise bias generator portion 902 and current multiplier portion 904. In various implementations, the bias generator portion 902 may receive input from LPF 906 and provide output to current multiplier portion 904. The LPF 906 also may provide input to VCO calibration unit 908, which provides input to current multiplier portion 904.

In various embodiments, the bias generator portion 902 of the PCSBM 900 may comprise a differential amplifier 910 and plurality of transistors, such as transistors 912-1-6, for example. In one embodiment, for example, the bias generator portion 902 of the PCSBM 900 may comprise n channel transistors 912-1 and 912-2 and p channel transistors 912-3-6. Each transistor may comprise a FET, BJT, or any other type of suitable transistor.

In various embodiments, the bias generator portion 902 of the PCSBM 900 may comprise a plurality of partial cascode circuits, such as partial cascode circuits 914-1-3, for example. As shown, partial cascode circuit 914-1 comprises n channel transistor 912-1 and n channel transistor 912-2, partial cascode circuit 914-2 comprises p channel transistor 912-3 and p channel transistor 912-5, and partial cascode circuit 914-3 comprises p channel transistor 912-4 and p channel transistor 912-6.

In various embodiments, the partial cascode circuit 914-1 may be connected to a ground supply voltage (Vss). The partial cascode circuit 914-2 and partial cascode circuit 914-3 may be connected to a power supply voltage (Vdd). In such embodiments, the partial cascode circuits 914-1-3 may implement partial cascode topology to provide a wider operating frequency range with increased PSRR and high common-mode noise immunity without a large overhead voltage penalty. For example, the partial cascode circuit 914-1 may reduce phase noise and provide improved PSRR with respect to the ground supply voltage (Vss). The partial cascode circuit 914-2 and the partial cascode circuit 914-3 may reduce phase noise and provide improved PSRR with respect to the power supply voltage (Vdd). In various implementations, the ground supply voltage (Vss) may be a few mV, and the power supply voltage (Vdd) may be 1.8V, for example.

In various embodiments, the current multiplier portion 904 of the PCSBM 900 may comprise a first transconductance (g_m—lpf) 916, a second transconductance (g_{m—Free—run}) 918, a summing unit 920, and a plurality of transistors, such as transistors 922-1-6, for example. In one embodiment, for example, the current multiplier portion 904 of the PCSBM 900 may comprise n channel transistors 922-1 and 922-2 and p channel transistors 922-3-6. Each transistor may comprise a FET, BJT, or any other type of suitable transistor.

In various embodiments, the current multiplier portion 904 of the PCSBM 900 may comprise a plurality of partial cascode circuits, such as partial cascode circuits 924-1-3, for example. As shown, partial cascode circuit 924-1 comprises n channel transistor 922-1 and n channel transistor 922-2, partial cascode circuit 924-2 comprises p channel transistor 922-3 and p channel transistor 922-5, and partial cascode circuit 924-3 comprises p channel transistor 922-4 and p channel transistor 922-6.

In various embodiments, the partial cascode circuit 924-1 may be connected to a ground supply voltage (Vss). The partial cascode circuit 924-2 and partial cascode circuit 924-3 may be connected to a power supply voltage (Vdd). In such embodiments, the partial cascode circuits 924-1-3 may implement partial cascode topology to provide a wider operating frequency range with increased PSRR and high common-mode noise immunity without a large overhead voltage penalty. For example, the partial cascode circuit 924-1 may reduce phase noise and provide improved PSRR with respect to the ground supply voltage (Vss). The partial cascode circuit 924-2 and the partial cascode circuit 924-3 may reduce phase noise and provide improved PSRR with respect to the power supply voltage (Vdd). In various implementations, the ground supply voltage (Vss) may be a few mV, and the power supply voltage (Vdd) may be 1.8V, for example.

In various implementations, the partial cascode circuits 914-1-3 and partial cascode circuits 924-1-3 of the PCSBM 900 may comprise partial cascode topology to provide a wider operating frequency range with increased PSRR and high common-mode noise immunity without a large overhead voltage penalty. In various embodiments, the bias generator portion 902 and the current multiplier portion 904 of the PCSBM 900 may be arrange to interface with each other and with a partial cascode differential inverter VCO, such as partial cascode differential inverter VCO 100. In such embodiments, the partial cascode differential inverter VCO 100 and the PCSMB 900 may implement fully partial cascode topology to ensure improved PSRR.

In various embodiments, if the voltage-controlled current sources have voltage to current linearity, then the transfer relationship may be expressed as follows: I_d=g_m(V_LPF)+g_m(V_Free—run) where $g_{\mathrm{m\_SB}}=\frac{d{I}_1}{d{V}_{\mathrm{lpf}}}$$V_{\mathrm{gs}}=V_{\mathrm{lpf}}$$I=\frac{\beta }{2}{\left(V_{\mathrm{gs}}-V_{\mathrm{th}}\right)}^2$$g_{\mathrm{m\_SB}}=\frac{d{I}_1}{d{V}_{\mathrm{lpf}}}$$g_{\mathrm{m\_SBM}}=\frac{d{I}_1}{d{V}_{\mathrm{lpf}}}+\frac{d{I}_2}{d{V}_{\mathrm{Free}}}$$g_{\mathrm{m\_SBM}}=g_{\mathrm{m\_lpf}}+\left(M\times g_{\mathrm{m\_Free}\mathrm{\_runf}}\right)$

With respect to the foregoing equations, g_m—SB is the self-bias transconductance, and g_m—SB is the is the transconductance of the self-biasing multiplier transconductance. In various embodiments, the LPF voltage is converted into transconduntance g_m—SBM which is used to set the operational frequency of the partial cascode differential inverter VCO 100.

In various embodiments, the PCSBM 900 may provide several multiplication ranges, such as N² multiplication ranges. In one embodiment, for example, the PCSBM 900 may provide 4-bit control and 16 multiplication ranges. In various implementations, the PCSBM 900 may achieve high tolerance for process variations by calibrating the current range in the PCSBM for a specific operational frequency request. In addition, the partial cascode topology in the analog cells provides an improvement in PSRR, without requiring a large overhead voltage. As a result, the PCSBM 900 may provide lower phase noise (e.g., VCO output jitter) without a large overhead voltage penalty and significant improvement for low supply voltage process.

In various implementations, the PCSBM 900 provides the necessary bias with lower sensitivity to temperature changes, process variations, and voltage drop on the power supply, while providing better PSRR and self-calibration current setting range with lower VCO gain (KVCO). The PCSBM 900 may be arranged to provide lower PLL noise, significant PSRR improvement, higher tolerance for lower power supply voltage without requiring a large overhead voltage penalty, increased output impedance to help matching current biasing for the VCO, and/or enhanced calibration functionality for the VCO to compensate process variation and store the result in local memory for process compensation.

FIG. 10 illustrates one embodiment of a PLL circuit 1000. In various embodiments, the PLL circuit 1000 may comprise PFD 1002, PFD buffer 1004, charge pump 1006, loop filter 1008 including capacitors (C1, C2) and resistor (R1), LPF 1010 (e.g., 1/RC circuit), PCSBM 1012, VCO calibration unit 1014, VCO 1016, frequency divider 1018, lock detect 1020, and loop reset 1022. The embodiments are not limited in this context.

In various implementations, the PFD 1002 determines the phase and frequency difference between a reference frequency F_ref and the divided output frequency signal F_o/N from the frequency divider 1018. If a difference is detected, the PFD 1002 sends error signals Up, Down to the charge pump 1006. The duration of the error signals may depend on the amount of phase and frequency error detected by the PFD 1002.

In various embodiments, the charge pump 1006 receives the error signals Up, Down and a reference bias voltage Vbp which control the charge pump output current. The output current generated by the charge pump 1006 charges or discharges the capacitors (C₁, C₂) of loop filter 106 to a voltage level V_LPF. The voltage V_LPF is used as a reference for the PCSBM 1012 to generate reference signals Vbp, Vbn to control the output frequency F_o of the VCO 1016.

In various embodiments, the PCSBM 1012 may comprise or be implemented by the PCSBM 900 of FIG. 9, and the VCO 1016 may comprise or be implemented by the partial cascode differential inverter VCO 100 of FIG. 1. In such embodiments, the PLL circuit 1000 may be arranged to provide lower PLL noise, significant PSRR improvement, higher tolerance for lower power supply voltage without requiring a large overhead voltage penalty, increased output impedance to help matching current biasing for the VCO, and/or enhanced calibration functionality for the VCO to compensate process variation and store the result in local memory for process compensation.

In various embodiments, the charge pump 1006 may comprise a partial cascode charge pump as described in co-pending U.S. patent application Ser. No. 11/186,000. In such embodiments, the partial cascode charge pump may implement a common current node and high output impedance architecture providing improved current matching between sink and source currents at the output. As a result, better matching in sink and source current improves phase noise and jitter as well as tolerance of process and temperature variations in matching the sink and source current outputs.

While the PLL circuit 1000 generally may be generally represented as a nonlinear system for the purpose of investigating its dynamic behavior, linear approximation may be useful to understand the functionality and trade-offs in the PLL circuit 1000.

FIG. 11 illustrates one embodiment of a linear model 1100 of PLL circuit FIG. 10. In various embodiments, the open loop transfer function of this model is: $\begin{array}{cc}H\left(s\right)=K_{\mathrm{PD}}{K}_F\left(\frac{K_{\mathrm{VCO}}}{S}\right)& \left(1\right)\end{array}$ where K_PD is phase frequency detector gain, K_F is LPF gain and $K_{\mathrm{VCO}}=\frac{{\omega }_{\mathrm{Max}}-{\omega }_{\mathrm{Min}}}{V_{\mathrm{Max}}-V_{\mathrm{Min}}}$ the gain for the VCO transfer function in radians per second. V_Max and V_Min are maximum and minimum input control voltage for maximum and minimum (ω_Max−ω_Min) output frequency of VCO.

Substituting the gain transfer function for the LPF, $K_F=\frac{S{c}_{1}R+1}{s^2{c}_1{c}_{2}R+S\left(c_1+c_2\right)}$ and PFD, $K_{\mathrm{PD}}=\frac{I_P}{2\pi }$ (Amp/radian) in to (1) yields: $H\left(s\right)=\frac{I_P}{2\pi }\frac{K_{\mathrm{VCO}}}{S}\frac{S{C}_{1}R+1}{S^2{C}_1{C}_{2}R+S\left(C_1+C_2\right)}$

In order to simplify the open-loop transfer function, the expression for H(jω) may be put in a standard form as follows: $H\left(j\omega \right)=\frac{I_P{K}_{\mathrm{VCO}}}{2\pi }\frac{j\omega C_{1}R+1}{{\left(j\omega \right)}^3{C}_1{C}_{2}R+{\left(j\omega \right)}^2\left(C_1+C_2\right)}$ where $K=\frac{I_P{K}_{\mathrm{VCO}}}{2\pi }$ is open loop gain in radians per second.

Next, H(jω) may be written in polar form as follows: $H\left(j\omega \right)=\frac{Kj\omega C_{1}R+1}{-j{\omega }^3{C}_1{C}_{2}R-{\omega }^2\left(C_1+C_2\right)}\angle \left({\tan }^{-1}\omega {\mathrm{Rc}}_1-{\tan }^{-1}\frac{\omega R{C}_1{C}_2}{C_{1}a+C_2}\right)$

Where magnitude and phase are: $H\left(j\omega \right)=\frac{I_P{K}_{\mathrm{VCO}}}{2\pi }\frac{j\omega c_{1}R+1}{-j{\omega }^3{c}_1{c}_{2}R+{\omega }^2\left(c_1+c_2\right)}$${\theta }_h\left(\omega \right)=+{\tan }^{-1}\frac{\omega }{Z_1}-{\tan }^{-1}\frac{\omega }{P_3}+{180}^{°}$${\theta }_h\left(\omega \right)=+{\tan }^{-1}\frac{\omega }{\frac{1}{c_{1}R}}-{\tan }^{-1}\frac{\omega }{\frac{c_1+c_2}{c_1{c}_{2}R}}+{180}^{°}$

Poles and zero of the open-loop may be defined as: $s^3{c}_1{c}_{2}R+s^2\left(c_1+c_2\right)$$P_1=0$$P_2=0$$P_3=\frac{c_1+c_2}{c_1{c}_{2}R}$$Z_1=\frac{1}{c_{1}R}$

Propagation delay, which is caused by the Poles and Zero, can be defined as: $t_{\mathrm{pd}}=\frac{1}{P_3}=\frac{c_1{c}_{2}R}{c_1+c_2}$$t_{\mathrm{dZ}}=\frac{1}{Z_1}=c_{1}R$

The simple model closed-loop transfer function is: φ_O=(φ_in−φ_Oβ)H(S)

Where H (S) is the Open-Loop transfer function of the system, and β is the loop divider: ${\varphi }_O=\left({\varphi }_{\mathrm{in}}-{\varphi }_O\beta \right)K_{\mathrm{PD}}{K}_F\left(\frac{K_{\mathrm{VCO}}}{S}\right)$${\varphi }_O\left(1+\beta K_{\mathrm{PD}}{K}_F\left(\frac{K_{\mathrm{VCO}}}{S}\right)\right)={\varphi }_{\mathrm{in}}{K}_{\mathrm{PD}}{K}_F\left(\frac{K_{\mathrm{VCO}}}{S}\right)$$H\left(s\right)=\frac{{\varphi }_O}{{\varphi }_{\mathrm{in}}}=\frac{K_{\mathrm{PD}}{K}_F\left(\frac{K_{\mathrm{VCO}}}{S}\right)}{\left(1+\beta K_{\mathrm{PD}}{K}_F\left(\frac{K_{\mathrm{VCO}}}{S}\right)\right)}$

Substituting the transfer function for the LPF, $K_F=\frac{{\mathrm{Sc}}_{1}R+1}{S^2{c}_1{c}_{2}R+S\left(c_1+c_2\right)}$ in to H(s) yields: $H\left(s\right)=\frac{V_O}{V_{\mathrm{in}}}=\frac{K_{\mathrm{PD}}{K}_{\mathrm{VCO}}\frac{{\mathrm{Sc}}_{1}R+1}{S^2{c}_1{c}_{2}R+S\left(c_1+c_2\right)}}{S+\beta K_{\mathrm{PD}}{K}_{\mathrm{VCO}}\frac{{\mathrm{Sc}}_{1}R+1}{S^2{c}_1{c}_{2}R+S\left(c_1+c_2\right)}}$$H\left(s\right)=\frac{V_O}{V_{\mathrm{in}}}=\frac{K_{\mathrm{PD}}{K}_{\mathrm{VCO}}\frac{{\mathrm{Sc}}_{1}R+1}{S^2{c}_1{c}_{2}R+S\left(c_1+c_2\right)}}{S+\beta K_{\mathrm{PD}}{K}_{\mathrm{VCO}}\frac{\hspace{0.17em}{\mathrm{Sc}}_{1}R+1}{S^2{c}_1{c}_{2}R+S\left(c_1+c_2\right)}}$$H\left(s\right)=\frac{V_O}{V_{\mathrm{in}}}=\frac{K_{\mathrm{PD}}{K}_{\mathrm{VCO}}\left({\mathrm{Sc}}_{1}R+1\right)}{S^3{c}_1{c}_{2}R+S^2\left(c_1+c_2\right)+\beta K_{\mathrm{PD}}{K}_{\mathrm{VCO}}\left({\mathrm{Sc}}_{1}R\right)+K_{\mathrm{VCO}}\beta K_{\mathrm{PD}}}$

Substituting the transfer function for the PFD, $K_{\mathrm{PD}}=\frac{I_P}{2\pi }$ in to H(s) yields $H\left(s\right)=\frac{K_{\mathrm{VCO}}\frac{I_P}{2\pi }\left({\mathrm{Sc}}_{1}R+1\right)}{S^3{c}_1{c}_{2}R+S^2\left(c_1+c_2\right)+\beta \frac{I_P}{2\pi }{K}_{\mathrm{VCO}}\left({\mathrm{Sc}}_{1}R\right)K_{\mathrm{VCO}}\beta \frac{I_P}{2\pi }}$

Dividing the numerator and denominator by RC₁ yields: $H\left(s\right)=\frac{K_{\mathrm{VCO}}\frac{I_P}{2\pi }\left(\hspace{0.17em}{\mathrm{Sc}}_{1}R+1\right)}{S^3{c}_1{c}_{2}R2\pi +S^2\left(c_1+c_2\right)2\pi +\frac{I_P}{2\pi }{K}_{\mathrm{VCO}}\left(1+\hspace{0.17em}S{c}_{1}R\right)}$

The expression for H(s) may be put in standard form by dividing out the poles and zeros to yield: $H\left(s\right)=\frac{K_{\mathrm{VCO}}{I}_P\left(S+\frac{1}{c_{1}R}\right)}{c_{2}2\pi \left(S^3+S^2\left(\frac{1}{c_{2}R}+\frac{1}{c_{1}R}\right)+S\frac{K_{\mathrm{VCO}}}{N}\frac{I_P}{2\pi N{c}_2}+\frac{I_P{K}_{\mathrm{VCO}}}{2\pi NR{c}_1{c}_2}\right)}$

The denominator of the closed-loop transfer function may be converted to a control theory form as follows: $\begin{array}{c}H\left(s\right)=\frac{K\left(S+z_1\right)}{\left(S+P_1\right)\left(S+P_2\right)\left(S+P_3\right)}\\ =\frac{K\left(S+z_1\right)}{\left(S+P_1\right)\left(S+\alpha -\mathrm{j\beta }\right)\left(S+\alpha +j\beta \right)}\end{array}$$H\left(s\right)=\frac{K\left(S+z_1\right)}{\left(S+P_1\right)(S^2+2S\alpha +\left({\alpha }^2+{\beta }^2\right)}$$\begin{array}{c}H\left(s\right)=\frac{K\left(S+z_1\right)}{\left(S+P_1\right)\left(S+2{\mathrm{\zeta \omega }}_{n}S+{\omega }_n^2\right)}\\ =\frac{K{z}_1\left(\frac{S}{z_1}+1\right)}{P_1\left(\frac{S}{P_1}+1\right){\omega }_n^2\left({\left(\frac{S}{{\omega }_n}\right)}^2+2\zeta \frac{S}{{\omega }_n}+1\right)}\end{array}$

The phase as function of co may be defined as follows: $\theta \left(\omega \right)={\tan }^{-1}\frac{\omega }{z_1}-{\tan }^{-1}\frac{\omega }{P_1}{\tan }^{-1}\frac{2\zeta \frac{\omega }{{\omega }_n}}{1-\frac{\omega }{{\omega }_n}}$ where ω_n=α²+β² is the comer frequency of the quadratic factor and Zeta $\zeta =\frac{{\omega }_n^2}{2{\omega }_n}$ is the damping coefficient of the quadratic term.

FIG. 12 illustrates one embodiment of a timing diagram 1200. As shown, it may be assumed that the input phase θ_Vin and the output (feedback) phase θ_Fb change little in any given frequency, where θ_Fb=θ_Vout/N. The embodiments are not limited in this context.

FIG. 13 illustrates one embodiment of a loop filter 1300. In various embodiments, small signal analysis may be used to calculate values of C₁, C₂ and R. In such analysis, the average charge which is follow in to the low-pass filter (e.g., LPF 1010 of FIG. 10) can be defined as: $I_{\mathrm{avg}}=\frac{\Delta {\varphi }_{\mathrm{IN}}}{2\pi }{I}_{\mathrm{ch}}$$\mathrm{where}\text{:}$$I_{\mathrm{avg}}=K_{\mathrm{pd}}\left({\varphi }_{\mathrm{IN}2}-{\varphi }_{\mathrm{IN}1}\right)=K_{\mathrm{pd}}\Delta {\varphi }_{\mathrm{IN}}\mathrm{and}$$K_{\mathrm{pd}}=\frac{I_{\mathrm{ch}}}{2\pi }$

To simplify the close loop transfer function, C₂ is assumed to equal to 0, and the expression for H(jω) is put in standard form as follows: $H\left(j\omega \right)=\frac{K_{\mathrm{PD}}{K}_{\mathrm{VCO}}\left(S{c_1}^{R+1}\right)}{S^2+S\frac{K_{\mathrm{PD}}{K}_{\mathrm{VCO}}R}{N}+\frac{K_{\mathrm{PD}}{K}_{\mathrm{VCO}}}{{}^N{C}_1}}$

Next, C₁ can be defined by ${\omega }_n=\frac{K_{\mathrm{PD}}{K}_{\mathrm{VCO}}}{{}_{}{}_{}{}^{N}C_1^{}}$ for desired transient settling time of the complete loop for phase or frequency changes given by $T_l=\frac{1}{{\omega }_n}$ and R by $\zeta =\frac{{\omega }_n}{2}{}_{}{}^{N}C_1^{}.$

Finally, C₂ may be set to around one-twentieth the size of C₁ to minimize glitches. It is noted that ω_n should not be selected very close to ω_c where the delay around the sampled feedback loop causes a loss in phase margin and takes the system into the unstable condition. Accordingly, ${\omega }_n\le \frac{{\omega }_c}{10}$ should be maintained in order have good phase margin.

With respect to the loop-filter, C₂ initially may be assumed equal to zero to simplify the loop-filter transfer function. In the second order loop-filter while capacitor C₂ is used to keep I_Cp×R from causing voltage jumps at the input of VCO, C₂ can lead to frequency jump at the output of VCO. Most of the voltage to bring the VCO to the proper frequency is provided by the C₁ in the loop filter. In general, C₂ is set about one-twentieth of C₁ or more. ${H\left(s\right)}_{\mathrm{LPF}}=\left(R+\frac{1}{{}^S{C}_1}\right)\left(\frac{1}{{}^S{C}_2}\right)=\left(\frac{{}^S{C}_1^{R+1}}{S^2{C}_1{C}_2^{R+S}{C}_2{+}^S{C}_2}\right)V_{\mathrm{in\_VCO}}=H\left(S\right)⨯I_{\mathrm{PDI}}$

Where ${H\left(s\right)}_{\mathrm{LPF}}=\frac{{}^S{C}_1^{R+1}}{S^2{C}_1{C}_2^{R+S}\left(C_2+C_1\right)}$ poles and zero would be: $s^3{c}_1{c}_2^{R+}{s}^2\left(c_1+c_2\right)$$P_1=0$$P_2=0$$P_3=\frac{c_1+c_2}{c_1{c}_2^R}$$Z_1=\frac{1}{c_1^R}$

Propagation delay, which is caused by poles and zero can be defined as: $t_{\mathrm{pd}}=\frac{1}{P_3}=\frac{c_1{c}_2^R}{c_1+c_2},$ it can be also called time constant. $t_{\mathrm{dZ}}=\frac{1}{Z_1}=c_{1}R$

Next, H(jω) may be written in polar form as follows: $H\left(\mathrm{jw}\right)=\frac{\sqrt{{\left(\omega C_{1}R\right)}^2+1^2}}{\sqrt{{\left(\omega C_2{C}_{1}R\right)}^2+{\left(\omega \left(C_2+C_1\right)\right)}^2}}{\mathrm{\angle tan}}^{-1}\omega C_{1}R-{\tan }^{-1}\frac{c_1{c}_{2}R}{c_1+c_2}$

Where magnitude and phase of LPF are: ${H\left(S\right)}_{\mathrm{LPF}}=\frac{\sqrt{{\left(\omega C_{1}R\right)}^2+1^2}}{\sqrt{{\left(\omega C_2{C}_{1}R\right)}^2+{\left(\omega \left(C_2+C_1\right)\right)}^2}}$$\theta \left(\omega \right)={\tan }^{-1}\omega C_{1}R-{\tan }^{-1}\frac{c_1{c}_{2}R}{c_1+c_2}$

In various embodiments, the key equations for close loop condition may be expressed as follows: $K_0=\frac{I_{\mathrm{CP}}{K}_{\mathrm{VCO}}R}{2\pi N}$${\omega }_n=\sqrt{\frac{I_{\mathrm{CP}}{K}_{\mathrm{VCO}}}{2\pi {\mathrm{NC}}_1}}$${\omega }_Z=\frac{1}{C_{1}R}$${\omega }_C=\frac{C_1+C_2}{{\mathrm{RC}}_1{C}_2}$$\xi =\sqrt{\frac{I_{\mathrm{CP}}{K}_{\mathrm{VCO}}{R}^2{C}_1}{8\pi N}}$$T_L=\frac{1}{{\omega }_n}$ The embodiments are not limited in this context.

While certain features of the embodiments have been illustrated as described herein, many modifications, substitutions, changes and equivalents will now occur to those skilled in the art. It is therefore to be understood that the appended claims are intended to cover all such modifications and changes as fall within the true spirit of the embodiments.

Claims

1. An apparatus comprising: a phase locked loop circuit comprising a plurality of partial cascode circuits, said plurality of partial cascode circuits including at least a first partial cascode circuit and a second partial cascode circuit, said first partial cascode circuit to be driven by a first bias voltage and to be connected to a ground supply voltage, said second partial cascode circuit to be driven by a second bias voltage and to be connected to a power supply voltage, wherein said first partial cascode circuit is to reduce phase noise from said ground power supply voltage and said second partial cascode circuit is to reduce phase noise from said power supply voltage.

2. The apparatus of claim 1, wherein said first partial cascode is to provide an output impedance to reduce said phase noise from said ground supply voltage.

3. The apparatus of claim 1, wherein said second partial cascode is to provide an output impedance to reduce said phase noise from said power supply voltage.

4. The apparatus of claim 1, wherein said first partial cascode circuit comprises a plurality of n channel transistors and said second partial cascode comprises a plurality of p channel transistors.

5. The apparatus of claim 1, wherein said plurality of partial cascode circuits comprises a third partial cascode circuit to be driven by said second bias voltage.

6. The apparatus of claim 5, wherein said third partial cascode circuit comprises a plurality of p channel transistors.

7. The apparatus of claim 1, said phase locked loop circuit comprising a voltage controlled oscillator including said plurality of partial cascode circuits.

8. The apparatus of claim 7, said voltage controlled oscillator comprising a plurality of delay cells.

9. The apparatus of claim 8, wherein at least one of said plurality of delay cells is to provide voltage outputs as voltage inputs to another one of said plurality of delay cells.

10. The apparatus of claim 7, said voltage controlled oscillator to receive said first bias voltage and said second bias voltage from a self-biasing multiplier.

11. The apparatus of claim 7, said voltage controlled oscillator comprising a first loaded capacitance and a second loaded capacitance to charge and discharge to provide a delay.

12. The apparatus of claim 1, said phase locked loop circuit comprising a self-biasing multiplier including said plurality of partial cascode circuits.

13. The apparatus of claim 12, said self-biasing multiplier comprising a bias generator portion, said bias generator portion comprising said plurality of partial cascode circuits.

14. The apparatus of claim 12, said self-biasing multiplier comprising a current multiplier portion, said current multiplier portion comprising said plurality of partial cascode circuits.

15. The apparatus of claim 12, said self-biasing multiplier to provide said first bias voltage and said second bias voltage to a voltage controlled oscillator.

16. A system comprising: a self-biasing multiplier; and a voltage controlled oscillator to receive a first bias voltage and a second bias voltage from said self-biasing multiplier, at least one of said self-biasing multiplier and said voltage controlled oscillator comprising: a plurality of partial cascode circuits including at least a first partial cascode circuit and a second partial cascode circuit, said first partial cascode circuit to be driven by a first bias voltage and to be connected to a ground supply voltage, said second partial cascode circuit to be driven by a second bias voltage and to be connected to a power supply voltage, wherein said first partial cascode circuit is to reduce phase noise from said ground power supply voltage and said second partial cascode circuit is to reduce phase noise from said power supply voltage.

17. The system of claim 16, wherein said plurality of partial cascode circuits comprises a third partial cascode circuit to be driven by said second bias voltage.

18. The system of claim 16, said self-biasing multiplier may to provide N2 multiplication ranges to calibrate a current range for a specific operational frequency of said voltage controlled oscillator.

19. The system of claim 18, wherein said N2 multiplication ranges comprises 16 multiplication ranges.

20. A method to control operation frequency of a variable controlled oscillator comprising a plurality of delay cells, the method comprising: converting input voltage from a low-pass filter into low-pass filter transconductance; determining a time constant for each of said plurality of delay cells based on said low-pass filter transconductance and free run transconductance; determining a total time delay based on a plurality of said time constants; and controlling said operational frequency based on said total time delay.